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IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 13, NO. 1, JANUARY 1998 47
Predicted (On-Time) Equal-Charge Criterion Scheme
for Constant-Frequency Control of Single-Phase
Boost-Type AC–DC Converters
Ramesh Oruganti, Member, IEEE, Kannan Nagaswamy, and Lock Kai Sang
Abstract—A new constant switching frequency control method
for single-phase boost-type ac–dc converters is presented. The
on times of the converter switches in each switching period is
determined such that the average input current tracks the refer-
ence template in every switching cycle. The problems encountered
in achieving smooth and stable operation and the modifications
made to overcome them are discussed. The simulation studies
done on the converter controlled with this method, which is given
the name predicted (on-time) equal-charge criterion (PECC)
method, indicate stable operation at different input-current and
voltage levels and power factors. The method was implemented
on an insulated gate bipolar transistor (IGBT) converter rated
for 1 kVA using a 80386 processor system for computations. The
experimental results are presented and discussed in this paper.
Index Terms—Boost ac–dc converters, constant-frequency
power converters, equal-charge criterion control, microprocessor
control of converters.
I. INTRODUCTION
THE DRAWBACKS due to harmonic-rich currents drawn
by the conventional ac–dc converters have been well
documented in literature. Considerable research has been done
in the recent past to improve the quality of input-current
waveforms. Of the many schemes and topologies which have
been proposed to address the problem, single- and three-phase
boost-type ac–dc power converters are popular [1]–[11]. Fig. 1
shows a single-phase boost-type converter with bidirectional
power-flow capability. By switching the appropriate pair of de-
vices (1, 4) or (2, 3), the input-current waveform is controlled
to keep close to a sinusoidal template, which is derived from
the input-voltage waveform and whose phase and magnitude
can be set as desired.
Single-phase ac–dc converters [10]–[12] find application
in areas such as traction drives, low-power-rated induction
motor drives, low-power uninterruptible power supply (UPS)
systems, and battery chargers. The converter in Fig. 1 is of
particular interest in traction drives, where its bidirectional
power-transfer capability can be used to recover energy when
the motors are braked.
Manuscript received June 13, 1996; revised January 31, 1997. Recom-
mended by Associate Editor, R. Steigerwald.
R. Oruganti and L. K. Sang are with the Center for Power Electronics,
Department of Electrical Engineering, National University of Singapore,
119260, Singapore.
K. Nagaswamy is with Hewlett-Packard, Singapore.
Publisher Item Identifier S 0885-8993(98)00484-0.
Fig. 1. Single-phase boost-type ac–dc converter with current control.
Fig. 2. Reference and actual current waveforms in HCC.
Many methods for input-current waveshaping have been
reported [1], [3], [5], [6], [8], and [9]. The most common
method is the hysteresis current control (HCC) [1], [2], [7],
[10], and [11]. Here, the input current is kept to within a band
about the reference current wave. The waveform of the input
current and reference current along with the
hysteresis band for part of a line cycle are shown in Fig. 2.
This method has the advantages of simple implementation,
fast current dynamics, and inherent peak-current limiting capa-
bility. However, the scheme has a major drawback in that the
switching frequency varies over a wide range. As a result,
issues such as design of input filter and switching losses
of the semiconductor devices assume significance. This has
motivated research into constant switching frequency methods
[3]–[5], [9] applicable to boost-type converters.
References [3] and [4] discuss a method called predicted
current control with fixed switching frequency (PCFF) for
three-phase converters. Here, the duty cycle of the legs of
0885–8993/98$10.00 1998 IEEE

48 IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 13, NO. 1, JANUARY 1998
converter bridge is controlled such that the input current
reaches the value set by the reference current template at the
end of the switching cycle. The method has the advantages
of constant frequency besides having good dynamic charac-
teristics. However, the method does not force the average of
the input current to be sinusoidal. In [5], the HCC method
is modified to achieve a constant-frequency control for a
dc–ac three-phase inverter. A phase-locked loop (PLL) keeps
the converter’s switching frequency constant by adjusting the
hysteresis band. Here, too, fast dynamic response is possible
as with HCC. However, the PLL with a large low-pass filter
tends to create stability problems. Also, the PLL may lose
synchronization during transients [6], which could cause large
changes in the switching frequency. Reference [6] discusses an
adaptive HCC method, where the hysteresis band is controlled
adaptively to result in a nearly constant switching frequency
together with fast dynamics. The method may be viewed as
controlling the average value of input current indirectly in
contrast to the method discussed in the present paper, which
controls the average input current directly. Another constant-
frequency control method is outlined in [9]. Here, the input
current’s phase and magnitude are controlled by controlling
the fundamental component of the rectifier input voltage using
the sine pulse-width modulation method. A major drawback
of this method is that the response of any current loop to step
changes in load is slow. There is a dc component in the input
current soon after the change, which dies down after a few
line cycles.
In this paper, a control method, which has been named as the
predicted (on-time) equal-charge criterion (PECC) method, is
proposed. This method seeks to combine the superior dynamics
of the HCC method and the advantages offered by constant-
frequency switching. Analysis-based simulation is adopted to
verify the operation of the converter under this control method.
A simple constant-frequency equal-charge criterion (ECC)
method (Method A) is initially proposed, which is then found
to be inherently unstable. The method is then modified by
predicting the on time for the ECC (Method B). This method
results in stable operation in only one half of a line-cycle wave-
form. A combination of two ways of implementing Method B
after some fine tuning gives the PECC method, which is the
main contribution of this paper. Simulated performance results
like input-current ripple, variable power-factor operation, and
comparison with the HCC method are presented. Details of
microprocessor-based implementation and some experimental
results are also given verifying the anticipated good perfor-
mance of the scheme. Simulation results are compared with
those of the HCC method also.
II. ANALYSIS/SIMULATION OF CONVERTER
UNDER A CONTROL METHOD
In order to study the performance of the boost-type con-
verter under different control methods, analysis-based simula-
tion method was applied. The network equations which result
when device set (1, 4) is on and (2, 3) is off or vice versa were
first solved assuming the switches and diodes to be ideal. The
Fig. 3. Current waveforms in simple ECC (Method A).
solved equations were then used to simulate the performance
of the converter under a specific control method.
The input ac voltage and reference input current were
assumed to be constant during a switching interval. This
assumption simplifies the analysis and is justified since the
switching frequency (20 kHz) is much greater than the line
frequency (50 Hz). Besides, a stiff dc bus (250 V) at the output
was assumed, again to simplify the analysis, and is justified in
most applications. Other parameters of the converter include
input voltage of 110 V (rms), input inductance of 2.5 mH, and
a voltampere rating of 1 kVA.
III. SIMPLE ECC: METHOD A
The motivation of this method is to make the average current
in the inductor equal to the average of the reference value on
a switching-cycle-by-switching-cycle basis. In Fig. 3, device
set (2, 3) of the converter is turned on (instants and )
once every period , causing the current to rise. At ,
the device set is switched off and (1, 4) switched on when the
following condition is satisfied:
or (1)
shaded area.
Thus, Method A is a simple control method, which can be
implemented using opamps. It is to be noted here that the
interval over which the average of is made equal to
that of is not equal to the switching interval .Itis
the time between two successive turn ons of (2, 3), which is
equal to .
When the method was simulated using the SABER simula-
tion tool, unstable waveforms (Fig. 4) were observed. As may
be noticed, after a few switching cycles, the device set (2, 3)
was switched on (as per the method) even before the integral
could go to zero or negative. After this occurs, (1) is never
satisfied. The current increases to large values (Fig. 4).
Since the simulation clearly demonstrated the instability of
this method, no further investigations regarding the same are
included. To overcome this problem, modifications were made
to the simple ECC method, which will be discussed in the
subsequent sections.

ORUGANTI et al.: CRITERION SCHEME FOR CONTROL OF BOOST-TYPE AC–DC CONVERTERS 49
Fig. 4. Waveforms of and under Method A (SABER simu-
lation).
(a) (b)
Fig. 5. Reference and actual input-current waveforms in Method B. (a) Mode
Sequence I. (b) Mode Sequence II.
IV. DEVELOPMENT OF PECC METHOD
In this section, the basic concept of the PECC method
and the problems faced in realizing good performance will
be discussed. The final control scheme will be discussed in
Section V.
A. Initial Proposed PECC Method
In Method A (Fig. 3), the on time – is determined such
that the ECC will be satisfied over the interval – (which is
not the cycle period ). In the modification proposed (Method
B), the on time – is determined such that ECC will
be satisfied over the cycle period , – . Two ways of
implementation (Mode Sequences I and II) of this control
method are possible [see Fig. 5(a) and (b)]. The following
explanation is made with reference to Mode Sequence I
[Fig. 5(a)]. A similar explanation can be made for Mode
Sequence II by interchanging the roles of device sets (1, 4)
and (2, 3) (Fig. 1).
In Fig. 5(a), device set (2, 3) is switched on at the start
of the switching cycle of period , followed by (1, 4) at time
. As stated earlier, the control method predicts the value of
such that the ECC is satisfied at the end of the cycle. Thus
or (2)
The assumption of constant input ac voltage during the switch-
ing cycle (Section II) results in linear variations of as
shown.
With Mode Sequence I, in the positive line-current half
cycle, the inductance charges up (stores energy) first and
then discharges into the dc bus. However, in the negative half
cycle, with the same sequence, the inductor first discharges
as it drives current against the dc bus and then charges. In
Mode Sequence II, the sequence of inductor charging and
discharging during the positive and negative line-current half
cycles is reversed. By setting the shaded area equal to zero
(see Appendix A), a quadratic equation in is obtained for
each mode sequence. They are
(3)
where
and (4)
for Mode Sequence I and
and (5)
for Mode Sequence II, where and are the input ac
voltage and reference input current for the switching interval
and is the current at the start of the interval . The
control system must solve (3) in each switching cycle to
obtain , which determines the switching instant within the
period .
1) Instability in Method B and Modification: Upon simu-
lation, it was found that if a single mode sequence (I or II)
is used throughout the line cycle, the system is unstable for
half a line cycle. For example, as in Fig. 6, instability is
noted in the negative half cycle if Mode Sequence I is used
throughout. Similarly, the system is unstable in the positive
half cycle when Mode Sequence II is used throughout. In
the unstable condition, the input current strays away from
the reference. At one point, the current goes so far away
from the reference that ECC cannot be satisfied for that
switching cycle, and the appropriate devices have to be kept
on or off throughout the period. The current no longer keeps
in step with the reference, resulting in very high values of
peak-to-peak values ripple current (Fig. 6). In Section IV-A2
and Appendix B, an analysis of the observed instability is
presented.
2) Stability Analysis for Method B: Ignoring the slow-line
waveform variation, when there is a small perturbation in
from the steady-state value, it results in a disturbance
in , which, in turn, causes the current at the end of the
sample to be perturbed by a factor . The perturbations in
Mode Sequence I are shown in Fig 7. The initial perturbation
may be due to noise. In simulation, the rounding-off errors

50 IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 13, NO. 1, JANUARY 1998
Fig. 6. Input-current waveform with Mode Sequence I used in both the
line-voltage half cycles.
Fig. 7. Perturbations in and resultant perturbations in and in
Mode Sequence I.
appear as perturbations. The system will be unstable if the
magnitude of the ratio to , which shall be denoted ,
is greater than or equal to one. That is, a disturbance in
results in a larger disturbance in , which is the for the
subsequent sample.
Upon analyzing the system equations with these perturba-
tions (Appendix B), the ratio is given by
(6)
The magnitude of is greater than one if is greater
than or, in other words, the duty ratio is greater than
50%. Since and are nearly constant over a switching
cycle, the current rise and fall in each switching cycle may
be equated. The following equation is then obtained for Mode
Sequence I:
Duty ratio (7)
Thus, with Mode Sequence I, it can be seen that duty ratio
50 in the negative line-voltage half cycle and 50 in the
positive half cycle. This is the reason for the system tending
to be unstable in the negative half cycle if Mode Sequence
I is used.
The above instability is somewhat similar to that seen
in peak-current control in dc–dc converters. The operation
in these converters can be stabilized by slope compensation
[13], where a slope is incorporated into the reference. It
is possible that such an approach may be effective here
Fig. 8. Input-current waveform when Mode Sequences I and II are com-
bined.
Fig. 9. Spectrum of waveform in Fig. 8 (fundamental not shown).
also. But, it was not pursued here for the following rea-
sons. The input voltage varies over a wide range, so the
slope required for stable operation would also vary widely.
Such slope compensation would add to the complication in
processing, which is undesirable. Furthermore, since there is
always a mode sequence that is stable in any half cycle,
it is not necessary to go in for any external stabilization
technique.
B. Combination of Mode Sequences I and
II for Stable Operation
By using Mode Sequence I for the positive line-voltage half
cycle and Mode Sequence II for the negative half cycle, stable
operation over a complete cycle may be expected. The input
current for this combination is shown in Fig. 8. The figure
shows that there is a considerable amount of transients, seen
as a 10-kHz component in spectrum (Fig. 9), just after the
zero crossings. This phenomenon is explained as follows. As
the reference current crosses zero, operation changes from one
mode sequence to the other, introducing an abrupt change in
the sequence of the devices being switched. The current is far
different from the steady-state value of at this point required
under the second mode sequence for a smooth operation.
By making the current vary in the same direction as it had
before the switching cycle, the current is made to go even
farther from the reference. This part of the waveform is shown
in detail in Fig. 10. The transient dies down eventually due
to the inherent stability of the mode sequence, but being
appreciable, it is observable on the spectrum. The transient
could be minimized by making the transition from one mode
sequence to the other after the current is close to a value of
, which will result in smooth operation. This is discussed in
the next section.

ORUGANTI et al.: CRITERION SCHEME FOR CONTROL OF BOOST-TYPE AC–DC CONVERTERS 51
Fig. 10. Input-current waveform during the zero crossing.
Fig. 11. Input-current waveform during transition from Mode Sequence I to
II under the PECC method.
V. PROPOSED PREDICTED ON-TIME ECC METHOD
The on-time value is used as the criterion for deciding to
changeover. When it goes beyond a threshold value (say, 55%
of ), mode sequence is changed as follows. Fig. 11 shows
a transition from Mode Sequence I to Mode Sequence II.
Mode Sequence I is used until , with Mode Sequence
II after .At , device set (2, 3) is turned on. It is
kept on for time interval – , which is equal to
calculated for Mode Sequence I -.At , the
input current also changes to a value that is close to
a value, which gives smooth operation in Mode Sequence II.
Then, at -is calculated based on Mode
Sequence II. Device set (1, 4) is turned on for a duration of
-. A similar changeover is carried out when operation
is to change over from Mode Sequence II to Mode Sequence
I. The resultant waveform is shown in Fig. 12 and the spec-
trum in Fig. 13. Essentially, during the transition, the switch-
ing period is constant until and after . There is
a transition interval – , which is not equal to the
switching interval. Note, however, that - - will
be nearly equal to a period . Thus, the operation is not
constant frequency for a very small part of the line cycle. The
current waveform is smooth, and the spectrum shows only the
sidebands due to the switching frequency, as is the case with
constant switching frequency. Stable operation is achieved,
minus the zero-crossing transients. This final method has been
named the PECC method.
Fig. 12. Input-current waveform under the PECC method (1-kVA output).
Fig. 13. Spectrum of waveform in Fig. 12 (fundamental not shown).
Fig. 14. Waveform of the ripple in current in Fig. 12.
VI. SIMULATION RESULTS
A. Input-Current Ripple
The ripple in the input current using the PECC method
is shown in Fig. 14. The ripple was calculated by simply
subtracting from . The magnitude of ripple de-
pends upon the switching frequency and boost inductance
alone and is independent of the input-current amplitude. A
small jump in ripple may be observed at the points where
the mode sequence transitions occur near 0.01 s. This may
be attributed to the fact that the current at the start of the
transitional period is not exactly the same as the steady-state
value for the later sequence. Fig. 12 shows the input-current
waveform at a power rating of 1 kVA and Fig. 15 shows the
input current at a light load of about 100 VA. It shows that
at light loads, the ripple swamps the fundamental component
as would be expected.
B. Variable Power Factor and Regenerative Operation
In the derivation of (3)–(5), no assumption was made regard-
ing the ac voltage or current waveforms. In other words, the
phase difference between the current and voltage waveforms
need not be fixed. Thus, using a boost-type converter with