mada

IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 13, NO. 1, JANUARY 1998
47
Predicted (On-Time) Equal-Charge Criterion Scheme
for Constant-Frequency Control of Single-Phase
Boost-Type AC–DC Converters
Ramesh Oruganti, Member, IEEE, Kannan Nagaswamy, and Lock Kai Sang
Abstract—A new constant switching frequency control method
for single-phase boost-type ac–dc converters is presented. The
on times of the converter switches in each switching period is
determined such that the average input current tracks the reference template in every switching cycle. The problems encountered
in achieving smooth and stable operation and the modifications
made to overcome them are discussed. The simulation studies
done on the converter controlled with this method, which is given
the name predicted (on-time) equal-charge criterion (PECC)
method, indicate stable operation at different input-current and
voltage levels and power factors. The method was implemented
on an insulated gate bipolar transistor (IGBT) converter rated
for 1 kVA using a 80 386 processor system for computations. The
experimental results are presented and discussed in this paper.
Index Terms—Boost ac–dc converters, constant-frequency
power converters, equal-charge criterion control, microprocessor
control of converters.
Fig. 1. Single-phase boost-type ac–dc converter with current control.
I. INTRODUCTION
T
HE DRAWBACKS due to harmonic-rich currents drawn
by the conventional ac–dc converters have been well
documented in literature. Considerable research has been done
in the recent past to improve the quality of input-current
waveforms. Of the many schemes and topologies which have
been proposed to address the problem, single- and three-phase
boost-type ac–dc power converters are popular [1]–[11]. Fig. 1
shows a single-phase boost-type converter with bidirectional
power-flow capability. By switching the appropriate pair of devices (1, 4) or (2, 3), the input-current waveform is controlled
to keep close to a sinusoidal template, which is derived from
the input-voltage waveform and whose phase and magnitude
can be set as desired.
Single-phase ac–dc converters [10]–[12] find application
in areas such as traction drives, low-power-rated induction
motor drives, low-power uninterruptible power supply (UPS)
systems, and battery chargers. The converter in Fig. 1 is of
particular interest in traction drives, where its bidirectional
power-transfer capability can be used to recover energy when
the motors are braked.
Manuscript received June 13, 1996; revised January 31, 1997. Recommended by Associate Editor, R. Steigerwald.
R. Oruganti and L. K. Sang are with the Center for Power Electronics,
Department of Electrical Engineering, National University of Singapore,
119260, Singapore.
K. Nagaswamy is with Hewlett-Packard, Singapore.
Publisher Item Identifier S 0885-8993(98)00484-0.
Fig. 2. Reference and actual current waveforms in HCC.
Many methods for input-current waveshaping have been
reported [1], [3], [5], [6], [8], and [9]. The most common
method is the hysteresis current control (HCC) [1], [2], [7],
[10], and [11]. Here, the input current is kept to within a band
about the reference current wave. The waveform of the input
current
and reference current
along with the
hysteresis band for part of a line cycle are shown in Fig. 2.
This method has the advantages of simple implementation,
fast current dynamics, and inherent peak-current limiting capability. However, the scheme has a major drawback in that the
switching frequency varies over a wide range. As a result,
issues such as design of input filter and switching losses
of the semiconductor devices assume significance. This has
motivated research into constant switching frequency methods
[3]–[5], [9] applicable to boost-type converters.
References [3] and [4] discuss a method called predicted
current control with fixed switching frequency (PCFF) for
three-phase converters. Here, the duty cycle of the legs of
0885–8993/98$10.00  1998 IEEE
48
IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 13, NO. 1, JANUARY 1998
converter bridge is controlled such that the input current
reaches the value set by the reference current template at the
end of the switching cycle. The method has the advantages
of constant frequency besides having good dynamic characteristics. However, the method does not force the average of
the input current to be sinusoidal. In [5], the HCC method
is modified to achieve a constant-frequency control for a
dc–ac three-phase inverter. A phase-locked loop (PLL) keeps
the converter’s switching frequency constant by adjusting the
hysteresis band. Here, too, fast dynamic response is possible
as with HCC. However, the PLL with a large low-pass filter
tends to create stability problems. Also, the PLL may lose
synchronization during transients [6], which could cause large
changes in the switching frequency. Reference [6] discusses an
adaptive HCC method, where the hysteresis band is controlled
adaptively to result in a nearly constant switching frequency
together with fast dynamics. The method may be viewed as
controlling the average value of input current indirectly in
contrast to the method discussed in the present paper, which
controls the average input current directly. Another constantfrequency control method is outlined in [9]. Here, the input
current’s phase and magnitude are controlled by controlling
the fundamental component of the rectifier input voltage using
the sine pulse-width modulation method. A major drawback
of this method is that the response of any current loop to step
changes in load is slow. There is a dc component in the input
current soon after the change, which dies down after a few
line cycles.
In this paper, a control method, which has been named as the
predicted (on-time) equal-charge criterion (PECC) method, is
proposed. This method seeks to combine the superior dynamics
of the HCC method and the advantages offered by constantfrequency switching. Analysis-based simulation is adopted to
verify the operation of the converter under this control method.
A simple constant-frequency equal-charge criterion (ECC)
method (Method A) is initially proposed, which is then found
to be inherently unstable. The method is then modified by
predicting the on time for the ECC (Method B). This method
results in stable operation in only one half of a line-cycle waveform. A combination of two ways of implementing Method B
after some fine tuning gives the PECC method, which is the
main contribution of this paper. Simulated performance results
like input-current ripple, variable power-factor operation, and
comparison with the HCC method are presented. Details of
microprocessor-based implementation and some experimental
results are also given verifying the anticipated good performance of the scheme. Simulation results are compared with
those of the HCC method also.
II. ANALYSIS/SIMULATION OF CONVERTER
UNDER A CONTROL METHOD
In order to study the performance of the boost-type converter under different control methods, analysis-based simulation method was applied. The network equations which result
when device set (1, 4) is on and (2, 3) is off or vice versa were
first solved assuming the switches and diodes to be ideal. The
Fig. 3. Current waveforms in simple ECC (Method A).
solved equations were then used to simulate the performance
of the converter under a specific control method.
The input ac voltage and reference input current were
assumed to be constant during a switching interval. This
assumption simplifies the analysis and is justified since the
switching frequency (20 kHz) is much greater than the line
frequency (50 Hz). Besides, a stiff dc bus (250 V) at the output
was assumed, again to simplify the analysis, and is justified in
most applications. Other parameters of the converter include
input voltage of 110 V (rms), input inductance of 2.5 mH, and
a voltampere rating of 1 kVA.
III. SIMPLE ECC: METHOD A
The motivation of this method is to make the average current
in the inductor equal to the average of the reference value on
a switching-cycle-by-switching-cycle basis. In Fig. 3, device
set (2, 3) of the converter is turned on (instants
and )
once every period , causing the current to rise. At
,
the device set is switched off and (1, 4) switched on when the
following condition is satisfied:
or
(1)
shaded area.
Thus, Method A is a simple control method, which can be
implemented using opamps. It is to be noted here that the
interval over which the average of
is made equal to
that of
is not equal to the switching interval . It is
the time between two successive turn ons of (2, 3), which is
equal to .
When the method was simulated using the SABER simulation tool, unstable waveforms (Fig. 4) were observed. As may
be noticed, after a few switching cycles, the device set (2, 3)
was switched on (as per the method) even before the integral
could go to zero or negative. After this occurs, (1) is never
satisfied. The current increases to large values (Fig. 4).
Since the simulation clearly demonstrated the instability of
this method, no further investigations regarding the same are
included. To overcome this problem, modifications were made
to the simple ECC method, which will be discussed in the
subsequent sections.
ORUGANTI et al.: CRITERION SCHEME FOR CONTROL OF BOOST-TYPE AC–DC CONVERTERS
Fig. 4. Waveforms of iref (t) and iact (t) under Method A (SABER simulation).
49
The assumption of constant input ac voltage during the switching cycle (Section II) results in linear variations of
as
shown.
With Mode Sequence I, in the positive line-current half
cycle, the inductance
charges up (stores energy) first and
then discharges into the dc bus. However, in the negative half
cycle, with the same sequence, the inductor first discharges
as it drives current against the dc bus and then charges. In
Mode Sequence II, the sequence of inductor charging and
discharging during the positive and negative line-current half
cycles is reversed. By setting the shaded area equal to zero
(see Appendix A), a quadratic equation in
is obtained for
each mode sequence. They are
(3)
where
and
(4)
for Mode Sequence I and
(a)
(b)
Fig. 5. Reference and actual input-current waveforms in Method B. (a) Mode
Sequence I. (b) Mode Sequence II.
IV. DEVELOPMENT
OF
PECC METHOD
In this section, the basic concept of the PECC method
and the problems faced in realizing good performance will
be discussed. The final control scheme will be discussed in
Section V.
A. Initial Proposed PECC Method
In Method A (Fig. 3), the on time – is determined such
that the ECC will be satisfied over the interval – (which is
not the cycle period ). In the modification proposed (Method
–
is determined such that ECC will
B), the on time
be satisfied over the cycle period ,
– . Two ways of
implementation (Mode Sequences I and II) of this control
method are possible [see Fig. 5(a) and (b)]. The following
explanation is made with reference to Mode Sequence I
[Fig. 5(a)]. A similar explanation can be made for Mode
Sequence II by interchanging the roles of device sets (1, 4)
and (2, 3) (Fig. 1).
In Fig. 5(a), device set (2, 3) is switched on at the start
of the switching cycle of period , followed by (1, 4) at time
. As stated earlier, the control method predicts the value of
such that the ECC is satisfied at the end of the cycle. Thus
or
(2)
and
(5)
for Mode Sequence II, where
and
are the input ac
voltage and reference input current for the switching interval
and
is the current at the start of the interval
. The
control system must solve (3) in each switching cycle to
obtain
, which determines the switching instant within the
period .
1) Instability in Method B and Modification: Upon simulation, it was found that if a single mode sequence (I or II)
is used throughout the line cycle, the system is unstable for
half a line cycle. For example, as in Fig. 6, instability is
noted in the negative half cycle if Mode Sequence I is used
throughout. Similarly, the system is unstable in the positive
half cycle when Mode Sequence II is used throughout. In
the unstable condition, the input current strays away from
the reference. At one point, the current
goes so far away
from the reference that ECC cannot be satisfied for that
switching cycle, and the appropriate devices have to be kept
on or off throughout the period. The current no longer keeps
in step with the reference, resulting in very high values of
peak-to-peak values ripple current (Fig. 6). In Section IV-A2
and Appendix B, an analysis of the observed instability is
presented.
2) Stability Analysis for Method B: Ignoring the slow-line
waveform variation, when there is a small perturbation
in
from the steady-state value, it results in a disturbance
in
, which, in turn, causes the current at the end of the
. The perturbations in
sample to be perturbed by a factor
Mode Sequence I are shown in Fig 7. The initial perturbation
may be due to noise. In simulation, the rounding-off errors
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IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 13, NO. 1, JANUARY 1998
Fig. 6. Input-current waveform with Mode Sequence I used in both the
line-voltage half cycles.
Fig. 8. Input-current waveform when Mode Sequences I and II are combined.
Fig. 9. Spectrum of waveform in Fig. 8 (fundamental not shown).
Fig. 7. Perturbations in ix and resultant perturbations in
Mode Sequence I.
TON
and iz in
appear as perturbations. The system will be unstable if the
magnitude of the ratio
to
, which shall be denoted ,
is greater than or equal to one. That is, a disturbance in
results in a larger disturbance in , which is the
for the
subsequent sample.
Upon analyzing the system equations with these perturbations (Appendix B), the ratio is given by
(6)
is greater than one if
is greater
The magnitude of
than
or, in other words, the duty ratio is greater than
50%. Since
and
are nearly constant over a switching
cycle, the current rise and fall in each switching cycle may
be equated. The following equation is then obtained for Mode
Sequence I:
Duty ratio
(7)
Thus, with Mode Sequence I, it can be seen that duty ratio
50 in the negative line-voltage half cycle and 50 in the
positive half cycle. This is the reason for the system tending
to be unstable in the negative half cycle if Mode Sequence
I is used.
The above instability is somewhat similar to that seen
in peak-current control in dc–dc converters. The operation
in these converters can be stabilized by slope compensation
[13], where a slope is incorporated into the reference. It
is possible that such an approach may be effective here
also. But, it was not pursued here for the following reasons. The input voltage varies over a wide range, so the
slope required for stable operation would also vary widely.
Such slope compensation would add to the complication in
processing, which is undesirable. Furthermore, since there is
always a mode sequence that is stable in any half cycle,
it is not necessary to go in for any external stabilization
technique.
B. Combination of Mode Sequences I and
II for Stable Operation
By using Mode Sequence I for the positive line-voltage half
cycle and Mode Sequence II for the negative half cycle, stable
operation over a complete cycle may be expected. The input
current for this combination is shown in Fig. 8. The figure
shows that there is a considerable amount of transients, seen
as a 10-kHz component in spectrum (Fig. 9), just after the
zero crossings. This phenomenon is explained as follows. As
the reference current crosses zero, operation changes from one
mode sequence to the other, introducing an abrupt change in
the sequence of the devices being switched. The current is far
different from the steady-state value of at this point required
under the second mode sequence for a smooth operation.
By making the current vary in the same direction as it had
before the switching cycle, the current is made to go even
farther from the reference. This part of the waveform is shown
in detail in Fig. 10. The transient dies down eventually due
to the inherent stability of the mode sequence, but being
appreciable, it is observable on the spectrum. The transient
could be minimized by making the transition from one mode
sequence to the other after the current is close to a value of
, which will result in smooth operation. This is discussed in
the next section.
ORUGANTI et al.: CRITERION SCHEME FOR CONTROL OF BOOST-TYPE AC–DC CONVERTERS
51
Fig. 12. Input-current waveform under the PECC method (1-kVA output).
Fig. 10.
Input-current waveform during the zero crossing.
Fig. 13. Spectrum of waveform in Fig. 12 (fundamental not shown).
Fig. 11. Input-current waveform during transition from Mode Sequence I to
II under the PECC method.
V. PROPOSED PREDICTED ON-TIME ECC METHOD
The on-time value is used as the criterion for deciding to
changeover. When it goes beyond a threshold value (say, 55%
of ), mode sequence is changed as follows. Fig. 11 shows
a transition from Mode Sequence I to Mode Sequence II.
Mode Sequence I is used until
, with Mode Sequence
II after
. At
, device set (2, 3) is turned on. It is
kept on for time interval
–
, which is equal to
calculated for Mode Sequence I
, the
- . At
also changes to a value that is close to
input current
a value, which gives smooth operation in Mode Sequence II.
Then, at
- is calculated based on Mode
Sequence II. Device set (1, 4) is turned on for a duration of
- . A similar changeover is carried out when operation
is to change over from Mode Sequence II to Mode Sequence
I.
The resultant waveform is shown in Fig. 12 and the spectrum in Fig. 13. Essentially, during the transition, the switching period is constant until
and after
. There is
a transition interval
–
, which is not equal to the
switching interval. Note, however, that
- will
be nearly equal to a period . Thus, the operation is not
constant frequency for a very small part of the line cycle. The
current waveform is smooth, and the spectrum shows only the
sidebands due to the switching frequency, as is the case with
constant switching frequency. Stable operation is achieved,
minus the zero-crossing transients. This final method has been
named the PECC method.
Fig. 14. Waveform of the ripple in current in Fig. 12.
VI. SIMULATION RESULTS
A. Input-Current Ripple
The ripple in the input current using the PECC method
is shown in Fig. 14. The ripple was calculated by simply
subtracting
from
. The magnitude of ripple depends upon the switching frequency and boost inductance
alone and is independent of the input-current amplitude. A
small jump in ripple may be observed at the points where
the mode sequence transitions occur near 0.01 s. This may
be attributed to the fact that the current at the start of the
transitional period is not exactly the same as the steady-state
value for the later sequence. Fig. 12 shows the input-current
waveform at a power rating of 1 kVA and Fig. 15 shows the
input current at a light load of about 100 VA. It shows that
at light loads, the ripple swamps the fundamental component
as would be expected.
B. Variable Power Factor and Regenerative Operation
In the derivation of (3)–(5), no assumption was made regarding the ac voltage or current waveforms. In other words, the
phase difference between the current and voltage waveforms
need not be fixed. Thus, using a boost-type converter with
52
IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 13, NO. 1, JANUARY 1998
TABLE I
COMPARISON OF HCC AND PECC SCHEMES AT 110 V,
9.09-A INPUT, 250 Vdc, AND 1-kW OUTPUT
Fig. 15.
Input-current waveform at 100-VA output.
TABLE II
COMPARISON OF HCC AND PECC SCHEMES AT 70 V,
9.09-A INPUT, 200 Vdc, AND 0.636-kW OUTPUT
Fig. 16. Input-current and voltage waveforms at zero power factor (capacitive load and = 90 ).
Fig. 18.
I).
Fig. 17.
iact (t) and Iref
for successive switching intervals (Mode Sequence
Input-current and voltage waveforms at regenerative operation
( = 180 ).
C. Comparison with HCC Method
the proposed control, operation at arbitrary power factors as
well as regeneration is possible. For this general case, the
input-current reference is given by
(8)
where
is the peak value of input-current reference, is the
fundamental angular frequency, and is the current phase shift
with respect to input voltage. A value between 90 and 270
indicates regenerative operation. For the PECC method, operation under different values of power factor were simulated and
waveforms studied. The waveforms for 90 (leading) phase
shift (capacitive load) and regenerative operation
are shown in Figs. 16 and 17, respectively. It can be observed
that the operation is stable and transient-free for these cases
also. Also, the behavior of the switching ripple is different
for different power factors. At unity power factor, the peakto-peak value of ripple is maximum at the zero crossings of
the current waveform and is minimum at the peaks. At zero
power factor, the ripple value is maximum at the peaks.
A brief comparison is made between the PECC and HCC
methods. Both were simulated for the same converter under
identical input and output conditions (see Section II).
The comparative results under full-load conditions are given
in Table I. In the HCC method, for a certain hysteresis band,
the switching frequency varies over a wide range, by a factor
of nearly two. The PECC operates at constant frequency,
making input filtering relatively easier, which is its significant
advantage. It should also be noted that with HCC, the range
of switching frequencies varies widely at different operating
conditions for a given hysteresis band. Furthermore, it has
been noted from simulation studies that when the input and
output voltages are lower, then, for a given input current,
PECC results in less ripple. This is another advantage of the
PECC method over the HCC method (see Table II).
On the negative side, the PECC method is more complex.
Also, the ripple current is higher at certain parts of the
waveform, as in all constant-frequency schemes. This could
result in slightly higher device stresses. More importantly,
this could also result in higher conducted EMI (electromagnetic interference). However, since the switching frequency is
constant, the filtering of this ripple would be simple.
ORUGANTI et al.: CRITERION SCHEME FOR CONTROL OF BOOST-TYPE AC–DC CONVERTERS
Fig. 19.
53
Block diagram of the 80 386 system used for implementing the PECC method.
D. Comparison with Average Current Control
The PECC method involves the control of the average
of input current in each switching interval. In the average
current control typically used in the single-phase power-factorcorrection systems [13], the average of the input current, after
filtering out the switching frequency components, is made to
follow the reference. The filtering results in poor currentloop dynamics. However, in the PECC method, the average
current is controlled in each switching cycle. Hence, the PECC
method is expected to exhibit good dynamic performance at
the expense of increased complexity. It has also been verified
experimentally that the dynamic performance of the PECC
method is very good, with the system responding very fast to
step changes in input-current reference.
VII. PRACTICAL IMPLEMENTATION
The implementation of the PECC method requires the
solution of a quadratic (3) based on the parameters such as
and . In the present work, it is carried out using
the Intel 80 386 microprocessor. This processor was chosen
for the 32-b data bus and the reasonably fast execution of
instructions.
The parameters
and
are acquired from the
actual quantities. Their acquisition and further computation
cannot be performed instantaneously. Hence, in each switching
period, the value of
for the subsequent cycle is calculated.
The following explanation is for Mode Sequence I (refer to
Fig. 18).
is calculated in the switching period prior to Period A.
In Period A,
is sensed using an analog–digital converter
(ADC) as well as
and
. With these quantities, the
value of
is estimated from (A-7).
and
are
is calculated during
also estimated. Using these values,
Period A itself. The value of
is fed to a logic circuit,
which converts this quantity to the gate-drive signals to
appropriate devices of the converter. Fig. 19 shows the block
diagram of the hardware.
Slight changes in the line frequency are accounted for by
sensing the zero crossings of the line voltage [zero-crossing
detector]. The line frequency is worked out from the squarewave output of the zero-crossing detector. The accurate value
of line frequency is used in generation of the reference current
value
.
Fig. 20. Flowchart for computation of
interval.
TON
for the subsequent switching
The flowchart in Fig. 20 shows how
is computed in
interval A. The method is the same throughout the line cycle
irrespective of the mode sequence. Only the equations used
for estimating
and
are different for different mode
sequences. In the case of a transition interval between mode
sequences,
is computed as per (A-3) using the existing
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IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 13, NO. 1, JANUARY 1998
(a)
(b)
Fig. 21. Input-voltage and current waveforms (scale: voltage—60 V/div; current—14 A/div). (a) Input current (rms) = 10:03 A. (b) Input current
(rms) = 5:38 A.
mode sequence and the computation of
for the subsequent
interval using the next mode sequence.
is shown to be
In the flowchart, processing for
interleaved with the analog–digital conversions. This is done
to save time while waiting for a certain quantity to be digitized.
However, it is not always possible, since it will be necessary
to wait for the digitized value of
in order to proceed
further in the computations. Thus, there is an overhead in
the processing in the form of ADC time. The ADC that was
used for the present work was slow (digitization time 15
s), hence, working with a switching interval of 50 s (for
20-kHz switching) as originally anticipated was not possible.
Therefore, a switching interval of 100 s (10-kHz switching)
was decided upon. It should be noted that this is not due
to a limitation of the method or the implementation. Having
decided upon 10-kHz switching, the boost inductance was
changed to 5 mH instead of 2.5 mH, which was used for
simulation studies. All other specifications of the converter
remain the same as in Section II. The switching frequency
can be raised by using faster ADC’s and processors with
faster clocks and processing speed. Due to limitations of space,
further details of implementation are not provided.
Fig. 22. Input-voltage and current waveforms [Vdc = 91:7 V and input
current (rms) = 7 A] (scale: voltage—30 V/div; current—14 A/div).
VIII. EXPERIMENTAL RESULTS
The PECC method was implemented on an insulated gate
bipolar transistor (IGBT) converter (using IRGTI050U06)
based on the 80 386 system explained in Section VII. The
converter was run at different power levels, input and output
voltages, and power factors. All claims made from the simulation studies were verified experimentally. This section will
describe the results.
A. Zero Phase Shift
Fig. 21 shows the input-voltage and current waveforms for
zero-phase-shift (unity power factor) operation, with
V(rms),
V, and two different input-current levels. Current waveforms were obtained from the output of the
circuit sensing
, which is a voltage signal proportional
.
to
The system was run at different input and output voltages
apart from the ones specified in Section II. It was seen to
Fig. 23. Spectrum of input-current waveform in Fig. 22 (without fundamental).
be satisfactory for different conditions. Just one condition
is shown here. Fig. 22 shows the waveforms for
V(rms).
The spectrum of the ripple component of input current was
examined. The fundamental component of the input current
was removed using a high-pass filter, giving a voltage signal
proportional to the ripple current. The spectrum of the inputcurrent waveform in Fig. 22 is given in Fig. 23.
The spectrum in Fig. 23 indicates components centered
around integral multiples of the switching frequency of 10
kHz. (The amplitude of the 10-kHz component is about 4%
of the fundamental.) This clearly indicates constant-frequency
switching. It may be noted from Fig. 23 that there is a
very small component around 5 kHz. This is due to slight
instability in the current waveform just after the zero crossings
ORUGANTI et al.: CRITERION SCHEME FOR CONTROL OF BOOST-TYPE AC–DC CONVERTERS
(a)
55
(b)
Fig. 24.
Input-current waveform with 60 phase shift. (a) Current leading. (b) Current lagging.
Fig. 25.
Response to a step change in input-current reference between 2–6.6 A (rms). (a) Step rise. (b) Step fall.
(a)
before switching over to the stable mode of operation. The
changeover between mode sequences is done once the value
of
goes beyond a threshold value, which is ideally 50%
of . This threshold value was decided based on the software
execution time. This value has to be at least equal to the
maximum execution time (including the ADC time). Due to
this consideration, the threshold was made 65 s. As a result
of this, the instability after zero crossings is allowed to persist.
This instability shows up as the component around 5 kHz. This
can be minimized by limiting this threshold to close to 50%
of . This is possible by using faster ADC’s and optimizing
the program for faster operation.
B. Nonzero Phase Shift
The system was run for nonunity power-factor operation
also. The software needed little change to accommodate the
nonzero phase shifts. Fig. 24 shows the input-current and
voltage waveforms with a 60 phase shift at input voltage
of 110 V(rms) and current of 7.7 A(rms).
Regenerative operation was not tried out due to nonavailability of active load that can sustain
at the dc link. Still,
nonunity power-factor operation demonstrates the feasibility
of bidirectional power flow using the PECC method.
C. Transient Response
Fig. 25 shows the response of the converter with the PECC
method for a step change in input current. Current is made to
(b)
rise and fall between 2–6.6 A(rms). The response is seen to
be extremely fast. It can be clearly seen from the plots that
the current is made to change to the new value of reference
in the fastest possible way.
IX. CONCLUSION
A control method for a single-phase boost-type ac–dc converter was developed and evaluated through simulation. Experimental results proved the workability of the method.
The method was shown to have the advantages of constant
switching frequency with fast dynamic response. The main
idea behind the method is to determine the duty cycle of
operation of the switches in a given switching period such that
the input average current tracks the input-current waveform
template on a cycle-by-cycle basis. The paper discussed how
the problems encountered in achieving stable and smooth
operation are overcome by incorporating changes in the control
method. The final method arrived at is named the PECC
method.
Both the PECC and conventional HCC methods have been
compared. The PECC method was also simulated and experimentally verified for arbitrary power factors and found to give
stable and smooth operation. The control implementation is
somewhat complex, necessitating the use of microprocessors
as it involves calculations for on time in each switching
interval. Nevertheless, the scheme offers several advantages
56
IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 13, NO. 1, JANUARY 1998
like simplified input filter design, fast dynamic response,
and stable operation under different power factors and inputcurrent levels. The switching frequency of 10 kHz can be
increased appreciably by using faster ADC’s and higher end
processors that are currently available.
APPENDIX B
DERIVATION OF (6)
and
is obtained.
First, the relation between
The variables
and
are substituted by the perturbed
values
and
in (3) and (4) to give
APPENDIX A
DERIVATION OF (3)–(5)
are derived. Then, from these
First, the equations for
equations, the areas under
and
are found and
equated to get the condition to satisfy ECC.
Refer to Fig. 5(a). During the time – , device set (2, 3)
is on. Then, the equation for voltage across the inductor is
(B-1)
The higher order perturbation terms can be neglected. Expanding the terms in (B-1) and substituting (A-14) in (B-1)
(B-2)
(A-1)
Perturbing the variables in (A-7)
Simplifying and integrating
(A-2)
In (A-2),
at
is the current
(denoted as
at
. The value of
) from (A-2) is
(A-3)
, device set (1, 4) is switched on. The voltage across
At
the inductor when (1, 4) is on is
(A-4)
Simplifying and integrating
(A-5)
at
The value of
(denoted as
) from (A-5) is
(A-6)
Substituting for
from (A-3) in (A-6) and simplifying
(A-7)
For ECC to be satisfied [see Fig. 5(a)]
Area ADEB
Area BEFC
Area AGHC
(A-8)
ADEB
(A-9)
BEFC
(A-10)
AGHC
(A-11)
Substituting (A-3), (A-7), and (A-9)–(A-11) in (A-8), (3) and
(4) are obtained.
Following the same approach for Mode Sequence II in
Fig. 5(b), the other condition for ECC can be worked out.
(B-3)
Substituting (A-7) in (B-3) and simplifying
(B-4)
Equation (6) is obtained by substituting (B-2) in (B-4) and
simplifying.
REFERENCES
[1] B. T. Ooi, J. C. Salmon, J. W. Dixon, and A. B. Kulkarni, “A 3-phase
controlled current PWM converter with leading power factor,” in Conf.
Rec. IEEE-IAS, 1985.
[2] J. W. Dixon, A. B. Kulkarni, M. Nishimoto, and B. T. Ooi, “Characteristics of a controlled current PWM rectifier-inverter link,” in Conf.
Rec. IEEE-PESC, 1986.
[3] R. Wu, S. B. Dewan, and G. R. Slemon, “A PWM AC to DC converter
with fixed switching frequency,” IEEE Trans. Ind. Applicat., vol. 26,
no. 5, 1990.
, “Analysis of a PWM ac–dc voltage source converter under the
[4]
predicted current control with fixed switching frequency,” IEEE Trans.
Ind. Applicat., vol. 27, no. 4, 1991.
[5] L. Malesani and P. Tenti, “A novel Hysteresis control method for
current-controlled VSI PWM inverters with constant modulation frequency,” in Proc. Conf. Rec. IEEE/IAS Annu. Meet., 1987.
[6] B. K. Bose, “An adaptive hysteresis-band current control technique of a
voltage-fed PWM inverter for machine drive system,” IEEE Trans. Ind.
Electron., vol. 37, no. 5, 1990.
[7] A. W. Green and J. T. Boys, “Hysteresis current-forced three phase
voltage sourced reversible rectifier,” Proc. Inst. Elect. Eng., vol. 136,
pt. B, no. 3, 1989.
[8] T. G. Habetler, “A space vector based rectifier regulator for AC/DC/AC
converters,” IEEE Trans. Power Electron., vol. 8, no. 1, 1993.
[9] J. W. Dixon and B. T. Ooi, “Indirect current control of a unity power
factor sinusoidal current boost type three phase rectifier,” IEEE Trans.
Ind. Electron., vol. 35, no. 4, 1988.
[10] O. Stihi and B. T. Ooi, “A single phase controlled-current PWM
rectifier,” IEEE Trans. Power Electron., vol. 3, no. 4, 1988.
[11] J. T. Boys and A. W. Green, “Current-forced single phase reversible
rectifier,” Proc. Inst. Elect. Eng., vol. 136, pt. B, no. 5, 1989.
[12] K. Thiyagarajah, V. T. Ranganathan, and B. S. R. Iyengar, “A high
switching frequency IGBT PWM rectifier/inverter system for AC motor drives operating from single phase supply,” IEEE Trans. Power
Electron., vol. 6, no. 4, 1991.
[13] Unitrode Power Supply Design Seminar Manual, Unitrode Corp., Lexington, MA, 1991.
[14] R. Oruganti, K. Nagaswamy, and K. S. Lock, “A constant frequency
variable power factor PWM scheme for single phase boost type ac–dc
converter,” in IEEE Int. Conf. Energy Management and Power Delivery,
1995.
ORUGANTI et al.: CRITERION SCHEME FOR CONTROL OF BOOST-TYPE AC–DC CONVERTERS
Ramesh Oruganti (S’83–M’85) received the
B.Tech. and M.Tech. degrees from the Indian
Institute of Technology, Madras, India. In 1987, he
received the Ph.D. degree from Virginia Polytechnic
Institute and State University, Blacksburg.
He worked for two years at the Corporate
R&D Division of the General Electric Company
on advanced power converter systems. Since
1989, he has been a Senior Lecturer in the
Electrical Engineering Department, National
University of Singapore. His research interests
in power electronics include soft-switched converters, active power-factor
improvement, and converter modeling and control. He is currently the Director
of the Center for Power Electronics in the Faculty of Engineering, National
University of Singapore.
Dr. Oruganti is a Member of the honor societies of Phi Kappa Phi and
Eta Kappa Nu.
Kannan Nagaswamy received the Master of Engineering degree in electrical engineering from the
Indian Institute of Science, Bangalore, India, in
1991.
He worked as a Senior Engineer (R&D) at the
Industrial Systems Group, National Radio and Electronics Co. Ltd., from August 1991 to August 1994,
where he was involved in the design, development,
and testing of microprocessor-based induction motor
drives. From September 1994 to September 1996,
he was a Research Scholar in the Department of
Electrical Engineering, National University of Singapore, Singapore, pursuing
research in single-phase boost-type ac–dc converters. He is now with HewlettPackard, Singapore, where he provides electrical engineering and systems
support for the assembly lines making inkjet cartridges.
57
Lock Kai Sang received the B.Sc. and Ph.D. degrees from the University of Strathclyde, U.K., in
1975 and 1979, respectively.
He was the Head of the Power and Machines
Division in the Department of Electrical Engineering, National University of Singapore, Singapore.
He is currently with the Center for Power Electronics, Department of Electrical Engineering, National
University of Singapore. His areas of specialization
are design and analysis of small electric machines,
control of electric drives, and power system harmonics. He has undertaken a number of industrial consultancy projects related to
these areas.
Dr. Sang is a Member of the Institution of Electrical Engineers, U.K., and
a registered Professional Engineer in Singapore.