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# Scattering and Transfer Matrix of Screening Effectiveness - modeling of the triaxial setup

publicité ```The scattering and transfer matrix of screening effectiveness,
modeling of the triaxial setup
Thomas H&auml;hner
Nexans
Draveil, France
+33 6 80 12 55 04, [email protected]
Abstract
The method described in the literature to model the triaxial setup to
measure the screening attenuation in the case of different coupling
sections is based on the principle of (voltage) wave propagation and
(multiple) reflections. This approach leads to cumbersome
formulae. The following article describes a more versatile way
based on the theory of cascaded multi-ports and the introduction of
the scattering and transfer matrix for the electromagnetic coupling
through the screen. In that way also the impact of impedance
variations in the device under test can be investigated. Furthermore
a method is described to extend the frequency range for transfer
impedance and screening attenuation measurements.
Note that the symbols used in the following text are described in
paragraph 7.
cable sheath
cable under test
input voltage U
cable screen
coupling length
app. 2m
1
tube
terminating
resistor R1= Z1
or network analyzer
signal
generator
U1
U2
Figure 1: triaxial setup
Keywords: transfer impedance; screening attenuation; screening
effectiveness, scattering matrix, transfer matrix, cascaded multiports, Mason’s gain formula, triaxial set-up, IEC 62153-4 series
2. Scattering matrix S of screening
attenuation
1. Introduction
The coupling between two lines (the inner and the outer circuit) can
be depicted as a four port coupling device (see Figure 2). Ports 1-3
represent the inner circuit (device under test) and Ports 2-4 represent
the outer circuit (surrounding).
The screening effectiveness of screened cables, connectors and
cable assemblies is described by the (surface) transfer impedance
 and screening attenuation .
The relation between the transfer impedance and the screening
attenuation of the screen, for a uniform device under test (DUT) is
well described in the literature . At IWCS 2017 the first
time a paper was presented dealing with non uniform devices like
cable assemblies or connecting hardware . That paper presented
formulae which allow to take into account variations of the
impedance (and velocity) in the outer circuit (cable surrounding).
However the given formulae are limited to 4 segments and exclude
impedance variations in the inner circuit (DUT).
A new, more versatile and elegant approach is based on the theory
of cascaded multi-ports and the use of the scattering matrix S and
transmission matrix T. It allows to evaluate the impact of variations
of the impedance and velocity in the inner and outer circuit for any
number of segments.
Today the triaxial method (see Figure 1) is the preferred method to
measure the transfer impedance  and screening attenuation .
Other methods are the line injection method  for transfer
impedance and the absorbing clamp  or injection clamp 
method for screening attenuation.
The maximum frequency up to which the transfer impedance can be
measured depends on the load conditions of the test configuration
and the test length . In general the frequency is limited to about
100MHz. The IEC standard 62153-4-16  describes a method to
extrapolate the test results to higher frequencies for transfer
impedance or to lower frequencies for screening attenuation. But the
method is only applicable in the case of a matched inner circuit
(device under test). However transfer impedance measurements are
often made with a mismatched inner circuit. A new approach
applicable to any load conditions is presented.
2.1 Coupling matrix
S42
Port 2
Port 4
S21
S41
Port 1
Port 3
S31
Figure 2: 4-port coupling device
For passive and reciprocal devices like cables, cable assemblies or
connecting hardware the forward scattering parameters Sxy equal the
reverse scattering parameters Syx .
The coupling between the inner and outer circuit is loose therefore
the forward and reverse scattering parameters of the inner circuit
(S31=S13) and outer circuit (S42=S24) equal the (wave) attenuation of
the inner respectively outer circuit:
S IC  S 31  S13  e  1 l
(1)
SOC  S 42  S 24  e  2 l
(2)
For matched circuits the reflection coefficients are zero:
S11  S 22  S 33  S 44  0
(3)
In the case of matched conditions in the inner and outer circuit ,
 describe the formulae for the voltages coupled from the inner to
the outer circuit at the near end and the far end. But scattering
parameters are ratios of (square root) of power waves. Therefore the
formulae for the voltage ratios need to be extended by introducing
the square root ratio of the impedances in the inner and outer circuit:
CN  
ZT  Z F
1
1  e  1   2 l
2 Z1   1   2 
CF  
Z F  ZT
1
   l
 1  e  1 2   e 2 l
2Z1   1   2 



Z1
(4)
Z2

(5)
Z2
(6)
C F  S14  S 41  S 32  S 23
(7)
The scattering matrix for the coupled four port device is then
expressed by:
SC
CN
S IC
0
CF
CF
0
S OC
CN
Z 2  Z 2n
Z 2  Z 2n
rOC , f 
Z2 f  Z2
Z2 f  Z2
(12)
2.3 Transfer matrix T
Z1
C N  S12  S 21  S 34  S 43
 0

C
 N
 S IC

CF
rOC , n 
CF 

S OC 
CN 

0 
(8)
It is important to note that the reference impedance of this scattering
matrix is not the reference impedance of the vector network
analyzer, in general 50 Ohm. But the reference impedance of the
inner circuit (port 1 to 3) equals the impedance of the inner circuit.
The reference impedance of the outer circuit (port 2 to 4) equals the
impedance of the outer circuit. Hence the Sxx of the coupling matrix
equal zero (see formula (8))
The conversion from the scattering matrix S to the transfer matrix
T (also known as cascade matrix or transfer scattering matrix) is
well known for two ports. Recent work extended the T-Matrix of
2-ports to multi-ports . It is interesting to note that already
in 1961 Wilhelm Klein described the conversion between the
scattering matrix and transfer matrix for M x M multi-ports 
(page 90 ff). In the following the notations described by Wilhelm
Klein are used. Here the conversions for a 4-port are shown:
S I , I S II1,I 

S II1,I 
(13)
TI , I  TI , II TII,1II TII , I 


 TII,1II TII , I
(14)
S
 S I , I S II1,I S II , II
T   I , II
 S II1,I S II , II

T T 1
S   I , II II , II
 TII,1II
Where the S-Matrix and T-Matrix are split into 4 sub-matrices:
  S11

S
S    21
  S 31

  S 41
S12 

S 22 
S 32 

S 42 
 S13

 S 23
 S 33

 S 43
The coupling matrix is valid for matched circuits. To take into
account impedance variations between different cascaded
segments or the load conditions of the triaxial setup1 a junction
matrix is introduced.
 T11

T
T    21
 T31

 T41
T12 

T22 
T32 

T42 
T13

T23
T33

T43
The formulae given in  (page 61 ff) for a two port can be
extended to a 4 port device and leads at the near end to:
2.4 Triaxial set-up as a cascade of near end
junction, coupling and far end junction

 rIC, N

0
S JN  

2
 1  rIC, N

0

The resulting transfer matrix T of the triaxial set-up and
consequently resulting scattering matrix is obtained from a simple
matrix multiplication:
2.2 Junction matrix
0
1  rIC, N 2
rOC, N
0
0
 rIC, N
1  rOC, N 2
0
0
1  rIC, F 2



1  rOC, N 2 

0


 rOC, N 
(9)



1  rOC, F 2 

0


 rOC, F 
(10)
0
rOC, N
0
0
 rIC, F
1  rOC, F 2
0
0
Where the reflection coefficients r are obtained for a wave
propagating from the near end (generator side) to the far end
rIC , n
Z  Z1 n
 1
Z1  Z1n
rIC , f 
Z1 f  Z1
Z1 f  Z1
T14  

T24    TI , I

T34   TII , I

T44  
S I , II 

S II , II 
TI , II 

TII , II 
Ttriax  TJN  TC  TJF
And at the far end to

 rIC, F

0
S JF  

2
1
r

IC, F


0

S14  

S 24    S I , I

S 34    S II , I

S 44  
(11)
(17)
It is to note, that the short circuit at the near end of the outer
circuit, i.e. rOC,N=1, leads to a sub-matrix SII,I of the near end
junction matrix which is singular. Thus the matrix inverse does
not exist and the conversion to a T-matrix is not possible (formula
(13). Therefore a value close to zero, e.g. 10-9 Ohm, shall be taken
for Z2,n.
Figure 3 shows simulation results for the configuration indicated
in Table 1:
Table 1: parameters for simulation of triaxial set-up
Z0=50Ω ; L=2m
Z2=150Ω ; Z2,n=0Ω ; Z2,f=Z0 ; εr2=1,0 ; α2=0
In the triaxial method the outer circuit is short circuited at the
near end and mismatched at the far end.
(16)
For the determination of the reflection factors [formulae (11)(12)]
of the junction matrices [formulae (9)(10)] one has to consider
that the outer circuit has a short circuit at the near end (Z2,n=0) and
a mismatch at the far end (Z2,f=Z0). For the inner circuit (device
under test) the load at the near is in general Z1,n=Z0 and at the far
end depending on the used method Z1,f=0 or Z1,f=Z1 or Z1,f=Z0.
Z1= Z0 ; Z1,n= Z0 ; Z1,f= Z0 ; εr1=2,25 ; α1=0
1
(15)
RT=13,6mΩ/m ; MT=0,93nH/m ; CT=0,04pF/m
0
0
classical formulae
‐20
‐20
‐30
‐30
‐40
‐50
‐60
‐40
‐50
‐60
‐70
‐70
‐80
‐80
‐90
‐90
‐100
‐100
0
500
1000
1500
2000
frequency in MHz
2500
3000
0
Figure 3: cascaded multi-port vs. classical formulae
The results of the resulting coupling Striax,41=Striax,14 are in perfect
concordance with the results obtained from the classical formulae
described in .
A non uniform device under test like a cable assembly, with
connectors at the cable ends, or a cable junction, with connectors
between two cable segments, may have variations of the
impedance (and propagation constant) in the outer and inner
circuit. In that case the resulting coupling can be obtained by
cascading the different uniform segments. The impedance
variations between the different segments are taken into account
by the junction matrix. The resulting transfer matrix T of the
cascaded segments and consequently resulting scattering matrix S
is obtained from a simple matrix multiplication; see Figure 4 and
formula (18).
Junction
Near end
Segment
1
Junction
S1,2
Segment
2
…
Segment
n‐1
Junction
Sn‐1,n
Segment
n
Junction
Far end
(18)
TR  TJ N  TC , s1  TJ , s1s2  TC , s2  ...  TJ , sn1 , sn  TC , sn  TJF
IWCS 2018
IWCS 2017
‐10
coupling in dB
coupling in dB
‐10
500
1000
1500
2000
frequency in MHz
2500
Figure 5: cascaded multiport approach vs. wave
propagation and multiple reflections
3. Extrapolation of measurement results
to higher frequencies for transfer
impedance and lower frequencies for
screening attenuation
The triaxial set-up can be used to measure both the surface
transfer impedance  and the screening attenuation . The
transfer impedance is in general measured with a coupling length
of 0,5 m resulting in an upper frequency limit of around 100 MHz
whereas the screening attenuation is in general measured with a
coupling length of 2 m to 3 m resulting in a upper frequency limit
for the transfer impedance of around 10 MHz and a lower
frequency limit for the screening attenuation of around 100 MHz
Figure 6 shows the grey zone between electrically short
(measurement range for the transfer impedance) and electrically
long (measurement range for the screening attenuation). In the
example, the transfer impedance can be measured up to around 30
MHz using a coupling length of 50 cm and the screening
attenuation can be measured starting from 150 MHz using a
coupling length of 200 cm.
10000
0
Figure 5 shows simulation results for the triaxial configuration
indicated in Table 2:
Table 2: parameters for simulation in triaxial set-up
S21 ‐ 50cm
‐10
Zt
grey zone
‐20
1000
‐30
‐40
mOhm/m
Z0=50Ω
4 segments
S21 ‐ 200cm
dB
reference
segments
3000
100
‐50
#
1
2
3
4
Z2
150Ω
100Ω
50Ω
150Ω
inner circuit
L
1m
0,25m 0,5m
0,25m
Z1= Z0 ; εr1=2,25 ; α1=0 ; equal for all segments
outer circuit
εr2=1,0 ; α2=0 ; equal for all segments
screen
RT=13,6mΩ/m ; MT=0,93nH/m ; CT=0,04pF/m ;
equal for all segments
The results of the resulting coupling SR,41=SR,14 are in perfect
concordance with the results obtained from the approach
presented at IWCS 2017 . I.e. the results and conclusions
presented there are confirmed and valid!
‐60
‐70
10
‐80
‐90
‐100
1
0,01
0,1
1
MHz
10
100
1000
Figure 6: simulation of the scattering parameter S21
(left hand scale) and the transfer impedance (right hand
scale) for a single braid screen
The IEC standard  describes a method to extrapolate the test
results of transfer impedance to higher frequencies and the test
results of screening attenuation to lower frequencies. But the
method is limited to a matched inner circuit, whereas transfer
impedance measurements are often made with a mismatched inner
circuit (device under test)
P1  S JN31 SC41 S JF42
The first idea was to use the described approach of cascaded
multi-ports to extract (de-embed) the coupling matrix (TC
respectively SC) from the measured resulting S-Matrix (SM) and
the corresponding resulting T-Matrix (TM):
P3  S JN31 SC31 S JF11 SC43 S JF42
TC  TJN 1  TM  TJF 1
(19)
However in the triaxial setup the scattering parameters can only
be measured from two ports, i.e. port 1 and port 4 (see Figure 2).
Although with the short circuit at the near end of the outer circuit
(port 2) we know S21=S12=S23=S32=S24=S42=0 and S22=1, we have
the problem that in the case of an inner circuit which is matched at
the far end (S33=0) we do not know S34=S43. In the case of an
inner circuit with a short circuit at the far end we know
S31=S13=S43=S34=0 and S33=1 and the measured matrix could be
written as:
 S11 0 0 S14 
0 1 0 0 

SM  
0 0 1 0 


 S 41 0 0 S 44 
(20)
But the sub-matrix SII,I is singular and the matrix inverse does not
exist and the conversion to a T-matrix is not possible. Simulations
revealed that it is not sufficient to replace the zeros by a value
close to zero.
Therefore a new approach is based on the signal flow graph of the
triaxial setup and Mason’s gain formula.
3.1 Signal flow graph of the triaxial setup and
Mason’s gain formula
The signal flow graph of the triaxial setup is illustrated in Figure 7.
The middle section corresponds to the coupling between both
circuits and the sections at the left and the right correspond to the
load conditions of the measurement setup. For screened devices the
coupling is loose (weak) between the inner circuit (DUT) and the
outer circuit (environment). Therefore it is reasonable, for the sake
of simplification, to neglect the feedback from the disturbed circuit
to the disturbing circuit.
P2  S JN31 SC21 S JN 44 SC42 S JF42
(21)
P4  S JN31 SC31 S JF11 SC23 S JN 44 SC42 S JF42
The signal flow graph contains 6 loops:
L1  S JN33 SC11
L4  S JN44 SC22
L2  SC33 S JF11
L5  SC44 S JF22
(22)
L3  SC31 S JF11 SC13 S JN33 L6  SC42 S JF22 SC24 S JN44
Applying Mason’s formula and taking into account that in SC, the
coupling matrix, S11= S22= S33= S44=0 leads to the resulting S41 of
the triaxial setup:
S R41 
P1  P2  P3  P4
1   L3  L6   L3 L6
(23)
After some further conversions and taking into account the short
circuit at the near end of the outer circuit one obtains:
S R41  1  r1n 2 1  r2 f 2 




CF 1  r1 f e 1  2  L  C N e  2 L 1  r1 f e 1  2  L 


2    L
1  r1n r1 f e 2 1L  r2 f e 2 2 L  r1n r1 f r2 f e  1 2 
(24)
Figure 8 shows the comparison between the results obtained from
the approach of cascaded multiports (with the conversion between S
and T matrix and multiplication of the T matrices) with the results
obtained using Mason’s formula. The configuration of the triaxial
setup is shown in Table 3 with a mismatched inner circuit having a
short circuit at the far end.
Table 3: parameters for simulation of triaxial set-up
Z0=50Ω ; L=2m
Z1= 75Ω ; Z1,n= Z0 ; Z1,f= 0 ; εr1=2,25 ; α1=0
Z2=150Ω ; Z2,n=0Ω ; Z2,f=Z0 ; εr2=1,0 ; α2=0
RT=13,6mΩ/m ; MT=0,93nH/m ; CT=0,04pF/m
0
Mason's formula
‐10
resulting S41 in dB
‐20
‐30
‐40
‐50
‐60
‐70
‐80
‐90
‐100
0
Figure 7: signal flow graph of the triaxial setup
To obtain the resulting scattering parameter S41 one has to find in
the signal flow graph all paths (P) from port 1 to port 4 and all loops
(L).
It exist 4 paths to get from port 1 to port 4:
500
1000
1500
2000
frequency in MHz
2500
3000
Figure 8: cascaded multi-port vs. Mason’s formula
The results of both approaches are in perfect concordance. Thus
formula (24) obtained from Mason’s gain formula can be used to
convert the measured scattering parameter SR41 to the transfer
impedance ZT.
3.2 Conversion between resulting (measured)
scattering parameter SR41 and transfer
impedance ZT
4. Measurement versus simulation results
In most cases the capacitive coupling through the cable screen can
be neglected. This is in particular true for screens composed of
metal foils, dense single braids or double braids. In this case
formula (24) can be rearranged to obtain the transfer impedance ZT:
ZT
Z F 0
 S R41
2 Z1Z 2
1  r1n 2 1  r2 f 2

1  r1n r1 f e21L  r2 f e 2 2 L  r1n r1 f r2 f e 21  2  L 


   L
1  e   1  2  L
1 e  1 2
   L
   L 
e 2 L 
1  r1 f e  1 2  
1  r1 f e  1 2  
1   2
 1   2




(25)

For low frequencies (L&lt;&lt;1) formula (25) can be rewritten as:
ZT  S R41
Z 2 f  Z1n  Z1 f

In formula (25) it is necessary to know the propagation constant of
the inner and outer circuit. A method to obtain them from
measurement is described in IEC 62153-4-16 .
(26)
2 L Z1n Z 2 f
Figure 9 shows the comparison between the results of transfer
impedance obtained from formula (25) and the commonly used
conversion formula between the measured forward transfer
scattering parameter and transfer impedance as described in IEC
62153-4-3 . The configuration of the triaxial setup is shown in
Table 3 with a mismatched inner circuit having a short circuit at the
far end.
Table 4: parameters for simulation of triaxial set-up
Figure 10 shows the results of the conversion from the measured
scattering parameter SR41 to the transfer impedance for a RG59 type
cable. The DUT was matched at the near end (generator side) and
far end. The conversion was done using a relative dielectric
permittivity of the inner circuit (DUT) of 2,3 (PE dielectric) and 1,1
of the outer circuit (PVC jacket in tube). The blue line and the green
line show the transfer impedance when measured with a 2m
coupling length respectively 0,5m and using the standard
conversion formula as described in  and formula (26). One can
see that the upper frequency limit for the validity of the transfer
impedance is 10MHz for the 2m length and 40MHz for the 0,5m
length. The red line shows the conversion to transfer impedance of
the 2m length using formula (25). Here the upper frequency limit
for the validity of the transfer impedance is 200MHz. The observed
peaks at higher frequencies are due to the capacitive coupling
impedance ZF which is not exactly zero (single braid with “low”
coverage) and due to uncertainties in the dielectric permittivity used
in the conversion formula (25).
Z0=50Ω ; L=0,5m
Z1= 75Ω ; Z1,n= Z0 ; Z1,f= 0 ; εr1=2,25 ; α1=0
Z2=150Ω ; Z2,n=0Ω ; Z2,f=Z0 ; εr2=1,0 ; α2=0
RT=13,6mΩ/m ; MT=0,93nH/m ; CT=0
The transfer impedance obtained from formula (25) corresponds, as
expected, to the transfer impedance obtained from the screen
parameter (RT, MT, and CT). But using the formula described in IEC
62153-4-3 to convert the measured forward transfer scattering
parameter to transfer impedance limits the upper frequency for the
transfer impedance in milli‐Ohm/m
100 000
ZT of screen
ZT from S41
ZT from IEC62153‐4‐3
10 000
Figure 11 shows the test results of transfer impedance for a single
braided coaxial cable with 50Ω impedance. The DUT is short
circuited at the far end (Z1f=0). The results are shown for 3
different coupling lengths 35cm, 100cm and 200cm. The
conversion from the measured scattering parameter SR41 to the
transfer impedance was done according to IEC 62153-4-3 . The
results show that the cut-off frequency for the transfer impedance
is decreasing with increasing length from 60MHz for 35cm to
10MHz for 200cm.
1 000
100
10
1
0
0,01
0,1
1
10
frequency in MHz
100
Figure 10: Example for the conversion of measured
scattering parameter SR41 to the transfer impedance of a
RG59 type cable; inner circuit matched; assuming
relative dielectric permittivity of 2,3 and 1,1 for the
inner, respectively outer circuit
1000
Figure 9: comparison of formulae for conversion
between forward transfer scattering parameter and
transfer impedance
The IEC standard 62153-4-16  to extend the frequency range is
only applicable for a matched DUT and cannot be used for a DUT
with a short circuit at the far end. The newly introduced equation
(25) however can be used for any load conditions. This is shown
for the above cable in Figure 12. The results for the coupling
length of 100cm and 200cm have been extrapolated using
equation (25). The conversion was done using a relative dielectric
permittivity of the inner circuit (DUT) of 2,3 (PE dielectric) and
1,0 of the outer circuit (cable jacket was removed). The cut-off
frequency has been increased from 10MHz for 200cm
respectively 20MHz for 100cm to 200MHz. The observed residual
peaks at higher frequencies are due to the capacitive coupling
impedance ZF which is not exactly zero and due to uncertainties in
the dielectric permittivity used in the conversion formula (25).
Furthermore a method is described to extrapolate measurement
results of transfer impedance to higher frequencies respectively
screening attenuation to lower frequencies. This covers now the
grey zone between electrically short and long devices and extends
the method described in  to any load conditions.
6. Outlook
A future project is to model the coupling attenuation of screened
balanced cables as a 6 port. The coupling attenuation is the
resulting coupling between first the differential mode to the
common mode and second between the common mode and the
surrounding.
7. Symbols
The symbols used throughout the paper are presented in Table 5.
Table 5: symbols
Figure 11: measurement of transfer impedance of a
single braided cable with 50Ω impedance; inner circuit
short circuit; coupling length 35cm, 100cm, 200cm
Figure 12: conversion of measured scattering
parameter SR41 to the transfer impedance of a single
braided cable with 50Ω impedance; equation (25); inner
circuit short circuit; assuming relative dielectric
permittivity of 2,3 and 1,0 for the inner, respectively
outer circuit
5. Conclusions
In this paper I have presented the scattering matrix S and transfer
matrix T for the coupling between two lines. The introduction of the
scattering matrix for the junction allows to model the different test
configurations (i.e. load conditions) for the measurement of the
transfer impedance and screening attenuation.
By using the theory of cascaded multi-ports it is possible to evaluate
the resulting screening attenuation of composed (non uniform)
devices like cable assemblies where the local impedance in the inner
circuit (device under test) and outer circuit (surrounding) may be
varying. The results presented at IWCS 2017  are confirmed and
in addition it is now possible to take into account impedance
variations in the inner circuit (device under test).
CN, F
coupling transfer function at the near respectively far
end
CT
capacitive coupling through the screen
DUT
device under test
f
frequency in MHz
L
coupling length
MT
magnetic coupling through the screen
rIC,n, f
reflection coefficient of the inner circuit at the near
respectively far end
rOC,n, f
reflection coefficient of the outer circuit at the near
respectively far end
RT
DC resistance of the screen
S
scattering parameter matrix
SC
scattering parameter matrix of the coupled four port
SIC
forward (reverse) transfer scattering parameter of
inner circuit S31, S13
SJ N, JF
scattering parameter matrix of the load conditions at
the near respectively far end
SJ, sn-1,sn
scattering parameter matrix of a junction between
section n and n-1
SOC
forward (reverse) transfer scattering parameter of
outer circuit S42, S24
SR41
resulting forward coupling scattering parameter of
the triaxial setup
Sxy
scattering parameter of multiport
T
transfer parameter matrix
TJ N, JF
transfer parameter matrix of the load conditions at
the near respectively far end
TJ, sn-1,sn
transfer parameter matrix of a junction between
section n and n-1
Z1
impedance of the inner circuit
Z1n, f
load impedance at the near respectively far end of
the inner circuit
Z2
impedance of the outer circuit
Z2n, f
ZF
ZT
load impedance at the near respectively far end of
the outer circuit
capacitive coupling impedance
Z F  jCT Z1Z 2
transfer impedance
ZT  RT  jM T
α1
attenuation in the inner circuit
α2
attenuation in the outer circuit
γ1
propagation constant in the inner circuit
γ2
propagation constant in the outer circuit
εr1
relative dielectric permittivity of the inner circuit
εr2
relative dielectric permittivity of the outer circuit

circular frequency (=2f)
8. Acknowledgments
This paper has been written in memory to Otto Breitenbach (†2003)
the father of the triaxial setup to measure the screening attenuation
of cable screens . Otto Breitenbach was the head of R&amp;D at
Kabelmetall Nuremberg (today Nexans) and had the idea to extend
the triaxial method – at that time used for transfer impedance
measurements up to 100MHz – to the GHz frequency range for the
measurement of the screening attenuation.
Thanks also to Lauri Halme, Thomas Schmid and Bernhard Mund
and the members of IEC TC46 WG5 for encouraging this laborious
work which is the continuation of the work presented at IWCS 2017
.
9. References
 IEC TS 62153-4-1 2014: Metallic communication cable test
methods - Part 4-1: Electromagnetic compatibility (EMC) Introduction to electromagnetic screening measurements
 IEC 62153-4-3: Metallic communication cable test methods Part 4-3: Electromagnetic compatibility (EMC) - Surface
transfer impedance - Triaxial method
 IEC 62153-4-4: Metallic communication cable test methods Part 4-4: Electro Magnetic Compatibility (EMC) - Test method
for measuring of the screening attenuation as up to and above 3
GHz, triaxial method
 L. Halme and B. Szentkuti, The background of
electromagnetic screening measurements of cylindrical
screens, Technische Mitteilung PTT Nr. 3 1988
 O. Breitenbach and T. H&auml;hner, Kabelschirmung im &Uuml;bergang
von MHz zu GHz-Frequenzen, NTZ Bd. 46 Heft 8 1993
 O. Breitenbach, T. H&auml;hner, B. Mund, Screening of cables in
the MHz to GHz frequency range, extended application of a
simple measuring method, Colloquium on screening
effectiveness measurements, Savoy Place London, 6 May
1998, Reference No: 1998/452
 T. H&auml;hner and T. Schmid, Modeling of the triaxial setup to
measure the screening attenuation in the case of different
coupling sections, IWCS proceedings 2017
 IEC 62153-4-16: Metallic communication cable test methods Part 4-16: Electro Magnetic Compatibility (EMC) - Extension
of the frequency range to higher frequencies for transfer
impedance and to lower frequencies for screening attenuation
measurements using the triaxial set-up
 IEC 62153-4-6: Metallic communication cable test methods Part 4-6: Electromagnetic compatibility (EMC) - Surface
transfer impedance – Line injection method
 IEC 62153-4-5: Metallic communication cable test methods Part 4-5: Electromagnetic compatibility (EMC) - Coupling or
screening attenuation - Absorbing clamp method
 IEC 62153-4-2: Metallic communication cable test methods Part 4-2: Electromagnetic compatibility (EMC) – Screening
and coupling attenuation - Injection clamp method
 W. Klein, Die Theorie des Nebensprechens auf Leitungen,
Springer Verlag 1961
 Hans-J&uuml;rgen Michel, Zweitor-Analyse mit Leistungswellen,
Teubner Studienb&uuml;cher, Stuttgart, 1981
 James Frei, Xiao-Ding Cai, Stephen Muller, Multiport SParameter and T-Parameter Conversion With Symmetry
Extension, IEEE Transactions on Microwave Theory and
Techniques, Vol. 56, No. 11, November 2008
 Janusz A. Dobrowolski, Scattering Parameters in RF and
Microwave Circuit Analysis and design, Artech House, 2016
 W. Klein, Grundlagen der Theorie elektrischer Schaltungen,
10. Author
Thomas H&auml;hner received in 1989 his
Diploma in Electrical Engineering (Dipl.Ing.
(FH))
with
emphasis
on
Telecommunications Technology from the
Georg Simon Ohm University of Applied
Science in Nuremberg (Germany) and in
2000
his
Diploma
in
from the University of Applied Science in
in Wildau (Germany).
Thomas H&auml;hner is in the cable business since 1990 where he
joined Nexans in Nuremberg (Germany) as an R&amp;D engineer of
radio frequency and data transmission cables and manager of the
RF test laboratory. Today he is still with Nexans and since
September 2015 Technical Manager for Aerospace-DefenseMedical based in Paris area (France).
He is active since more than 20 years in standardization
committees. In 2010, Thomas H&auml;hner was granted for the 1906
IEC Award in recognition of his outstanding technical
contribution in developing, writing and finalizing TC 46’s IEC
62153-4 series (Metallic communication cable test methods Electromagnetic compatibility). Thomas is Chairman of IEC
technical committee TC46 (Cables, wires, waveguides, R.F.
connectors, R.F. and microwave passive components and
accessories) and Chairman of CENELEC technical committee
TC46X (Communication Cables).
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