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Coarse WDM/CDM/TDM concept for optical packet transmission inmetropolitan
and access networks supporting 400 channels at 2.5 Gb/speak rate
Article in Journal of Lightwave Technology · January 2001
DOI: 10.1109/50.908792 · Source: IEEE Xplore
7 authors, including:
Thomas Pfeiffer
Jörg-Peter Elbers
Nokia Bell Labs (Stuttgart Germany)
ADVA Optical Networking SE
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Coarse WDM/CDM/TDM Concept for Optical Packet
Transmission in Metropolitan and Access Networks
Supporting 400 Channels at 2.5 Gb/s Peak Rate
Thomas Pfeiffer, Jens Kissing, Jörg-Peter Elbers, Bernhard Deppisch, Martin Witte, Harald Schmuck, and
Edgar Voges
Abstract—To improve the networking flexibility in the
metropolitan and access area, the granularity in the optical domain has to be increased above that in the core network requiring
more channels at lower bit rates. Pure dense wavelength division
multiplexing (DWDM) as it is applied in the core does not seem to
meet this requirement at affordable cost. We propose and analyze
a network based on hybrid optical multiplexing techniques including wavelength, code, and time division multiplexing. Applied
to optical packet transmisssion this approach enables several 100
all-optical channels between end users and headend with average
bit rates up to 100 Mb/s per channel while keeping the installation
and maintenance cost at a minimum.
Index Terms—Code division multiplexing, multiaccess communication, network reliability, optical crosstalk, optical noise, packet
switching, time division multiplexing, wavelength division multiplexing.
N metropolitan area networks (MAN) the required number
of independent optical channels is likely to increase to 100
with channel bit rates up to the Gb/s range within the near future.
This fine granularity enables flexible utilization of fiber bandwidth and simple network reconfiguration. Network operators
will be able to lease single optical channels or groups of optical channels to service providers on demand. Even in access
networks with multiple optical channels, the increase of total
throughput to the Gb/s range is anticipated [1]. It is, however,
questionable whether the straight forward scaling of today’s
dense wavelength division multiplexing (DWDM) technology,
as it is presently used in the core network, to the specific requirements of MAN and access networks is an affordable option for the realization of 100 optical channels. Specially for
packet-based services like Internet with average data rates per
user that are up to two orders of magnitude lower than the peak
rate [2] the implementation of DWDM in network parts close
to the end user will not be a realistic option. The spectral efficiency of such networks would be extremely poor making this
approach very expensive.
Manuscript received April 19, 2000. This work was supported in part by the
German Ministry for Research within the Photonik II and KomNet projects.
T. Pfeiffer, B. Deppisch, M. Witte, and H. Schmuck are with Alcatel Corporate Research Center, D-70499 Stuttgart, Germany
J. Kissing J.-P. Elbers, and E. Voges are with Lehrstuhl für Hochfrequenztechnik, University Dortmund, Germany.
Publisher Item Identifier S 0733-8724(00)11609-3.
With increasing channel number and taking into account the
optical bandwidth of fiber networks that is mostly limited by
the gain bandwidth of optical amplifiers the channel spacing
will rapidly drop below the current 100 GHz and 50 GHz spacings. The technological and economical effort that has to be
spent to realize even smaller optical channel spacings increases
dramatically and is hardly justified by the required channel capacity. The main issues to be encountered at the optical transmitters and receivers, when the channel spacing drops down to
the 10 GHz range, are selection, tuning and control of the laser
center frequency, the requirement for external modulation even
for low channel bit rates and the fabrication and control of narrowband optical filters. Even without considering additional impairments from fiber transmission (e.g., channel crosstalk due to
nonlinear effects) and optical amplification (gain flatness, transient effects) these systems are expected to become extremely
demanding with respect to component specifications and network management.
For packet-based transmission systems the additional application of time and/or code division multiplexing (TDM,
CDM) techniques in the network is a favorable way to increase
the channel number. TDMA (time division multiple access] is
a viable option for increasing the channel number in simple
fiber network structures with limited capacity to several 100
[3]. CDM techniques can also be applied to more complex
network architectures, since some optical CDM approaches
can be realized to support asynchronous operation of channels.
Moreover, with asynchronous CDM a statistical multiplexing
gain can be achieved in packet transmission networks enabling
a large number of channels to be allocated. The bursty nature
of the traffic helps to reduce the performance degradation due
to multiple access interferences. Alternatively, optical TDMA
can be applied to a limited part of the network, typically in
the access part, whereas in the MAN part WDM and CDM
techniques are implemented.
There have been different proposals how to implement optical CDM applying time domain [4], [5] or wavelength domain
[6], [7] coding. Also, the combination of WDM and CDM techniques has been proposed in some papers [8]–[10]. We favor
spectrally encoded CDM due to the fact that, in contrast to most
time domain encoded CDM approaches, the different channels
do not have to be synchronized to each other to achieve full
orthogonality between codes. This guarantees independence of
the optical multiplexing scheme from the network architecture.
Moreover, in time domain CDM the optical pulses would have
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to be much shorter than the bit period, so that the optical components must satisfy Gb/s specifications even for Mb/s channel
bit rates.
In this paper, we present a detailed system proposal for the
implementation of CDM enhanced WDM based on previous
experimental and theoretical studies that we have performed on
a special form of spectrally encoded optical CDM. Our CDM
approach applies periodic spectral encoding of broadband
optical sources like light emitting diodes (LEDs) [11]. This
approach has some advantageous features like stability and
insensitivity of system performance with respect to component
drift and specifications making it very attractive for MAN and
access applications. On the networking side the benefits are
asynchronous operation of different optical channels, support
of different signal formats and ease of redistributing electrical
bandwidth among optical channels. After a short review of the
operating principle these aspects will be discussed in the next
section with reference to experiments that have been performed
in multichannel system demonstrators transmitting continuous
data streams. Then the basic system limitations, namely
channel crosstalk and optical intensity noise, will be discussed
yielding some simple rules of thumb to estimate the achievable
channel number and bit rate. A full numerical simulation is
used for system evaluation when using different transmitter
and receiver configurations. From this analysis two network
scenarios for optical multichannel transmission are derived that
also include time domain features like statistical multiplexing
and synchronous TDMA within limited network areas. At the
end impacts from transmission of broadband optical signals are
discussed, specially from chromatic dispersion induced signal
The concept of periodic spectral encoding as proposed by
Möller [11] is a generalization of coherence multiplexing
[6] in the sense that all kind of optical filters having periodic power transmission characteristics are considered for encoding and decoding. The optical output power of a thermal
light source, like an LED, is intensity modulated by the
electrical data. After emission the broadband spectrum is optically shaped using passive filters with periodic transmission
functions like Mach–Zehnder (MZ) or Fabry–Perot (FP) filters (Fig. 1). The periodicity of the transmission function
for these type of filters is given in terms of the free spectral range (FSR) which is defined by the filter round trip
. The round trip time is determined
time via
by the filter geometry and given as
for FP filters, where
MZ filters and
and are the differential delay and the cavity length of the
is the group index of the cavity mafilters, respectively,
terial and is the vacuum speed of light. In the network
different FSR are allocated to different optical transmitters
and define the codes in the system. At the receiver an optical filter with periodic transmission function is tuned by
matching its FSR to the desired channel. The filter types
at transmitter and receiver do not have to be identical, only
, respectively,
the FSR or the round trip times
must be matched (Fig. 2). To achieve nearly perfect orthogonality between channel codes a differential receiver set-up is
used that removes the offset of the received signal power in
. In the theoretical analysis of this
Fig. 2 at
paper, (cf. Section III) it will be assumed that the common
mode rejection ratio (CMRR) of the differential receiver is
infinite, i.e., the signal level and group delay for the undesired channels are assumed to be identical on the signal arm
and on the reference arm of the receiver. In our 155 Mb/s
differential receiver the CMRR was better than 40 dB for
frequencies up to 200 MHz [12]. The optical source spectra
are assumed to be much broader than the FSR of the transmitter filters, or in other words, the source coherence time
defined by the source optical power density spectrum
[13] be much smaller than
. Then the relative positioning
the filter round trip time
of the source spectrum center frequency and the filter transmission function is irrelevant which leads to the first benefit
of this approach: the selection of the source spectra and the
filter transmission is not critical as long as
for every
the difference of the round trip times
is in the order of the source
pair of transmitters
coherence time
or larger. [In practice the gain spectra of
the optical amplifiers define the coherence time, since usually
they are narrower ( 30 nm) than the source spectra ( 50
nm).] This has been experimentally verified in an 8 155
Mb/s system where the transmitter LEDs were taken from two
different suppliers (MRV and Anritsu) with different spectral
53–68 nm]
width [FWHM (full width at half maximum)
and center wavelengths (1539–1551 nm) [12]. The filter FSR
in this system (MZ at the transmitters, FP with finesse 4 at
the receiver) were chosen in the range 10–20 GHz, the power
extinction of the transmitter filters was 10–13 dB and slightly
polarization dependent, the FP receiver filter extinction ratio
was only 9 dB. Nevertheless the system operated with bit
. The second benefit of peerror rate (BER)
riodic encoding is that only the difference in filter roundtrip
for any two transmitters needs to be larger
is not
than some picoseconds. The absolute value of the
important as long as they are much larger than . From that
it follows that even temperature changes of more than 100
C do not affect the system performance as proven by experiment [14]. The reason for this unsensitivity to temperature
changes is the low temperature coefficient of the refractive
C for fused silica [15]). Our
index of glass (about 10
MZ encoding filters were made from standard 3 dB fiber
for the above tempercouplers, so that
ature range. The fine tuning of the receiver filter to drifting
transmitter filter FSR was accomplished by applying a local
low-frequency dithering technique in the receiver [16], but
no control of the transmitter filter FSR was required to keep
The robustness of the system with respect to component specifications and drift as demonstrated above is unique to the approach applying periodic spectral encoding with broad source
spectra and small filter FSR. Other code multiplex systems like
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Fig. 1.
Principle set-up of CDM system applying periodic spectral encoding.
Fig. 2. Principle of periodic spectral encoding and decoding (left). Received optical power with single-ended detection around point of optimum tuning (right).
Encoding and decoding filter type have been exchanged with respect to Fig. 1 for more instructive visualization. Gaussian source spectrum is assumed with 2 ln(2)= .
-sequence encoded systems [7] are not expected to show similar robustness due to the requirement of exact matching of the
aperiodic spectra at transmitter and receiver.
With systems as shown above, where the individual CDM
channels were modulated with independent continuous data
streams, different transmission experiments have been per155 Mb/s over 111 km field
formed. Transmission of 8
installed standard single-mode fiber (SMF) was demonstrated [17] as well as over 204 km Alcatel TeraLight™ fiber
ps/nm km @ 1550 nm) with
each channel (Fig. 3). Transmission and splitting losses were
compensated by three cascaded erbium-doped fiber amplifiers
(EDFAs) and the chromatic dispersion of the fiber link was
compensated by using matched lengths of dispersion compensating fiber (DCF). In more complex networks including
parallel groups of transmitters in a tree-like configuration the
capability of the chosen CDM approach to support different
channel bit rates was experimentally demonstrated [12]. In
these experiments, it was also found that the length tolerance
of the dispersion compensation using DCF was at least 4 km
SMF at this channel bit rate (cf. Section V). Even transmission
of quasi-analog electrical signals (16-QAM subcarriers) over
optical CDM parallel to baseband signals within the same
network was experimentally demonstrated [14]. These last
features cannot easily be realized with CDM systems based on
time domain coding.
After having reviewed the basic features of the optical CDM
approach applying periodic spectral encoding we will now dis-
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Fig. 3. Transmission of 8 3 155 Mb/s optical CDM signals over 204 km Alactel
TeraLight™ fiber.
cuss the main factors that pose an upper limit to the achievable
system capacity.
The BER is evaluated using the -factor approximation
, where erfc denotes the complementary error function and is defined as
with the current
for the “1” and “0” signal and the
and . The noise on the “1”
associated standard deviations
and on the “0” is assumed to be identical and is set to 0, so
with the differential receiver current and its
standard deviation . The main contributions to
from (thermal) receiver noise, shot noise, optical intensity
noise, and crosstalk. The variance of the latter two disturbances
increases linearly with electrical signal power and are responsible for the potential occurrence of a BER floor, hence they
decide on the feasibility of a system configuration. They will
now shortly be discussed, under simplifying assumptions, how
they are influenced by the width of the optical source spectra
and by the optical filters. We first focus the discussion on
Gaussian source spectra and on combinations of MZ and FP
filters. The results are characteristic of the discussed system
and are only slightly modified if other spectra or filters are
used. In the next section of this paper realizations are presented
that are more useful in systems with many channels.
When in the following we talk about source spectra, it is assumed that they are not modified during transmission due to e.g.,
optical amplifiers. Otherwise they have to be replaced in the formulae by the respective spectra at the receiver. Some basic considerations about crosstalk have been discussed in [18] (note:
the symbols used here slightly differ from those in [18]). Here,
we derive the number of optical channels that could be allocated
in a system, if crosstalk was the only limiting factor taking into
account the data statistics. For broad source spectra
compared to the filter FSR) with total power
photocurrent after the receiver filter is given by the transmitter
and receiver filter transmission functions
denotes the average over the
optical frequency . The photodetector response is omitted here
for simplicity. With a differential receiver the signal current
is given by
, where
is the difference between the receiver transmission function in the filtered arm
and the reference arm
and with the periodicity
is detuned from the transmitter
, the signal
varies in an oscillatory manner (Fig. 2) the details of which depend on the specific choice of filters and source spectra. Specially with a FP–FP combination for transmitter and receiver
filter the envelope of the tuning curve yields maxima also around
is a rational number. This is in contrast
points, where
to systems, where, for example, the receiver filter is an MZ filter
is an inyielding maxima only around points, where
teger. So in an FP–FP system the allocation of transmitter FSR
as the ratio of large
is preferably done by choosing
prime numbers, whereas in an –MZ system ( being either
MZ or FP) the
can be allocated to the transmitters
equidistantly within one octave
or larger [18]. To simplify the discussion the latter case is assumed with identical Gaussian source spectra for all transmit.
ters described by
Then the envelope of the tuning curve for an MZ receiver filter
is described as function of
as [18]
. The
with FWHM given by
in (1) is the second coefficient of the transmitter
filter cos-series
with the
for MZ or
for FP with
finesse . In the following, it is assumed that all transmitters
continuously send NRZ signals (non return to zero) with equal
for “0” and “1.” This results in a
binomial probability distribution of “1” signals at the receiver
which for many interfering transmitters is approximated by
a Gaussian distribution of the crosstalk current
with the mean value
, where
is calculated from (1) for all transmitters
. The standard deviation
is given
, where
for channels
in the
transmitting synchronously to each other and
asynchronous case [19]. With these results the crosstalk-limited
-factor for the channel with
denotes the
integer part of its argument) can now be expressed as
with source FWHM
The number of addressable codes is very large (cf. Fig. 5)
if transmitter or receiver filter or both are MZ like filters. In
case of an FP/FP combination the crosstalk around points
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is a rational number strongly reduces the
useful number of codes. However, even with MZ filters not all
codes can be simultaneously used, since the intensity noise at
the receiver imposes a more restrictive limit to the achievable
BER. For single ended detection, i.e., no differential receiver,
the intensity noise of a broadband thermal light source with
for Gaussian and
coherence time
for rectangular spectra) is given by
[20]. Here is the photocurrent and
is the noise
equivalent bandwidth defined via the normalized electrical
. It
receiver transfer function
was assumed that the detected light is linearly polarized. For
partly polarized light the electrical noise power is reduced
denotes the degree of polarization
for polarized and unpolarized light,
respectively [13]. If the source spectrum is passively filtered by
a periodic optical filter with transmission function
the relative intensity noise is enhanced by a factor
for MZ and
for FP] which can be regarded as the noise figure of the
optical filter for thermal light. The effective coherence time
independent passively filtered thermal light sources is
with the superposed optical
. For
power density spectrum
identical source spectra
with coherence time
this reduces to
denotes the average over the optical
where again
frequency . This spectrum is filtered at the receiver and differentially detected. With the effective receiver filter function
the electrical noise power at the
output of the differential receiver is
denotes the photocurrent that would be measured if the total
was directly detected with
a single photodiode:
. With the signal current
, given by
, the optical
for transmitter
in case of identical
noise limited -factor for transmitter
Gaussian source spectra and with MZ receiver filter is
It was assumed that the noise originating from all transmitters
is slightly larger
is identical although the photocurrent from
than the others. In practice, however, the above approximation
is valid as supported from experiments. Furthermore, the optical
has only entered the formula by 50% to account for
the data statistics [19]. The -factor in (5) can be optimized
Fig. 4. Optimum transmitter filter finesse for FP–MZ filer combination.
by the choice of the transmitter filter type and specifications.
For FP transmitter filters with finesse [
] the term in the denominator
is minimized with the optimum finesse
This relation is depicted in Fig. 4 showing optimum finesse
values that can be easily realized from a technological point of
view. The reason for the existence of an optimum finesse lies
in the fact that with increasing the differential signal current
degrades and the noise figure of the transmitter filters increases,
but on the other side the interchannel beat noise is reduced. With
the optimum finesse the optical noise limited -factor in the
floor is related to the number of active channels by
. EquaThe last approximation is well suited up to
tion (7) indicates that systems, where
is valid, show the same error floor under the above assumptions, specially that the filter finesse is optimized to the number
of active transmitters. This is, of course, not feasible in practice. However, the dependence of the -factor on filter finesse
is weak at large
so that
10–20 is a good choice in
many cases. Equation (7) indicates that the system employing
FP/MZ filters performs slightly better as compared to MZ/MZ
systems like coherence multiplexed systems [6]. The electrical
signal-to-noise ratio (SNR) with respect to MZ/MZ systems is
improved by about 3 dB for six channels and depends more
weakly on channel number as compared to MZ/MZ systems.
This improvement originates from the use of FP transmitter filters which reduce the optical interchannel beat noise.
that are allowed to be active
The number of channels
at the same time is reduced considerably as compared to the
. The noise limited -factor
number of addressable codes
for given
is determined by the ratio
[see (7)].
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Fig. 5. Number of addressable codes
(for parameters see text).
and of active channels
For a given number of channels
to be addressed, on the
other hand, the crosstalk limited -factor is determined by the
[see (2)]. The number of addressable codes
and the number of active channels are compared as function of
the source spectral width in Fig. 5 for FP transmitter filters with
FSR 10–20 GHz and optimized finesse. The requirement to
in the floor (
be met was
at 155, 622, and 2500 Mb/s (noise equivalent electrical bandwidth 122, 488, 1961 MHz with fourth-order Bessel filters). It
was assumed that the channels were synchronized and that the
detected light was linearly polarized, both assumptions yielding
lower numbers as compared to the opposite cases.
There are many simplifications made in the calculations
shown in Fig. 5, as indicated in the text. However, comparison with results from a full numerical model show that the
results presented in the figure can serve as a guideline to the
performance to be expected with such kind of systems. The
calculations presented in the next section are based on a more
complex mathematical description.
In this section, we discuss how an optical CDM network must
be dimensioned to allow large channel numbers. Using a similar
but more complete numerical model [19]–[21] as described in
the last section we evaluated the number of channels that can be
, if the source spectra and the opsupported with
tical coding filters are optimized. Throughout this section, the
source spectra are assumed to be unpolarized and the different
CDM channels are assumed to be unsynchronized to each other.
For reference (Fig. 6) the optical channel number is evaluated
that may transmit simultaneously with Gaussian source spectra
THz, i.e.,
nm) and for transmitter/receiver
filter combinations MZ/MZ, FP/MZ, and FP/FP. The FSR were
allocated within 10–20 GHz, the inverse FSR being equidistantly distributed within one octave for /MZ combination and
with large prime number ratios in the case FP/FP. The FP filter
in all cases. The FP/ combination
finesse was set to
Fig. 6. Allowed number of simultaneously active channels with a 45-nm-wide
Gaussian spectrum (FP finesse 10).
allows a larger number of channels as compared to the MZ/MZ
system by up to a factor of 2. But obviously with this system design the channel numbers are not large enough for future metro
and access networks, specially at high channel bit rates.
To increase the number of channels we take advantage of the
fact that FP/MZ systems scale with
(cf. last section) implying that the total channel number is increased if the available optical bandwidth is divided into several WDM slices and optical CDM is applied within each of
those slices. As discussed in the last section with /MZ combinations large numbers of codes can be addressed, while the
number of simultaneously active transmitters must be considerably lower. Such systems are attractive for optical packet mode
transmission, taking advantage from statistical multiplexing to
increase the optical channel number. In Fig. 7, nearly rectangular optical spectra are considered (mathematically described
) with
by Super-Gaussian spectra
spectral width of 10 nm and 1.6 nm (200 GHz). In the former
case, 84 codes can be addressed (FSR 10–20 GHz), in the
latter case, there are 27 codes (FSR 5–10 GHz) allowed before
crosstalk increases the BER to
. The number of transmitters that may be active at the same time is limited to 34
and 11, respectively, due to optical noise limits. The 10-nm
slice can be used to transmit optical packets with, e.g., 622 Mb/s
), the 1.6 nm slice, with, e.g., 155 Mb/s
peak rate (
). The channel spacing required in an equivalent
DWDM system would be 190 GHz and 40 GHz, respectively, if
the number of simultaneously active channels is taken as reference. However, in packet mode operation the required channel
spacing for an equivalent DWDM system would be 16 GHz and
. In a packet mode
7.4 GHz for
CDM system either the packet statistics or the network manchannels are
agement must assure that no more than
received at the same time. Violations of these limits might be
corrected applying forward error correction in the receivers.
If the available optical bandwidth is, for example, 50 nm, it
can be divided into five (10 nm) or 31 (1.6 nm) WDM slices.
These system proposals then enable
optical packet mode channels within that bandwidth, not
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Fig. 7. Allowed number of simultaneously active channels in 10 nm wide (top,
84) and 1.6 nm (200 GHz) wide (bottom,
= 27) nearly
rectangular WDM slice.
taking into account guard bands. This is more than an order
of magnitude increase as compared to the system with broad
Gaussian source spectra (see Fig. 6). To receive these many
channels, the WDM slices are demultiplexed using conventional
WDM filters before applying the afore mentioned differential
detection scheme to each CDM channel. The average channel
bit rate amounts to about 29 Mb/s for 155 Mb/s peak rate and to
46 Mb/s for 622 Mb/s peak rate and is almost independent of
the two slice bandwidths considered in Fig. 7. These throughput
values are attractive for access networks. The total system capacity ranges from 12.2 Gb/s to 38.5 Gb/s depending on the
specific configuration. From a networking point of view the
grouping of channels into wavelength bands might have some
added value for the management and reconfiguartion of the network. It also allows the parallel operation of additional Gb/s
DWDM channels for high end users over the same network. The
1.6-nm slice option was selected, because of its compatibility
with the ITU DWDM grid. However, it must be mentioned that
problems are to be expected with the optical power budget in
that case. In the above simulations, it was assumed that the optical source power was 1 mW within the considered bandwidth
before the optical encoding filter at the transmitters.
In the network scenario developed above there is one differential receiver for each optical channel. Since mechanically tunable filters are used, this requires much floor space and will be
an expensive solution as long as the average channel capacity is
well below 100 Mb/s. In this section, we propose an approach
where the number of optical CDM channel is again reduced,
but now each CDM channel carries many optical TDM signals.
The proposed network comprising metropolitan and access fiber
networks is depicted in Fig. 8 [22]. The optical CDM technique
is supposed to be only applied in the upstream direction from
the optical network units (ONU) to the headend, so the discussion is restricted to transmission in that direction. For downstream transmission a combination of DWDM and TDM seems
better suited [22]. There are three cascaded optical multiplexing
techniques in the network. Starting from the headend, there is a
coarse WDM on the primary ring and optical CDM on the secondary ring. The link between both rings is realized by the indicated a coarse WDM coupler. The combined WDM/CDM operation is the same as discussed above, with the only exception
that now the CDM channels are made up of sequential optical
packets from a TDM network (cf. below) and are quasi-continuously active. So the CDM system must be optimized for continuous data transmission in contrast to the situation discussed in
the preceding section. The best performance is achieved, if we
use FP filters both at the transmitters and at the receiver (FP/FP
combination). The finesse is set to
. To reduce crosstalk
and optical noise originating from the overlap of multiple inteand
gers of the filter FSR in terms like
FSR are allocated within the semi-octave 15–20 GHz using ratios of large prime numbers. The number of continuously active
CDM channels that can now be supported within broad WDM
slices is depicted in Fig. 9. Now transmission even at 2.5 Gb/s
channel bit rate is possible with six (10) CDM channels within
10 nm (16 nm) WDM slices. However, the modulation transfer
function of the FP filters is reduced to bandwidths in the order
of 0.75–1.0 GHz leading to signal distortions at 2.5 Gb/s. Using
an electrical decision feedback equalizer (DFE) after detection
these distortions can be compensated. DFEs have been successfully applied to compensate for even larger ratios of electrical
signal bandwidth to equivalent electrical filter bandwidth in dispersive LED transmission [23].
Each optical CDM channel in Fig. 8 carries optical packets
from a TDM PON (passive optical network). The all optical
conversion from TDM to CDM applies a technique that has
been recently demonstrated for bit rates up to 10 Gb/s [24]. It
is based on the optical modulation of the backward ASE (amplified spontaneous emission) output power of an SOA (semiconductor optical amplifier) or an LED by means of an intensity modulated laser diode (Fig. 10). The logic of the modulated
ASE signals is inverted with respect to the input. This is, however, easily reverted in the electrical domain after detection. The
modulated ASE is optically encoded using FP filters as above
and now behaves like the other optical CDM channels discussed
in this paper. Since the laser diode emission wavelengths may
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Fig. 8.
Optical network incorporating different cascaded optical multiplexing techniques (WDM denotes a wavelength selective coupler).
Fig. 9. Allowed number of simultaneously active channels in different slice
widths for optimized FP/FP filter combinations.
vary over a 50-nm wide window without significantly degrading
the conversion efficiency a tight selection of transmitter lasers is
not necessary [24]. We demonstrated the feasibility of the concept by time domain multiplexing 2 4 DFB lasers with 127
bit packets at 622 Mb/s into two CDM channels (8.6 nm spectral width) and achieved factors of 10 [22] (the width of the
guard bands between packets in Fig. 10 can be substantially reduced in practice).
Now the network in Fig. 8 comprises 20 TDM channels per
PON at 2.5 Gb/s peak rate, collected into five optical CDM
channels that are multiplexed into four WDM slices (10 nm
wide Super-Gaussian with 3 nm guard bands between them).
This gives an all optical connection between 400 ONUs and
headend with 100 Mb/s average channel bit rate (including 20%
TDM guard bands between packets). The number of optical
CDM receivers is now reduced to 20, because demultiplexing
of the TDM packets is performed electrically after optical decoding and detection. Although the optical transmission length
between ONU and headend may be up to 100 km the stringent timing requirements of the TDM approach apply only to the
PON section that is assumed to be spread over 20 km. With the
TDM-in-CDM approach the primary and secondary rings can
be managed as is usually done including any kind of protection
switching schemes. This could not be achieved, if the TDM approach was applied throughout the entire network, e.g., within a
common DWDM channel for each secondary ring. In that case
the temporal ranging of the ONUs would have to be changed, if
the optical path length changes due to protection switching requiring a reinitialization of the network after a failure.
The advantage of the proposed system approach lies in the
low cost per channel and the inherent stability and robustness
of the applied multiplexing techniques. At the ONUs almost
unspecified laser diodes can be used and the effort for generation
and encoding/decoding of broadband signals is shared among
20 high-speed ONUs additionally taking advantage of the above
mentioned temperature stability of the CDM technique. WDM
applying 10-nm slices requires only loosely specified optical
filters that are expected to operate in a very stable way.
We will now highlight two issues arising from the broad spectral width of the optical signals: 1) optical amplification and
2) transmission over dispersive fibers.
Since both crosstalk and noise depend critically on the shape
and width of the source spectrum as seen by the receiver, any
deformation of the spectrum during transmission may degrade
the system performance. However, the spectral gain shape of
the optical amplifiers is not as critical as it might seem from
first sight. With the exception of one fluoride-based EDFA we
used conventional silica-based EDFAs with the well-known
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Fig. 10.
All optical TDM-to-CDM conversion (top) and experimental demonstration at 622 Mb/s (bottom).
ASE peak around 1530 nm for low input signals. At high input
10 dBm) the corresponding gain peak vanishes
power (
leaving a sufficiently flat gain response over almost 30 nm
width. If the signal input power was so low that the gain peak
around 1530 nm became too large we added a small piece of
unpumped erbium-doped fiber after the silica-based EDFAs,
thus equalizing the gain shape in a very simple yet effective
way. This measure was accompanied by a 2–3 dB loss in total
optical signal power. With narrowband optical signal spectra
as proposed in the last section the requirement of gain flatness
is more related to power equalization among the WDM slices,
rather than to CDM specific crosstalk and noise issues. So the
impact on system performance is not CDM specific in that case.
In the remainder of this section, we will concentrate on chromatic dispersion induced signal distortions.
The chromatic dispersion of the fiber link has a potential
impact both on the signal and on the noise in systems operating with thermal light sources [25]. However, for broad source
spectra the electrical noise power in the receiver is not significantly affected by the chromatic dispersion, even if transmission
takes place far from the wavelength of zero fiber dispersion [19].
The degradation of the SNR is primarily due to distortions of
the signal shape. In our experiments we used matched lengths
of DCF to restore the optical signal shape before detection.
The question with optical dispersion compensation is,
how exactly the DCF dispersion value must match the SMF
dispersion to enable error free transmission, or in other words,
how large is the length tolerance of optical compensation
schemes. To answer that question we investigated the optical
power penalty (eye opening penalty, EOP) that is induced,
when high-speed thermal light pulses around 1550 nm are
transmitted over uncompensated SMF links. The EOP is the
additional optical power
that is required to maintain a
given BER in the presence of signal distortions. The value
after transmission over fiber length can be estimated
from the relative pulse peak amplitude for a single isolated
“1” (losses neglected) [26]. It is calculated from
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Assuming an electrical Gaussian pulse shape after low pass filtering in the receiver the peak amplitude of the detected op, can be expressed
tical pulse, relative to the input value at
in terms of the parameter
. is a normalized dispersion and yields the asymptotic behavior of the relative broadof NRZ pulses at bit
ening of the width
rate . Here is the fiber dispersion parameter (usually given
in ps/nm km) and
is the FWHM of the optical signal spectrum. is taken to be constant over the entire optical spectrum
considered which is a valid assumption for transmission far from
the wavelength of zero dispersion. The factor is given as
for Gaussian source spectrum
for rectangular source spectrum
where erf( ) denotes the error function. The corresponding
power penalties as calculated from (8) are valid as long as
, or equivalently as long as
Gaussian and rectangular spectra, respectively. After these
values are reached a BER floor will be measured.
In Fig. 11 the calculated penalties are depicted for Gaussian
and rectangular source spectra as function of the normalized dis) the calculated curves are
persion . At small values (
verified by experimental values that were extracted from transMb/s NRZ signals in
mission measurements of
the range around 1550 nm over different SMF lengths using
nm wide Gaussian and
nm wide rectangular optical spectra. These types of spectra correspond to
the situations discussed in the section about the optical CDM
system designs.
For large values, approaching the limits of uncompensated
, the experiments showed
higher penalties than predicted by the simple model. The measured fiber length limits, where a BER floor set in, were somewhat smaller than calculated (around 8 km and 77 km instead
of 9.3 km and 93 km). This may be due to not taking into account electrical noise in the calculations or that the Gaussian
pulse shape approximation is not good enough. Nevertheless,
the model is useful for a quick estimation of the expected system
performance in terms of chromatic dispersion induced degradations.
If the fiber lengths are larger than the above given limits
or if the penalties are too large to be tolerated, an electrical
postcompensation applying a DFE can be used to improve the
performance. Using a two stage DFE a normalized dispersion
km SMF,
nm with Gaussian
spectrum), i.e., beyond the practically measured limit (cf.
Fig. 11), has been successfully compensated with only 4.0 dB
residual penalty [23]. Approaches applying linear electrical filters to compensate the low pass characteristic of the fiber-LED
system have been found not to be as effective as the DFE
approach in that the power penalty easily exceeds 10 dB [27].
In practical system implementations the larger part of the
transmission fiber dispersion should be compensated using appropriate lengths of DCF. The residual dispersion with normal-
Fig. 11. Optical power penalty versus normalized dispersion
BDL for dispersive LED transmission with rectangular and with Gaussian spectrum.
Measurements were performed at 155 Mb/s with 71 nm wide Gaussian and with
8.6 nm wide rectangular spectrum.
ized values up to around
can then either be tolerated
or the induced power penalty may be reduced by applying the
DFE approach. To give a number that is relevant for the proposed systems of the preceding section the theoretical limit for
uncompensated transmission of 2.5 Gb/s signals with 10 nm
rectangular spectra lies at 4.9 km. This is the order of magnitude to which the length of the DCF must be matched to the
SMF length.
The dispersion slope of commercially available DCF is well
enough matched to the slope of the SMF to enable compensation over broad spectral ranges for long fiber links. In 8 155
Mb/s system experiments with about
nm wide spectra
we compensated the dispersion of up to 111 km field installed
SMF [17]. With a dispersion slope of 0.09 ps/nm km for the
ps/nm km) and 0.2 ps/nm km for the DCF
ps/nm km) the maximum differential group delay
due to mismatched dispersion slope alone for the wavelength
nm and
nm wide spectra
components of
amounts to 6.4 ps and 0.7 ps per kilometer SMF, respectively.
So if the dispersion is perfectly compensated at the center wavelength of the spectra, severe signal degradation due to dispersion
slope mismatch would only be measured with signals at 2.5 Gb/s
with 30 nm spectra after about 30 km SMF. In the case of narrow
rectangular spectra as they have been proposed in the system
design section of this paper the residual mismatch in total dispersion is irrelevant even at 2.5 Gb/s over 200 km SMF.
We have proposed and analyzed a hybrid optical multiplexing
approach for metropolitan and access networks. Based on a mix
of wavelength, optical code and time division multiplexing a
very flexible and robust implementation of multiuser optical
networks can be realized. The basic technologies applied have
been demonstrated experimentally. The network was analyzed
using a detailed theoretical approach that takes into account
crosstalk and intensity noise in wavelength sliced optical CDM
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networks. By taking advantage of the fact that optical CDM supports more codes than can be used simultaneously an optical
packet transmission system with more than 400 optical channels was proposed that relies on statistical multiplexing gain.
Alternatively, using a novel optical TDM-to-CDM conversion
technique the implementation of a 400-channel system with average channel bit rates up to 100 Mb/s was discussed. In the
latter approach, the optical packet transmission is realized sequentially within each CDM channel. The applied optical CDM
technology as well as the TDM-to-CDM conversion has been
shown to be very insensitive to poor component specifications
and drift, thus keeping the cost low for components and network
supervision in the physical layer.
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Thomas Pfeiffer, photograph and biography not available at the time of publication.
Jens Kissing, photograph and biography not available at the time of publication.
Jörg-Peter Elbers, photograph and biography not available at the time of publication.
Bernhard Deppisch, photograph and biography not available at the time of
Martin Witte, photograph and biography not available at the time of publication.
Harald Schmuck, photograph and biography not available at the time of publication.
Edgar Voges, photograph and biography not available at the time of publication.
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