Variable Frequency AC Motor Drive Systems

IET Power and Energy Series 8
This book is intended to explain the technical principles involved in
the many AC variable speed drive systems available today. It deals
with all the DC link inverter and direct AC to AC converter systems
that are in commercial use. The principles of AC motors are
considered specifically from the variable frequency point of view,
and this chapter concentrates on the effects of harmonics. The
different types of power semiconductor switches are considered
separately from the drive systems in which they are used.
A total of seven separate and technically different drive systems
are considered in such a way that their principles can be fully
understood and their performance capabilities explained. Square
wave and pulse width modulated DC link inverter systems,
cycloconverters and slip power recovery drives are all included in
this comprehensive book.
This book has been written so that it can be understood by general
engineers, not just by experts in the field. It should therefore be
of great use to any engineer involved with variable speed drives in
any capacity. It should also be of interest to university and college
electrical engineering departments and students.
David Finney, B.Sc., CEng., FIEE, is
division manager and chief engineer,
responsible for large variable speed drive
systems, at the G.E.C. Industrial Controls
plant in Rugby, England. In this position he
is responsible for the development, design
and manufacture of large drive systems
for use in mining, metals, paper, oil, and
chemical industries throughout the world.
He has been technically involved in the
power semiconductor field since 1958,
when thyristors were only just emerging,
and during this time he has worked on all
types of thyristor converters and inverter
drives from a few kilowatts up to 10,000
kW using natural and forced commutation
techniques and operating in square wave
and pulse modulated modes.
He has published a number of articles
and given lectures around the world in his
chosen subject.
Variable Frequency
AC Motor Drive Systems
Variable Frequency
AC Motor Drive Systems
David Finney
Finney
The Institution of Engineering and Technology
www.theiet.org
0 86341 114 2
978-0-86341-114-4
Variable Frequency
AC Motor Drive
Systems
IET Power and Energy Series 8
Series Editors: Prof. A.T. Johns
G. Ratcliff
J.R. Platts
Variable Frequency
AC Motor Drive
Systems
Other volumes in this series:
Power circuit breaker theory and design C.H. Flurscheim (Editor)
Industrial microwave heating A.C. Metaxas and R.J. Meredith
Insulators for high voltages J.S.T. Looms
Variable frequency AC motor drive systems D. Finney
SF6 switchgear H.M. Ryan and G.R. Jones
Conduction and induction heating E.J. Davies
Statistical techniques for high voltage engineering W. Hauschild and
W. Mosch
Volume 14 Uninterruptable power supplies J. Platts and J.D. St Aubyn (Editors)
Volume 15 Digital protection for power systems A.T. Johns and S.K. Salman
Volume 16 Electricity economics and planning T.W. Berrie
Volume 18 Vacuum switchgear A. Greenwood
Volume 19 Electrical safety: a guide to causes and prevention of hazards
J. Maxwell Adams
Volume 21 Electricity distribution network design, 2nd edition E. Lakervi and
E.J. Holmes
Volume 22 Artificial intelligence techniques in power systems K. Warwick, A.O. Ekwue
and R. Aggarwal (Editors)
Volume 24 Power system commissioning and maintenance practice K. Harker
Volume 25 Engineers’ handbook of industrial microwave heating R.J. Meredith
Volume 26 Small electric motors H. Moczala et al.
Volume 27 AC-DC power system analysis J. Arrill and B.C. Smith
Volume 29 High voltage direct current transmission, 2nd edition J. Arrillaga
Volume 30 Flexible AC Transmission Systems (FACTS) Y-H. Song (Editor)
Volume 31 Embedded generation N. Jenkins et al.
Volume 32 High voltage engineering and testing, 2nd edition H.M. Ryan (Editor)
Volume 33 Overvoltage protection of low-voltage systems, revised edition P. Hasse
Volume 34 The lightning flash V. Cooray
Volume 35 Control techniques drives and controls handbook W. Drury (Editor)
Volume 36 Voltage quality in electrical power systems J. Schlabbach et al.
Volume 37 Electrical steels for rotating machines P. Beckley
Volume 38 The electric car: development and future of battery, hybrid and fuel-cell
cars M. Westbrook
Volume 39 Power systems electromagnetic transients simulation J. Arrillaga and
N. Watson
Volume 40 Advances in high voltage engineering M. Haddad and D. Warne
Volume 41 Electrical operation of electrostatic precipitators K. Parker
Volume 43 Thermal power plant simulation and control D. Flynn
Volume 44 Economic evaluation of projects in the electricity supply industry H. Khatib
Volume 45 Propulsion systems for hybrid vehicles J. Miller
Volume 46 Distribution switchgear S. Stewart
Volume 47 Protection of electricity distribution networks, 2nd edition J. Gers and
E. Holmes
Volume 48 Wood pole overhead lines B. Wareing
Volume 49 Electric fuses, 3rd edition A. Wright and G. Newbery
Volume 51 Short circuit currents J. Schlabbach
Volume 905 Power system protection, 4 volumes
Volume 1
Volume 4
Volume 7
Volume 8
Volume 10
Volume 11
Volume 13
Variable Frequency
AC Motor Drive
Systems
David Finney
The Institution of Engineering and Technology
Published by The Institution of Engineering and Technology, London, United Kingdom
First edition © 1988 Peter Peregrinus Ltd
Reprint with new cover © 2006 The Institution of Engineering and Technology
First published 1988
Reprinted 1991, 2006
This publication is copyright under the Berne Convention and the Universal Copyright
Convention. All rights reserved. Apart from any fair dealing for the purposes of research
or private study, or criticism or review, as permitted under the Copyright, Designs and
Patents Act, 1988, this publication may be reproduced, stored or transmitted, in any
form or by any means, only with the prior permission in writing of the publishers, or in
the case of reprographic reproduction in accordance with the terms of licences issued
by the Copyright Licensing Agency. Inquiries concerning reproduction outside those
terms should be sent to the publishers at the undermentioned address:
The Institution of Engineering and Technology
Michael Faraday House
Six Hills Way, Stevenage
Herts, SG1 2AY, United Kingdom
www.theiet.org
While the author and the publishers believe that the information and guidance given
in this work are correct, all parties must rely upon their own skill and judgement when
making use of them. Neither the author nor the publishers assume any liability to
anyone for any loss or damage caused by any error or omission in the work, whether
such error or omission is the result of negligence or any other cause. Any and all such
liability is disclaimed.
The moral rights of the author to be identified as author of this work have been
asserted by him in accordance with the Copyright, Designs and Patents Act 1988.
British Library Cataloguing in Publication Data
Finney, David
Variable frequency AC motor drive systems.
1. Alternating current electric motors 2. Variable speed drives
I. Title II. Series
621.46’2
ISBN (10 digit) 0 86341 114 2
ISBN (13 digit) 978-0-86341-114-4
Printed in the UK by Short Run Press Ltd, Exeter
Reprinted in the UK by Lightning Source UK Ltd, Milton Keynes
Contents
Preface
Page
ix
1
AC motors
1.1 Introduction
1.2 The induction motor
1.2.1 Induction motor principles
1.2.2 The variable frequency induction motor
1.2.3 The equivalent circuit
1.2.4 The vector diagram
1.2.5 Equations and relationships
1.2.6 Examples of calculations
1.3 The synchronous motor
1.3.1 Synchronous motor principles
1.3.2 Equivalent circuits and vector diagrams
1.3.3 Equations and relationships
1.3.4 Examples of calculations
1.4 Harmonics in AC motors
1.4.1 Harmonic power losses
1.4.2 Torque pulsations
1.4.3 Harmonic equivalent circuits
1.5 Motor power losses
1.6 Motor voltages to earth
1
1
2
3
9
14
20
21
26
32
33
36
40
42
45
46
46
47
49
52
2
Power switching devices
2.1 Introduction
2.2 The thyristor
2.2.1 Capabilities and performance
2.2.2 The available thyristors
2.2.3 Thyristors in AC motor drive circuits
2.3 The transistor
2.3.1 Capabilities and performance
2.3.2 The available transistors
2.3.3 Transistors in AC motor drive circuits
54
54
55
57
62
65
73
75
83
86
vi
Contents
2.4
The gate turn off thyristor
2.4.1 Capabilities and performance
2.4.2 The available GTO thyristors
2.4.3 GTO's in AC motor drive circuits
91
93
98
100
3
Power switching circuits
3.1 Introduction
3.2 The 3 phase, naturally commutated bridge
3.2.1 As a rectifier
3.2.2 As an inverter — regeneration
3.2.3 Switch voltages
3.2.4 DC voltage harmonics
3.2.5 AC current harmonics
3.3 The three phase bridge inverter
3.3.1 The voltage source bridge inverter
3.3.2 The current source bridge inverter
3.4 Isolation of electronics
104
104
104
104
111
113
115
115
119
120
124
126
4
The six step voltage source inverter for induction motors
4.1 Introduction
4.2 Principles of operation
4.3 Detailed analysis of the system
4.3.1 Circuit waveforms
4.3.2 Relationships and equations
4.3.3 Examples of calculations
4.4 Practical circuit design considerations
4.4.1 Overcurrent protection
4.4.2 Overvoltage protection
4.4.3 Factors affecting specifications
4.4.4 Circuit variations
4.5 Overall control methods
4.5.1 Supply convertor control
4.5.2 Inverter control
4.5.3 Typical control schemes
4.6 Performance and application
4.6.1 Torque/speed characteristics
4.6.2 Speed control accuracy
4.6.3 Supply power factor and harmonics
131
131
131
135
136
142
148
151
152
154
154
156
158
159
159
160
161
162
164
164
5
The Pulse Width Modulated voltage source inverter for induction
motors
5.1 Introduction
5.2 Principles of operation
5.2.1 Pulse width modulation
5.2.2 The PWM drive system
166
166
166
167
174
Contents
5.3
vii
Detailed analysis of the system
5.3.1 Motor waveforms
5.3.2 Inverter circuit waveforms
5.3.3 Circuit relationships and equations
5.3.4 Examples of calculations
Practical circuit design considerations
5.4.1 Overcurrent protection
5.4.2 Regeneration
5.4.3 Factors affecting specifications
5.4.4 Typical circuit diagram
Overall control methods
Performance and application
5.6.1 Torque/speed characteristics
5.6.2 Efficiency
5.6.3 Supply power factor
5.6.4 Motor and supply harmonics
5.6.5 Accuracy and transient performance
177
177
182
185
188
192
192
193
194
194
196
198
199
199
200
200
201
6
The six step current source inverter drive
6.1 Introduction
6.2 Principles of operation
6.3. Detailed analysis of the system
6.3.1 Circuit waveforms
6.3.2 The motor vector diagram
6.3.3 Circuit relationships and equations
6.3.4 The standard current source inverter circuit
6.3.5 Examples of calculations
6.4 Practical circuit design considerations
6.4.1 Overcurrent protection
6.4.2 Overvoltage protection
6.4.3 Circuit variations
6.4.4 Factors affecting specifications
6.5 Overall control methods
6.6 Performance and application
6.6.1 Torque/speed characteristics
6.6.2 Efficiency
6.6.3 Supply power factor
6.6.4 Torque pulsations
202
202
203
206
207
213
215
219
223
229
229
230
230
231
232
235
235
236
236
236
7
The six step synchro-convertor system for synchronous motors
7.1 Introduction
7.2 Principles of operation
7.2.1 Starting and low speed operation
7.2.2 Normal running conditions
239
239
241
242
245
5.4
5.5
5.6
viii
Contents
7.3
7.4
7.5
7.6
8
123 Reversing and regeneration
7.2.4 Motor excitation
Detailed analysis of the system
7.3.1 Convertor and motor waveforms
7.3.2 Armature reaction
7.3.3 Motor vector diagram
7.3.4 Relationships and equations
7.3.5 Examples of calculations
Practical circuit design considerations
7.4.1 Overcurrent protection
7.4.2 Factors affecting specifications
7.4.3 Circuit variations
Overall control methods
7.5.1 Supply convertor control
7.5.2 Motor convertor control
7.5.3 Excitation control
Performance and application
7.6.1 Torque/speed characteristic
7.6.2 Efficiency
7.6.3 Speed control accuracy
7.6.4 Stability and transient performance
7.6.5 Supply power factor
7.6.6 Torque pulsations
247
248
249
249
251
252
254
257
261
262
263
265
266
268
268
269
269
270
271
272
272
273
274
The current source inverter for the capacitor self-excited induction
motor
275
8.1 Introduction
8.2 Principles of operation
8.2.1 High speed running
8.2.2 Lower speed running
8.3 Detailed analysis of the system
8.3.1 Circuit waveforms
8.3.2 The motor vector diagram
8.3.3 Relationships and equations
8.3.4 Examples of calculations
8.4 Practical circuit design considerations
8.4.1 Protection
8.4.2 Commutation methods
8.4.3 Factors affecting specifications
8.5 Overall control methods
8.5.1 Supply convertor control
8.5.2 Motor convertor control
8.5.3 Motor magnetisation control
8.5.4 Typical overall control scheme
275
275
278
279
281
284
286
288
292
295
295
296
299
301
302
302
303
303
Contents
8.6
9
ix
Performance and application
8.6.1 Motor current waveforms
8.6.2 Torque/speed capability
8.6.3 Supply power factor
305
305
306
307
The cycloconvertor
9.1 Introduction
9.2 Principles of operation
9.2.1 The fundamental principles
9.2.2 3 phase systems
9.2.3 Reversal and regeneration
9.2.4 Supply side conditions
9.3 Detailed analysis of the system
9.3.1 Circuit waveforms
9.3.2 Current reversal
9.3.3 The motor vector diagram
9.3.4 Relationships and equations
9.3.5 Examples of calculations
9.4 Practical circuit design considerations
9.4.1 Overcurrent protection
9.4.2 Convertor polarity switching
9.4.3 Alternative power circuits
9.5 Overall control methods
9.5.1 Firing control
9.5.2 Typical control schemes
9.6 Performance and application
9.6.1 Speed range
9.6.2 Dynamic performance
9.6.3 Supply power factor
9.6.4 Harmonics
308
308
309
309
312
313
315
318
318
325
326
327
331
332
333
334
335
338
339
340
343
343
344
344
346
.
10 The slip energy recovery system for wound rotor induction motors
10.1 Introduction
10.2 Principles of operation
10.3 Detailed analysis of the system
"
10.3.1 Circuit waveforms
10.3.2 The motor equivalent circuit
10.3.3 The motor vector diagram
10.3.4 Circuit equations and relationships
10.3.5 Examples of calculations
10.4 Practical circuit designs
10.4.1 Overcurrent protection
10.4.2 Overvoltage protection
10.4.3 Circuit variations
349
349
350
354
355
357
360
360
364
367
368
369
370
x
Contents
10.5 Overall control methods
10.6 Performance and application
10.6.1 Efficiency
10.6.2 Power factor
10.6.3 Torque capability
10.6.4 Harmonics in the system
370
373
373
374
376
377
Bibliography
380
Index
390
Preface
During recent years there has been a surge of interest in the subject of AC
Variable Frequency Motor Drives and this has been mainly due to the many
technical and financial benefits which can be derived from being able to vary the
speed of a process. The plant can be operated under its optimum condition
whatever its loading and in many cases considerable energy savings can be made
compared to other drive arrangements.
During the same period there has also been considerable technical advance in
the capabilities of such drive systems due mainly to the emergence of high
quality semiconductor power switches and control microprocessors. This has
caused the cost of these drive systems to reduce so that the overall economics
of their application can be favourable in an increasing range of potential uses.
In writing this book my aim has been to explain the technicalities of these
drives in such a way that they can be understood by as wide a range of people
as possible so as to encourage the increasing use of these systems.
It has not been written just for the technical expert in this area of drives but
also for the people who will use, apply and maintain such systems as well as
those who only have a general interest in the subject. I have also included
information which will be of particular interest to the college and university
departments dealing with power electronic equipment and I hope this book
helps them widen the scope of their curriculum to include variable speed drives.
The preparation of this book was greatly assisted by my developing a set of
computer programmes designed to model the individual drive systems. As a
result I have decided to complete the development of these programmes and to
make them available to others.
These programmes model the steady state behaviour of the drive systems and
using them it is possible to:
a) Model any drive, of any size, of any speed range operating at any
voltage level.
b) Operate the computer as though it were the drive, using the keyboard
to input your requirements and observing the drive operation on the
screen.
xii
Preface
c) Establish all the variable parameters of the drive under any condition
of operation. All the supply convertor, motor convertor and motor
currents, voltages and power factors, etc., are available at any
operating speed and torque.
d) Observe the switching sequences of the power circuits while controlling the drive model from the computer keyboard.
e) Obtain printed graph plots of the variation of all the drive parameters
from a printer connected to the computer.
f) Carry out experiments on the drive model under a variety of conditions, as though it was a set of laboratory equipment. It is possible
to start with a simplified system, e.g. neglecting power losses, etc. and
to gradually increase the system complexity until a full practical drive
is being modelled and studied.
These programmes are a very important aid to the full understanding of these
drive systems. Further details can be obtained from ORANGE ENTERPRIZES, 20, BADBY ROAD, DAVENTRY, NORTHANTS. NN11 4AP,
ENGLAND.
I would like to thank all my colleagues at G.E.C. Industrial Controls, Rugby,
for the help they have given me, this book would not have been possible without
their help, specifically I would like to thank Mr. David Martin for much expert
advice. Special thanks are due to my wife, Lesley, for being patient during the
many hours of writing and for the time she spent transferring my untidy
handwriting into our word processor and hence into the typed manuscript.
Acknowledgement is also given to The General Electric Company of England
and to G.E.C. Industrial Controls, Ltd, for permission to publish this book, the
contents of which I learned while in their employ.
May I hope that all readers find this book interesting, informative and
readable.
DAVID FINNEY
DAVENTRY 1987
Chapter 1
AC motors
1.1 Introduction
It is impossible to investigate the operation and performance of the many AC
variable frequency drive systems without first of all considering the motor itself.
It is the motor which carries out the useful mechanical work that is the important end result of all such systems. The aim of the power electronic drive
controller is to obtain the optimum performance from the motor, to obtain the
maximum power from it over as wide a speed range as is required, to achieve
the highest operating efficiency from the motor and to obtain the best dynamic
performance possible. In all cases it is necessary for the motor and controller to
be matched together carefully if this overall optimum performance is to be
achieved. Hence the starting point of this exploration into variable frequency
drives must be the motor, how it works, how it develops torque and how to
understand it when operating as a variable speed drive.
Traditionally variable speed motors have been DC motors and they have
reigned supreme in this field since electricity has been put to practical use.
They are still used for a wide range of applications where the high quality
performance they can produce is needed. However there is an increasing area of
application where the DC motor is unable to satisfy the performance required
or cope with the environment specified. In some cases it is the lack of a
commutator or brushgear which can decide on the use of an AC motor. In
others it is the need for speeds above those achievable with a DC motor. In yet
others it may be the wish to apply a variable speed controller to an existing fixed
speed motor. It may even be the ready availability of an AC motor which is the
deciding factor. Whatever the reason may be, the availability of a wide range of
variable frequency drive systems is leading to a ste? iy increase in the use of AC
variable speed motor drives throughout industry and this trend is clearly going
to continue.
This chapter is not intended for motor designers; it does not go into the
details of winding factors and specific loadings, nor does it deal with tooth
saturation or sub-transient reactance. It is aimed at explaining the motors in
simple terms with particular reference to their use with variable frequency
2
AC motors
controllers; the ways of getting the best out of them and the adverse features of
their performance.
This book as a whole discusses drive systems which are able to be used with
motors that are manufactured in relatively large quantities by a number of
manufacturers. It does not cover special systems which need unusual motors.
Hence this chapter deals with only conventional AC machines with three phase
windings, machines which have been designed for use on standard power
frequency supply networks or which are derived from such machines. This
means cage or wound rotor induction motors and synchronous machines of the
salient pole, cylindrical, slip ring or brushless types.
1.2 The induction motor
The 3 phase AC induction motor is the most widely used motor in industry
today and it has been so since the original decision at the beginning of this
century to standardise on an alternating current transmission system for electric
power. It is a relatively simple motor which only requires power to be connected
to it's stator winding; no auxiliary supplies or independent field excitation
systems are needed. As a result it can be made by rugged and economic methods
and it is found to be extremely reliable even when used in the severe and adverse
environments which are experienced in many industrial applications.
The fact that the motor is self starting and that most can be started just by
direct on line switching is a feature of importance and the fact that it can
continue operating even with significant disturbances on the mains supply adds
to it's in service reliability.
When used on fixed frequency mains power supplies it is basically a fixed
speed motor, the speed only changing slightly even when large changes in load
and torque are applied. It is also capable of accepting high overload levels
without being damaged and without tripping off.
On the whole therefore the induction motor has for a long time been seen as
a real workhorse capable of working hard in the worst of surroundings, under
heavy load conditions and even on poor mains power supplies. As a result it has
been very widely used and a large proportion of the power generated in the
power stations of the world is used to drive the many millions of such motors
in service.
Its widespread use has led to the design being optimised to reduce size,
material and cost and to its availability in a wide range of powers, voltages and
enclosures. Motors from less than 1 KW to more than 15,000 KW have been
made and voltages from 208 volts to 13.8 KV are regularly available. Enclosures
range from open type machines through totally enclosed designs to the extreme
of explosion proof constructions. Induction motors are regularly installed outdoors, exposed to rain and sandstorms and they are even installed at the bottom
of oil wells.
AC motors
3
It is therefore natural that such motors should be considered for operation at
variable speed when suitable variable frequency controllers became available.
See Fig. 1.1.
Fig. 1.1 This shows a typical totally enclosed cage type induction motor, which can be used
fora very wide range of industrial applications. Although they are designed principally
for fixed speed operation they can be used with many of the variable speed systems
described in this book. (G.E.C. Small Machines, Ltd.)
1.2.1 Induction motor principles
The basis of the three phase induction motor is for the stator winding to produce
a continuously rotating field in the iron and air gap and for this to induce
currents in the rotor conductors such as to generate a torque which will make
the rotor turn and allow the electrical power supplied to the stator to be
converted into rotational mechanical energy which can be drawn from the
motor shaft.
The stator
The stator windings can be of various designs but the essence of them all is that
each phase winding occupies two, 60 electrical degree sections of the iron
perifery, these two sections being separated by 180 electrical degrees. The
4
AC motors
3 phase windings are then arranged in sequence as shown in Fig. 1.2, the
sequence of the physical windings corresponding to the sequence of rotation of
the voltage vectors applied to the phases. Fig. 1.3 shows the arrangement of a
typical single layer stator winding to demonstrate how such windings are
arranged in practice. This diagram shows a two pole section of the stator
flattened out for clarity. With more pole pairs in the stator this sequence is
repeated with the coils of each phase usually being connected in series.
A phase
go
ArAolololq/o/
B phase
return
07
-27
c
"o
o
o
j>
a
C phase
return
/ Y
/
60°
\
\
/
/
A
/ / \\
'
VJ
I
\
L
o
o^
o~
B phase
go
C phase
go
A phase-return
Fig. 1.2 Two pole stator winding space allocation
The aim of the stator winding is to produce afieldfluxwhich rotates smoothly
around the air gap so as to induce voltages and currents into the rotor conductors.
If such a winding as shown in Fig. 1.3 is supplied with three phase currents,
displaced by 120 electrical degrees from each other and changing sinusoidally,
then it will produce just two flux poles of opposite polarity which will move
along the winding and hence rotate in the air gap space, at a speed dependent
on the cyclic frequency of the currents.
To understand this very key feature of AC motors it should be appreciated
that the magneto-motive force (MMF) or ampere-turns produced by each phase
winding is trapesoidal in shape, with the magnitude of the MMF being dependent
on the level of currentflowingin the winding. Fig. 1.4 has been drawn to show
AC motors
5
the MMF's produced by the individual phase windings and the total summation
of the three, at one instant in time when the phase currents are as shown. The
three windings therefore produce a single pair of flux poles.
\
N
J
J
A
C2
A1
C-
A+
I
_J
"BI
A2
A-
l
c1
B2
c+
Fig. 1.3 Typical stator winding
Fig. 1.5 demonstrates how the changing levels of currents in the three phases
results in a total MMF waveform which rotates smoothly around the air gap.
You will see that the MMF waveform is not completely sinusoidal and that its
shape varies slightly with time. In practice its shape is also affected by the fact
that the coils are contained in discrete slots and by the particular arrangement
of the coils. However, the principle point is that analysis of these waveforms
show that with sinusoidal currents the fundamental component does not alter
in size and that the harmonic components are relatively small and insignificant.
This rotating MMF waveform causes a corresponding flux waveform in the
iron and air gap and this interacts with the rotor conductors to generate the
necessary torque. Thefluxwaveform is not identical to the MMF waveform due
to the various saturation effects of the iron caused by the slots, and by variations
in the air gap, etc.
6
AC motors
B-
I
A+
C-
B+
A-
C+
B-
I
stator perifery
A phase
M.M.F
B phase
M.M.F
IQ= - 0 . 5
C phase
M.M.F
1 s a5
X
C -
\
total M.M.F
Fig. 1.4 Stator MMF waveform
The rotor
The rotor of an induction motor consists of a set of rotor conductors which may
be connected together as a 3 phase winding similar to the stator as in the case
of a wound rotor with connecting slip rings. In the majority of cases, however,
it just consists of a set of conductors which are all short circuited together at
both ends of the rotor iron core. In this case the three phases can only be
distinguished in the rotor by the pattern of rotor conductor currents.
AC motors
B_
c_
A+
B+
A_
C"V
c+
B_
A+
7
c_
/—
--,-?
+15 degrees
+ 30 degrees
+ 45 degrees
•7"
f 60 degrees
Fig. 1.5 Stator rotating MMF
Torque production
When the rotor is at standstill, the rotating flux caused by the stator induces
voltages in the rotor conductors and as the rotor windings are short circuited
significant currents are caused to flow. In effect it is operating like a short
circuited transformer, the rotor currents being balanced by equal and opposite
stator winding currents, so that the magnetic MMF andfluxis maintained close
to its original value. The rotor currents react with the air gap flux to generate
forces which try to turn the rotor and which try to reduce the induced effects in
8
AC motors
the rotor. Hence the rotor starts to rotate in the same direction as the rotating
flux field.
As the rotor speed increases the rotor conductors cut the rotatingfield(which
is rotating at a constant speed decided by the stator frequency) more slowly and
the result is that the frequency of the rotor currents is reduced. In general, the
rotor frequency is equal to the difference between the speed of rotation of the
stator field and the speed of rotation of the rotor itself.
The value of the voltage induced into the rotor conductors will also depend
on the relative speed between the rotating field and the rotor. This voltage and
the resulting rotor currents will also reduce as the speed increases.
The value of the torque produced is more complex as it will also depend on
the relative phase relationship between the rotor currents and the stator flux. If
the inductive effect in the rotor is significant the current will be delayed in phase
and the torque produced will reduce. When the rotor is at standstill the rotor
frequency is high. Hence the effect of inductance on the rotor is more significant
than it is when running at high speed when the rotor frequency is very low.
Finally in this brief review of basic principles it should be noted that if the
motor eventually managed to rotate at the same speed as the rotatingfield,then
nothing would be induced into the rotor and no torque would be generated.
Hence the motor never operates in this state: the nearest condition is on a light
load when the speed difference is very small.
Operation on a fixed voltage and frequency supply
The curves of Fig. 1.6 are typical of present day motor designs and these show
the performance obtainable when the motor is connected to a fixed frequency
fixed voltage supply network. The critical point on these curves is the peak value
of the torque curve. At speeds above this the rotor inductance is relatively
insignificant and the value of torque generated per amp is high. At speeds less
than this the inductance has the dominating effect causing the torque to reduce
as the speed drops. In this particular machine special steps have been taken to
make the torque generated at standstill be sufficiently high that the motor will
be able to self start against a significant load. Skin effect has been used to cause
the rotor resistance to increase at low speeds, so preventing the rotor inductance
being allowed to dominate and reduce the torque. This principle is used in the
majority of cage motors.
The curves show very clearly that the conditions to the right of the peak
torque point are very much better than at other speeds. The efficiency is high,
the power factor is high, the torque per amp is high. In addition the speed torque
curve indicates that stable operation is possible because an increase in torque
corresponds to a slight reduction in speed. For these reasons the induction
motor is always used near to the maximum synchronous speed with the actual
speed of operation being dependent on the torque demanded by the load.
Although the rotor always rotates at a slower speed than the rotatingfield(in
normal motoring operation) the magneticfieldsproduced by both the stator and
AC motors
rotor currents rotate at exactly the same speeds in the air gap. The rotor always
produces a rotating field which rotates at rotor frequency with respect to the
rotor. The sum of the rotor speed and rotor frequency must always equal the
speed of rotation of the stator field, i.e. the fields are always in synchronism
whatever the speed of the rotor.
500KW 4 pole
3300 volt, 3phase, 50 hertz
motor
10,000 — 1000 —
tres
efficiency
/
1/ [
80 -
E
ton
/
90
-
tc
100 —
•
S
o
%
70
8.
60
5,000 -
500 —
50
•
400 -
40
2. -
300
30
^
200
20
100 —
10
'
S.
tore
/
\
/
'
\_ \ / />7
//
1
- >v
0
powe^/ X ^
factor
\.
rotor
/
lib
per unit
50
05
1\
\
>
><
frequency
percent
I
\
—
/
1,000 —
torque
power
losses
>v
speed
100
slip
b
Fig. 1.6 Fixed frequency curves of an induction motor
From an electrical point of view therefore the motor operates very much like
a transformer in that the rotor current is balanced by an equal and opposite
stator current. The stator current consists of two components, the direct reflection of the rotor current and the necessary value of magnetising current to
generate the core flux. This magnetising current is much larger than one would
get in a transformer due to the relatively large air gap between the rotor and
stator through which the magnetic field has to pass.
1.2.2 The induction motor as a variable frequency motor
The above assessment of the motor when supplied from afixedfrequency mains
supply still applies when sinusoidal voltages of any frequency are applied to the
10
AC motors
motor, as long as the conditions are such that the air gap flux remains at a
similar value. This means that when such a motor is supplied from a variable
frequency, variable voltage supply there can be an infinite number of sets of
curves like those of Fig. 1.6, one for each of the possible supply frequencies.
Fig. 1.7 shows a typical sequence of torque curves which can be obtained from
such a motor when it is supplied with different frequencies with the applied
voltage being altered in proportion to the frequency. The important result is that
it is possible to always operate in the area to the right of the peak torque point,
i.e. in the area of maximum efficiency, maximum torque per amp and inherent
stable operation, whatever speed the motor happens to be running at. With such
a variable supply, therefore, the motor can always be operated under its most
advantageous conditions at any speed from standstill up, and the regions of high
currents, low torques and low efficiency can be ignored. These conclusions assume
that the supply used to feed the motor is of similar nature to the mains power
supply, i.e. the voltage waveforms are sinusoidal and the source impedance of
the supply is low. In practice most inverter systems, as this book will be
explaining later, are not equivalent to mains sources and the capabilities and
method of operation of the inverter may prevent the motor being used over the
whole of its potential range.
•
" 1
\
30
20
10
40
50
hertz
hertz
l
i
speed
i
i
*•
Fig. 1.7 Torque/speed curves at variable frequency
Another very important point to get clear at the start is that most static
inverter systems used with induction motors allow almost all the parameters of
circuit operation to be chosen at will,
i.e. frequency
voltage level
current level
AC motors
11
can usually all be altered independently by the controller in order to achieve the
optimum point of operation for the total system. Let us therefore consider these
variables in turn.
Frequency
Variation of the frequency supplied to the motor will alter the speed of rotation
of the stator rotating field and hence the synchronous speed of the motor. As
torque is generated by the speed difference between the rotor and the rotating
field the motor speed will normally be slightly slower than the synchronous
speed. However, the motor can always be made to run at the exact speed
required by applying an appropriate frequency slightly higher than that corresponding to the desired speed. Now therefore it is possible to compensate for the
slight speed drop inherent in the motor so that it can be accurately speed
controlled.
Clearly the motor is not now limited to the mains frequency of 50 or 60 hertz.
As long as it is mechanically capable of operating at the higher speeds there is
no reason why 69.77 hertz should not be used.
One consequence of this increased flexibility of operation is that a motor
which was designed to operate on afixedfrequency mains supply may well now
be used over a wide range of speed. The user has to take account of the
consequences of doing this and one of the most important is that the motor
cooling may be dependent on the rotor speed and hence at low speed it may not
be capable of handling the same levels of current and torque as it can at high
speed.
An additional advantage of the ability to alter the frequency at will is that it
is always possible to reduce the applied frequency so that the motor is running
supersynchronously and as a result is capable of regenerating energy from the
motor back into the inverter supply system. The motor can therefore be braked
under full control as long as the inverter supply is capable of absorbing the
energy from the motor and load.
Voltage
As with all magnetic windings and circuits the stator winding of an induction
motor operates by inducing a voltage within it (due to the core flux) which is
approximately equal and opposite to the applied voltage. The current flows
in the winding due to the small difference between the applied and induced
voltages, limited only by the winding impedances. The stator winding therefore
obeys the normal magnetic circuit laws which state that induced voltage is
proportional to:
flux x frequency x number of turns
and hence if an optimum and constant value offluxis to be maintained then the
induced voltage will have to be varied in proportion to the frequency. As, in
general, the supply voltage is usually only slightly higher than the induced
12
AC motors
voltage this means the supply voltage would normally be increased with the
frequency.
However, as I have said before, this does not have to be the case. The level
of voltage and therefore flux can easily be altered if it is advantageous. An
increase in the flux level will mean that more torque can be generated and the
only limitations to the use of higher flux levels are the higher iron losses and the
higher magnetising current which result.
At first sight the reduction of flux by reducing the applied voltage would not
seem to be worthwhile but it can be used effectively for two purposes. If the
motor is being used for significant periods of time at light load the iron losses
in the motor can be reduced significantly if the flux is reduced. The more
frequent reason for allowing the flux to reduce is to enable higher speeds and
frequencies to be used without having to apply higher than rated supply voltage,
and therefore insulation stress to the motor. Operation at frequencies of 50 per
cent above the rated value can usually be used at these reduced flux levels.
Reduction in the applied voltage andfluxis also used as a means of protection
for the motor and drive system. If excessive current conditions are experienced
for any reason (such as the motor stalling) the ability to reduce the voltage
enables damaging currents to be avoided.
Fig. 1.8 shows the basic relationships associated with variation of flux. Due
to saturation the magnetising current has to increase by a larger factor than the
increase in flux. The torque generated will increase in proportion to the flux if
the torque component of current is kept constant.
Fig. 1.8 Flux relationships
Current
The torque developed in an induction motor is the result of interaction between
the air gapfluxand the currents in the rotor conductors. During operation near
AC motors
13
to synchronous speed with low levels of slip the power factor of the rotor current
is high and the torque developed is almost proportional to the current flowing.
Therefore control over the rotor current provides direct control of the motor
torque under optimum low slip operating conditions.
The stator current contains an equivalent current to the rotor current as
dictated by the turns ratio between rotor and stator, but it also contains a
magnetising component which causes the MMF and hence the flux in the iron
circuit. For the purposes of our study the iron circuit is unaffected by the
rotation of the rotor and by the currents flowing in the rotor and hence it is
reasonable to consider that the magnetising current requirement of the motor
is almost independent of the speed of the motor, i.e. the MMF required to
produce the rated flux is the same whatever the speed the motor happens to be
running at.
However the effective impedance of the magnetic circuit does change with
applied frequency and as mentioned before it is necessary to alter the voltage in
proportion to the frequency if the required constant magnetising current and
flux is to be obtained.
In general therefore, control over the current flowing in an induction motor
will control the level of torque generated as long as low slip levels are maintained, i.e. the applied frequency corresponds closely to the speed of the rotor.
per unit
flux
t
p ratio at 1 p.u. flux
12
/
/
o
g
0.8
I
1
0-6
applied stator v
"6
frequency — * •
Fig. 1.9 Voltage variationwith frequency
Variable frequency characteristics
When used under variable frequency sinusoidal conditions with the control
arranged so that operation at low slip values is guaranteed, the most important
characteristic curves are those shown in Figs. 1.9, 1.10 and 1.11.
14
AC motors
Fig. 1.9 shows the relationship between applied voltage and frequency
necessary to achieve the required values of air gap flux. Over most of the range
a constant value of applied voltage divided by frequency will guarantee a
constant flux. At low speeds the voltage drop in the stator resistance becomes
more significant in relation to the applied voltage and a higher value of applied
voltage is necessary to ensure the correct flux level.
The relationship between torque and current is almost independent of
frequency and Fig. 1.10 applies to any frequency and motor speed. It shows that
the torque and current have a linear relationship if the flux level is maintained
constant. The initial value of current required before any torque is generated
shows the magnetising requirement.
per unit
flux
•
-
I
1
b*
B
0-8
0-6
10
12
j
i / /
i /ft'
/
/
y
/
at any
frequency
.
.
.
I
torque •
Fig. 1.10 Torque/current curves
The final set of curves which define an induction motor's variable frequency
capabilities is the torque against slip speed curves of Fig. 1.11. Slip speed is the
speed difference between the rotor and the stator rotating field in, for example,
RPM, this makes these curves again apply at any frequency. As shown, the
torque or slip speed are directly related but reduction of flux causes the slip
speed for a particular value of torque to increase.
1.2.3 The equivalent circuit of an induction motor
The understanding of the electrical operation of this motor under all conditions
of operation is best achieved by developing an equivalent circuit which can
AC motors
15
fully describe the way it works. From this an appropriate vector diagram and
a set of equations which define its operation can be deduced to enable the
performance of a motor to be estimated and assessed.
As the magnetic fields produced by both the stator and rotor always rotate
at the same speeds in the air gap it is possible to represent the windings as a
transformer with stator turns Tl and the rotor as the secondary with T2 turns.
However the stator winding operates at the supply frequency Fl while the rotor
operates at the slip frequency, dictated by the difference between the speed of
the rotor and the speed of the stator rotating field.
per unit flux
/\2
\
at any frequency
/
/
torque
t
/
/
:
0-8
/
/
0-6
:
/
/
/
—^ — - ^
0-4
^
1
slip speed -R.P.M
Fig. 1.11 Torque Islip speed curves
Fig. 1.12(a) shows the equivalent of one phase of an induction motor drawn
on this basis, in this:
R2 is the actual rotor resistance per phase,
L2 is the actual rotor leakage inductance per phase,
Rl is the stator resistance per phase,
LI is the stator leakage inductance per phase.
The losses in the magnetic circuit, the iron losses, are represented by
the presence of the resistor RL and the magnetising impedance dictating the
magnetising current is shown connected across the transformer primary.
In this figure the rotor frequency F2 is given by:
F2 = SI x Fl, where SI is the per unit slip.
16
AC motors
and E2 is related to the stator induced voltage El by:
E2 = SI x El x T2/T1.
If the motor is a wound rotor slip ring one then the values of the rotor
resistance and inductance, R2 and L2 must include the external circuit connected
to the slip rings.
stator
resistance ^
T1
turns
R ^ stator
Inductance
V1
R
F1
frequency
.
11
fc
iron
loss
R1
L1
r - —i
r\^\n
I
J
magnetising
impedance E1
v~
T2
L2
I
R2
t
frequency
F
F2
I
T1
T1
mag
V1
E1;
frequency
F1
11
R1
J
L1
12
IL
V1
;
L2'
*—almag
1
Z
M
X2'=2xfTxFxL2'
E:i
'si
frequency
F1
J
V
\
Fig. 1.12 Induction motor single phase equivalent circuits
Although this equivalent circuit can be used to assess performance it is not
easy due to the two different frequencies. So it is normal to simplify the circuit
further by referring all the rotor parameters to the stator. This can be done in
two stages as shown in Figs. 1.12(b) and 1.12(c). The first stage is to alter the
values to allow for the difference in turns ratio so that stator and rotor voltages
AC motors
17
can be more directly related. This is shown in Fig. 1.12(b) where R2' is now the
rotor resistance value as referred to the stator and L2' is the equivalent of L2.
If we neglect the detailed effects of the windings which may mean the coupling
between them is not ideal then approximately R2' = R2 x (T1/T2)2 and
L2' = L2 x (T1/T2)2 and E2' is now equal to El x SI.
The second stage of simplification is to dispense with the transformer representation altogether by dividing all the rotor parameters by the secondary to
primary voltage ratio, the slip SI.
This results in Fig. 1.12(c) where the rotor frequency is now equal to Fl and
the rotor voltage El.
This is the traditional equivalent circuit of an induction motor and the
necessary referred rotor parameters are usually available from the manufacturers
to match this figure in any specific case.
In this figure:
(a) Imag represents the stator magnetizing current required to produce
the necessary flux so as to generate the correct value of back emf El
in the stator winding.
(b) (II) 2 x RL represents the iron loss.
(c) (II)2 x Rl equals the stator copper loss.
(d) The total power passed across the air gap is represented by
(I2)2 x R27S1.
(e) The rotor copper loss is equal to (I2)2 x R2'.
(f) The total mechanical output to the rotor shaft (including the friction
and windage losses) is therefore given by the difference between (d)
and (e) and this equals.
(I2)2 x R2'((l - S1)/S1)
A couple of further points should be noted if this equivalent circuit is going
to be truly representative of all variable frequency conditions.
1) In present day standard cage type motors it is normal to include deep
rotor bars whose resistance changes with rotor frequency due to skin
effects so that its resistance at standstill is higher than at running
speed when the rotor frequency is very low. Standstill resistance
values of four times those at running speed are not unusual. The rotor
inductance may also vary with rotor frequency. Although this information may be available and required in order to study operation at
high slip, it is not needed for variable frequency operation where the
slip is normally very low. For variable frequency operation R2' and
L2' can be the low frequency equivalent values.
2) The magnetising current Imag is not usually linear with induced
voltage, as motors are now designed to run with the iron flux density
close to saturation. The curve of Fig. 1.13 is typical and will be
referred to later.
Table 1.1 shows a typical set of parameters for standard cage motors being
supplied at the present time, these are not intended to be particularly precise but
Table 1.1 Typical equivalent circuit parameters of induction motors
Motor
power
No. of poles
STAR/
DELTA
KW
0-75
40
150
300
300
550
1100
1320
25-0
550
1320
500
1000
1000
1500
1500
2060
4
4
2
4
6
4
4
4
4
4
4
4
4
6
6
8
4
STAR
STAR
DELTA
DELTA
DELTA
DELTA
DELTA
DELTA
STAR
STAR
STAR
STAR
STAR
STAR
STAR
STAR
STAR
Line
volts
Supply
frequency
Rl
LI
R2'
L2'
Volts AC
Hz
OHMS
mH
OHMS
mH
OHMS OHMS
415
415
415
400
400
415
415
415
660
660
660
3300
3300
3300
3300
3300
3300
50
50
50
50
50
50
50
50
50
50
50
50
50
50
50
50
50
15-4
1-76
0-72
0-28
0-27
012
004
003
0-39
015
004
0-26
0104
0100
•059
•047
•039
47-4
90
7-3
3-7
40
2-1
0-83
0-8
1-2
2-5
10
6-8
2-9
41
2-7
3-6
1-68
10-2
118
0-77
0-28
0-27
010
004
003
0-37
012
0-035
0-26
0105
0-087
•048
•064
•041
38-2
8-9
9-5
5-9
6-5
3-8
1-6
1-6
1-3
21
10
8-0
3-7
3-2
2-3
2-8
2-2
1870
493
964
367
310
456
233
219
427
320
250
2123
1139
1125
833
541
725
RL
Xm
232
500
81-8
380
251
21 6
10-7
8-9
27-1
19-3
10-2
62-3
31-6
32-9
23-2
191
191
Rated
slip
Per unit
•067
•036
•027
•021
•020
•012
•009
•008
•024
•019
•013
•014
•Oil
•010
•008
•Oil
•009
1
AC motors
19
are just to give some idea of the relationship between these parameters in
normally designed motors. The information is based on that kindly supplied to
me by G.E.C. Machines Coy. Ltd. of Blackheath and Rugby, England.
Isat
Vsat
—
~l
I
I
XI
magnetising current
Fig. 1.13 Magnetising saturation curve
Further simplification
Calculations based on this equivalent circuit can still be more than is necessary
for some uses and it is possible to deduce a further approximate equivalent
circuit which can be useful in these variable frequency applications where the
slip is always low.
The approximation can be made by taking account of the following:
1) Referring to Fig. 1.12(c) as the slip is always low the magnitude of the
rotor resistance component R2'/S1 is always much larger than XT and
XT can be ignored without large errors being introduced.
2) Over the majority of the load and frequency range, the voltage drops
in the stator resistance and inductance are small compared to the
value of El and VI and they can be ignored if you are only looking
for general trends in performance.
3) In a similar way the iron losses are usually relatively small and can be
ignored if-simple assessment is being made.
The result is the simplified equivalent circuit of Fig. 1.14.
20
AC motors
h
»
magnetising
impedance
Xm
R2
7s,
Fig. 1.14 Simple equivalent circuit of an induction motor
1.2.4 The vector diagram
The vector diagram has been found to be a useful concept in the understanding
of alternating current systems, where the electrical quantities are all varying
sinusoidally. In this book the normal conventions of these diagrams will be
observed, i.e.
(a) The vectors will be assumed to be rotating in an anticlockwise
direction.
(b) Flux vectors will be shown leading the voltage they induce by 90
electrical degrees, i.e. in quadrature. It should be noted that the flux
vector on a single phase motor vector diagram is not a single vector
but it is the result of all 3 phase MMFs.
(c) The magnetising current causing the flux is shown in phase with the
flux vector.
(d) Voltage drops in resistance are shown in antiphase with the current.
(e) Voltage drops in inductance are shown lagging the current by 90
electrical degrees.
Bearing these in mind and the fact that we wish to appreciate the operation of
the motor over a wide range of frequency, supply voltage and current the vector
Fig. 1.15 has been drawn.
The flux vector Flm is shown horizontally and this induces a voltage El into
the stator winding which is seen by the supply as being a back emf of — El as
shown. The rotor current vector 12 is shown lagging —El due to the effect of
the rotor inductance L2'.
The magnetising current Imag is in phase with thefluxvector and the iron loss
current IL is in phase with the — El. Adding these vectorially to 12 gives II, the
stator current.
This stator current causes voltage drops in Rl and LI and the difference
between the applied voltage VI and the induced voltage — El is II x Rl +
II x XI as shown.
AC motors
21
This diagram includes all the basic parameters and can be used to deduce
performance over a wide range of operating conditions with a reasonable degree
of accuracy.
V1
Mm
t El
Fig. 1.15 Induction motor vector diagram
Simplified vector diagram
Referring back to the simplified equivalent circuit, the same approximations can
be used to derive the vector diagram corresponding to Fig. 1.14. In practice,
angle An2 is small due to the low slip operating conditions and it can be
approximated to zero. Also the angle De is small over most of the range.
Fig. 1.16 shows the simplified vector diagram corresponding to the simplified
equivalent circuit of Fig. 1.14.
22
AC motors
V=-E1
11
12
Flm
Mm
I mag
Fig. 1.16 Simple vector diagram
The circle diagram
Because under variable frequency control the slip value is. normally kept small
the traditional circle diagram concept is not so important. However, I have
shown the circle locus of II in Fig. 1.15 as the load torque is changed as the line
ABC at a specific frequency. At zero load the II vector would be OA with a
typical rated load condition being OB.
The vector diagram as the frequency changes
When used in variable frequency motor drives it is usual to maintain the flux
Flm constant over the whole frequency range by altering the applied voltage
with frequency. The result is that the only serious factor to change on the vector
diagram Fig. 1.15 is the relative magnitudes of the voltage vectors in relation to
the current vectors. The voltage vector VI and — El reduce with frequency but
all the vectors related to currents remain almost the same at the same levels of
load torque.
The reason for this is that as the frequency is reduced the rotor reactance, X2
reduces and as at the same torque the slip increases, the rotor relationship
remains unchanged so that angle An2 is not seriously altered. The only other
factor is that IL reduces with frequency so that at low frequency the locus of II
will be very little different to the original ABC locus.
1.2,5 Induction motor equations and relationships
The steady state calculation of the motor currents and voltages can best be
achieved by writing down the equations related to the equivalent circuit and
vector diagram and Figs. 1.12(c) and 1.15 will be used in the following.
AC motors
23
The most straightforward approach is to initially make an assumption of a
constant rated value of air gap flux being used at all times because this is the
condition in which the majority of motors are operated when used with variable
speed drives. An alternative, more rigorous solution allowing assessment of any
flux level is given later.
Equations at constant flux
Under constant flux conditions the value of the induced voltage El will always
be directly proportional to the applied frequency. Also the slip speed will be
almost linearly related to the torque. If the slip at rated torque, Tr and rated
frequency Fr is given as Sir then the slip speed under this condition is given by:
slip speed = Sir x 120 x Fr/P RPM
where P is the number of poles on the motor, and therefore the slip speed under
any particular torque condition T is given by:
SS = T/Tr x (Sir x 120 x Fr/P)
and therefore the slip SI at any frequency F will be given by:
SI = slip speed divided by synchronous speed
T
Sir x 120Fr/P
f r X 120 x F/P
SI = Sir x I- x §
0)
Z2 = J(2 x n x F x L2')2 + (R27S1)2
(2)
Ir
r
It is now possible to solve the rotor circuit under any condition. The rotor
impedance Z2 is given by
as the flux is assumed to be constant then
m
T
_ Er
" Fr
where Er is the rated induced voltage. Therefore
El = Er x F/Fr
(3)
The rotor current is then found from
12 = E1/Z2
(4)
The power factor angle of the rotor current An2 can be found from
TAXT/A O\
2 X 71 X F X L2' X SI
TAN(An2) = —
—
24
AC motors
Therefore
If thefluxis assumed constant then the value of the magnetising current will be
the same for all operating conditions.
The loss current IL will vary with the value of El
IL = E1/RL
(6)
The vertical component of II will therefore be given by
I ^
= 12 x COS (An2) + IL
(7)
and the reactive component by
Ireact = 12 x SIN(An2) + Imag
(8)
Therefore
II = Viewer) 2 + (Ircac,)2
(9)
Bnl = ATN(Ipower/Ireact)
(10)
and
Finally VI can be found by adding the voltage drops in Rl and LI as follows:
The vertical component of VI will be given by
V ^
= El + II x XI x COS (Bnl) + II x Rl x SIN (Bnl)
and the reactive component
Vreact = II x XI x SIN (Bnl) - II x Rl x COS (Bnl)
and the angle
(11)
(12)
y w ) 2 + (vreact)2
De = ATN(Vreact/Vpower)
(14)
And therefore from this the motor input power factor equals
Pfm = SIN (Bnl - De)
(15)
The electrical power losses in this phase of the motor are given by:
power loss = (II)2 x Rl + (I2)2 x R 2 + (E1)2/RL
(16)
and the total power loss for the motor is three times this value. The speed of the
motor is given by:
S = (1 - SI) x (120 x F/P) RPM
(17)
AC motors
25
The mechanical power out of the motor is given by:
Power out = 2 x n x S x T/60 watts.
(18)
If the torque is in Newton metres and the speed in RPM.
Calculations for any conditions
Calculations for any conditions of applied voltage, induced voltage, slip and
torque are best carried out by computer and I give below a listing of a BASIC
computer program which enables all the motor parameters to be assessed with
any of four sets of input conditions.
7000 REM Computer calculation of induction motor from the equivalent circuit
7010 REM Calculations from the VECTOR DIAGRAM
7020 IF SI=1 THEN LET SI=.9999
7030 LET X2=2* PI *F*L2/1000
7040 LETZ2=((R2/SI)f2+(X2)f2)*.5
7050 LET An2= ATN ((X2*SI)/R2)
7060 LET S=Nss*F/Fr*(1 -SI)
7070 LET I2=E1/Z2
7080 LET lmag=-.75*lsat* LN (1-(E1*Fr)/(F*Vsat))
7090 LET Ireact=lmag+I2* SIN (An2)
7100 LET RL=RLr*F*4/(3*Fr+F)
7110 LET IL=E1/RL
7120 LET lpower=IL+l2* COS (An2)
7130 LET Bn1 = ATN (I power/1 react)
7140 LET I1=((lpower)*2+(lreact)~2)\5
7150 LETX1=2* PI *F*L1/1000
7160 LET Vreact=l1*X1* SIN (Bn1)-I1*R1* COS (Bn1)
7170 IF Vreact<0 THEN LET Vreact=-Vreact: LET K 1 = - 1 : GO TO 7190
7180 LET K1=1
7190 LET Vpower=E1+M*X1* COS (Bn1) + I1*R1* SIN (Bn1)
7200 LET V1 = ((Vpowerf 2+ (Vreactf 2)\5
7210 IF K2=0 THEN GO TO 7250
7220 IF ABS (V1 -V10) *= .005*V10 THEN GO TO 7250
7230 LET E1=E1*(1-(V1-V10)/(V10))
7240 GO TO 7000
7250 LET De= ATN (Vreact/Vpower)
7260 LET Loss=(Hf2*R1+(l2f2*R2+((E1)"2)/RL
7270 LET Losst=3*Loss
7280 LET Pin=3*((l1)"2*R1+(l2)"2*(R2)/SI + ((E1)*2)/RL)
7290 LET Pout=Pin-Losst
7300 LET Torque=((Pout)*60)/(2* PI *S)
7310 IF K3=0 THEN GO TO 7350
7320 IF ABS (Torque-T10) <*= .01*T10 THEN GO TO 7350
7330 LET SI=SI*(1 -(Torque-T10)/(T10))
7340 GO TO 7000
7350 LET Efm=100*(Pout)/Pin
7360 LET Pfm= SIN (Bn1-KUDe)
7370 IF V1 >Vs AND K2=0 THEN LET K2=1: LET V10=Vs: GO SUB 7000: LET K2=0
7500 RETURN
8000 REM end of Subroutine
26
AC motors
10
100
1000
motor power KW
Fig. 1.17 Rated efficiencies of standard totally enclosed three phase induction motors (up to
660 volts, 50/60 hertz)
1) If K2 is set to zero the calculation will work out all the parameters if
the following three are set prior to the calculation
Induced voltage El
Slip
SI
Frequency
F
2) If K2 is set to 1 with K3 set to zero the subroutine will work out all
the parameters if
An applied voltage VI0
Slip
SI
Frequency
F are set beforehand.
3) If K2 is zero and K3 is 1 it is necessary to initially input
Induced voltage
El
Frequency
F
Torque required
T10
An initial value of slip SI
in this case the programme will alter the slip until the required torque
value is obtained.
AC motors
27
100
No. of
poles
90
2
80
70
10
100
1000
motor power - KW
Fig. 1.18 Rated power factors of standard totally enclosed three phase induction motors (up
to 660 volts, 50/60 hertz)
4) With K2 and K3 equal to 1 the programme will again adjust the slip
to achieve your requested torque but it will, at the same time, work
from a value of terminal voltage. You need to input
Required applied voltage V10
Frequency
F
Required torque
T10
An initial value of slip
SI
You will note from line 7050 that in this calculation the magnetising current
is calculated from an equation.
This is the equation of the graph drawn in Fig. 1.13 and from this either Imag
or El can be obtained, from
Imag =
El
- - 7 5 x Isat x LN{1 - (El x Fr)/(F x Vsat)}
(19)
2-71(-133xImag/I-t)}
(20)
= Vsat x - ^ x {1 -
28
AC motors
1.2.6 Examples of induction motor calculations
1) Frequencies and speeds
Question
A 20 KW 4 pole induction motor is designed for operation off a 460 volt 60 hertz
3 phase supply system and when operating at full power on this supply it runs
at 1770 PM. What supply frequency will be needed to make this motor run at
1355 RPM, while delivering a shaft power of 12 KW. Assume operation at rated
flux.
Answer
The synchronous speed at 60 hertz operation will be
=
120 x 60/4
=
1800 RPM
Therefore slip speed at rated power
=
1800 - 1770 = 30 RPM
Torque is proportional to power divided by speed, therefore
Torque (1355) _
Torque (1770)
H
1770
20 X 1355
as the motor is operating at constant flux then slip speed will be proportional
to torque. Therefore slip speed at -784 x rated torque
=
-784 x 30 RPM
= 23-51 RPM
In order to run at 1355 RPM the applied frequency must correspond to a
synchronous speed of
1355 + 23-51
=
1378-51
Therefore
1378-51 = 120 x F/4
F
=
1378-51
4
= 45-95 hertz.
AC motors
29
2) Currents and voltages
Question
If the above motor is assumed to be supplied from a sinusoidal variable
frequency source approximately what voltage and current will need to be
supplied to it when running at the 1355 RPM, 12KW condition if the rated
power factor of the motor is 0*84. Assume a motor efficiency of 86 per cent
under both conditions.
Answer
Refer to the simplified equivalent circuit of Fig. 1.14 and the vector diagram of
Fig. 1.16.
Under rated operating conditions the applied voltage is 460 volts, 60 hertz.
At 45-95 hertz the applied voltage will be approximately
= 460 x 45-95/60 = 352-3 volts line
Under rated operating conditions the input KW to the motor will be given by:
Power in = 20/-86 = 23-3 KW
As the power factor under this condition is 0-84 then the KVAR supplied at
rated load will be
KVAR in = 23-3/-84 x SIN (ACS -84)
=
15 KVAR
This is mainly magnetising current and it will remain approximately a constant indicator of magnetising current. Magnetising current under rated load
conditions will be given by
Imag =
=
15 x 1000/(460 x
18-8 amps line
Under the 12KW output condition the power into the motor is
Power in =
12/86 =
13-95KW
and this is equivalent to a power current of
Wer =
13-95 x 1000/(352-3 x ^3)
= 22-87 amps line
Therefore the input line current will be given by:
II = VCW) 2 + (Imag)2
= V(22-87)2 + (18-8)2
= 29-6 amps line.
30
AC motors
3) Equivalent circuit calculations
Question
If the equivalent circuit of one phase of a 3 phase induction motor is as shown
in Fig. 1.19 find the following when a sinusoidal voltage at a frequency of
20 hertz is applied to the motor so that the induced voltage per phase is 150 volts
and the motor operates at a slip of 002 per unit.
Determine the input phase current to the motor,
the shaft torque in Newton metres,
the total electrical power losses in the motor.
R1=0.2A./>.
415 volts/phase
50Hz
:O.23O
L2=8.5mH R2-
L1=3.7mH
R8500n-
r
[I
Vsat = 450V
Isat = 6 -5 amps
2 pole, 30 KW induction motor
Fig. 1.19 Motor equivalent circuit
Answers
Under these conditions the rotor referred impedance can be found from
equation (2)
Z2 =
v /(0-23/-02)
2
+ (2 x n x 20 x 8-5/1000)2
=
7(132-25 + 114)
=
11-55 ohms.
Therefore from equation (4)
12 =
150/11-55 =
13 amps.
From equation (19)
Imag =
- - 7 5 x 6-5 x LN{1 - (150 x 50)/(20 x 450)}
= 8-7 amps.
From equation (6)
IL =
150/500 = 0-3 amps.
From equation (5)
An2 = 5-3 degrees.
AC motors
31
Therefore from equation (7), (8) and (9)
Ipower =
13 X COS (5'3) + 0-3
=
13-24
Ireact =
13 x SIN (5-3) + 8-7
=
9-9
+ (9-9)2
^
=
16-5 amps.
The total power in the rotor is given by:
Rotor power/phase = (I2)2 x R2'/S1
and the rotor power loss will be
Rotor power loss/phase = (I2)2 x R2'.
The total mechanical power output from the three phases of the motor
therefore equals
Power out = ((I2)2 x R2'/S1 - (I2)2 x R 2 ) x 3
= (13 x 13 x -23/-02 - 13 x 13 x -23) x 3
= (1943-5 - 38-9) x 3
=
1904-6 x 3
=
5714 watts.
Motor speed = 3000 x 20/50 x (1 =
02)
1176RPM
From equation (18)
Torque = 5714 x 60/(2 x n x 1176)
= 46-4 Newton metres.
From equation (16)
Power loss per phase
= (16.5)2 x -24 + (13)2 x -23 + (150)2/500
=
149-2
Total motor power losses for the three phases
=
149-2 x 3
= 447.6 watts.
32
AC motors
1.3 The synchronous motor
The synchronous motor has not gained such widespread use in Europe as the
induction motor, possibly due to its not being a self starting motor, and maybe
due to the higher cost of manufacture.
The situation has been different in North America and this may be due to the
higher operating power factor and the stronger influence of electricity tariffs in
this respect.
Its history to date has mainly been affected by the almost universal acceptance
of AC motors as fixed speed motors and the lack of availability of variable
frequency controllers. The torque is only generated in this motor when the rotor
is running at the synchronous speed and when connected to a fixed frequency
mains supply it is impossible to start it as a synchronous motor. The result is
that most synchronous motors which were used, were a combination of induction
motors and synchronous motors, the motor starting up using its induction
motor arrangements and then synchronising at full speed to gain the benefits of
its more efficient, higher power factor synchronous motor condition. The design
of such motors was a compromise between the two requirements and this
resulted in it not being the ideal design for either condition. They often do not
possess a high enough starting torque and the presence of a starting winding has
complicated the motor design.
When the synchronous motor is used with a variable frequency controller
there is no need for special starting features to be included because the necessary
starting torque can be generated by its synchronous motor action by gradually
increasing the applied frequency. The motors being considered in this section
are motors suitable for use with variable frequency controllers where torque is
available at all speeds and no inherent starting capabilities in the motor itself are
necessary. In this respect the motors which will be considered are almost the
same as the AC synchronous generators which are produced in very large
quantities for local diesel generation sets where the prime mover is available to
accelerate the machine up to its synchronous speed.
The synchronous motor has a stator winding of the same type as with an
induction motor but now the rotor has a DC winding which produces the air
gapfluxdirectly. Torque generation only occurs when the rotor field winding is
supplied with power and the rotor is rotating in synchronism with the rotating
field caused by the stator MMF.
The motor is therefore supplied with two sources of power, 3 phase AC to the
stator and DC to the rotor field and both of these can normally be adjusted
independently so that the motor conditions can be more accurately controlled.
The motor always runs in absolute synchronism with the stator rotating MMF,
changes in torque only causing an alteration in the angular displacement
between the rotor and the rotating MMF.
AC motors
33
1.3.1 Synchronous motor principles
The synchronous motor stator winding is identical to that described for an
induction motor in Section 1.2.1. It produces an approximately sinusoidal
MMF in the air gap rotating at a speed decided by the applied frequency, see
Figs. 1.4 and 1.5.
When DC current is passed through the rotor windings a similar field MMF
is produced and when the rotor is rotating at the same speed as the stator
rotating field, then the two fields interact to generate torque. Effectively the
stator field drags the rotor around due to the force of attraction between the
field poles of opposite polarity. When zero torque is being produced the rotor
poles coincide with the stator poles. As more torque is demanded the rotor lags
behind the statorfield.If it is possible to demand a large torque so that the rotor
lags behind by more than 90 electrical degrees then the rotor will drop out of
step and torque generation abruptly ceases.
In the induction motor thefieldin the air gap is produced by the stator current
being larger than the rotor current, the difference producing the air gap MMF.
In the synchronous motor the air gapfluxis the result of the difference between
the stator MMF (produced by the stator winding current) and the rotor MMF
(produced by thefieldwinding). These twofieldscan vary in magnitude (due to
changing levels offieldand armature current) and in phase (due to the application
of torque as above).
Fig. 1.20 shows the formation of the air gap MMF under a number of
operating conditions. The plots show the air gap straightened out with the
magnitude of the MMF around it. In (a) the stator current (and therefore
MMF) is low and very low torque is being generated. In (b) a larger stator
current is producing torque by virtue of the load angle displacement y; the
resultant flux has been reduced and its position moved with respect to the
appliedfield.In (c) the stator MMF is even larger and is generating more torque
due to the increased y angle. The total air gap MMF is now nearer the applied
field in magnitude but again it is displaced with respect to the applied field.
Therefore the resultant field in the air gap varies widely between operating
conditions and it is thisfieldwhich causes the appropriate voltage to be induced
into the stator winding.
When this machine is used as a motor a voltage is applied to its stator winding
and under steady running conditions the voltage induced into the stator windings
has to be almost equal and opposite to the applied voltage in order that only a
modest current willflow.The angle y between the rotor and the stator MMF will
change automatically until the applied and induced stator voltages do balance.
While the balance condition is being located the additional current which flows
will be such as to cause the rotor to move to the desired stable position with
respect to the stator rotating field.
An increase in the applied field will also cause a stronger attraction between
the stator MMF and rotor and the load angle y will reduce so as to reduce the
resultant field to its original value.
34
AC motors
In general therefore the angular position of the rotor with respect to the
rotating stator MMF varies significantly as the applied voltage, field strength
and stator currents vary. As long as y remains in the region zero to 90 electrical
degrees stable operation will be possible.
^applied field MMF
, resultant MMF
MMF
Fig. 1.20 MMFs in the air gap
As I mentioned earlier these conditions can only occur if the rotor is rotating
at the same speed as the stator rotating MMF, i.e. in synchronism with the
stator frequency.
AC motors
35
Torque production
Torque is produced as a result of the interaction between the resultant air gap
field and the currentflowingin the stator windings. It will therefore be affected
by the magnitude of these two items and the angle which occurs between them.
It is in fact the component of the stator current which is in quadrature to the
resultant field which generates the torque directly.
Referring to Fig. 1.20 the torque generated is proportional to the magnitude
of the stator current, the magnitude of the resultant flux and the cosine of the
angle 0, i.e.
Torque ex stator current x flux x cos $
Construction
The stator core and windings are very similar to that of an induction motor. The
core will be made from laminated iron sheets punched with slot space for the
windings which will cover the complete perifery. The connections at the ends of
the core will be arranged so that the conductors are connected into the three
windings displaced by 120 electrical degrees from each other.
The rotor carries a DC winding which produces an appropriate number of
poles to match the number of MMF poles generated by the current flowing in
the stator. Most motors operating at less than 2000 RPM will have a salient pole
arrangement where the coils are wound around distinct iron poles usually
having soft iron caps of a shape suitable to produce the required air gap field
profile. Motors used with variable frequency drives will usually have laminated
steel rotors in order to minimise the adverse effects produced by harmonics in
the current and voltage waveforms.
The rotors of higher speed motors are more likely to be cylindrical in order
that they can withstand the higher forces involved. In these cases the rotor coils
will be inserted into slots in the rotor surface. The very large high speed motors
have rotors machined from solid steel forgings and it is necessary to take special
measures to ensure that any harmonic currents do not cause overheating in the
rotor iron. DC current has to be passed through the rotor windings and it may
be transferred to the rotor via two slip rings with brushes.
In such cases a controlled source of DC can be placed away from the motor
to provide the field power. An alternative approach is to mount an additional
small exciter machine on the same shaft of the motor so that the power can be*
transferred magnetically to the rotor rather than via brushes and slip rings. One
such system is described more fully in Chapter 7. Another alternative which is
becoming more possible particularly with small motors is a permanent magnet
rotor, but clearly it is not then possible to control the level of air gap flux and
this may lead to a rather special variable frequency drive/motor combination
being used.
Even though it is not necessary to build in an induction type rotor cage
winding for starting purposes, one may still be included to help in stabilising the
36
AC motors
operation of the motor or to control the value of subtransient reactance. In a
synchronous motor the rotor is effectively held by the magnetisation between
rotor and stator and this behaves like a variable tension spring. The result is that
while rotating at a 'fixed' speed the rotor can oscillate about a mean position
under the influence of the magnetic pull. If the rotor is provided with a short
circuited cage type winding such oscillatory movement of the rotor will cause
currents to flow in the cage and these will help to stop the oscillation quicker
than would otherwise be the case. It is usual to put copper rods into the pole
faces and connect them together at the two ends of the rotor.
13.2 Synchronous motor equivalent circuits and vector diagrams
When a synchronous motor is being considered in relation to operation at
variable speed with variable frequency supplies it is most satisfactory to consider
the rotor and stator conditions separately. The MMFs in the air gap should be
considered independently to the stator winding effects.
R1
12
E1
E1
V1
-11X1
El
-I1R1
-El
applied
field
vi
tv-m1
/-I1-R1
-E1
11
/ *|IL
>xi
^12
leading
unity
/
lagging
Fig. 1.21 Synchronous motor stator equivalent circuit and vector diagrams
AC motors
-E1
resultant MMF
flux
arm.reaction
MMF
I
applied MMF
resultant MMF
arm.reaction MMF
applied MMF
resultant MMF
\
\
Fig. 1.22 Flux and MMF diagrams
arm.reaction
37
38
AC motors
Stator conditions
The stator winding can be represented in a similar manner to an induction
motor. The winding can be represented as a coil with resistance and with some
leakage reactance mainly caused by the end winding connections.
The iron losses are also taken into account by the inclusion of the resistance
RL and the loss current IL.
Fig. 1.21 shows the single phase equivalent circuit of the statoir winding with
appropriate vector diagrams corresponding to leading, unity power factor and
lagging current conditions. The induced voltage El is produced by the rotating
air gap flux cutting the stator conductors. The current which flows causes
voltage drops in the resistance and leakage reactance of the winding so causing
the applied voltage to differ from the induced voltage.
Two points to note here are:
1) The input current to the motor can be at any power factor depending
on the air gap flux conditions. Clearly the normal aim will be to
arrange conditions so that a power factor as near to unity as possible
is maintained in order to achieve maximum torque per amp of current
flow.
2) In these vector diagrams the value of El, the induced voltage, is
shown at a constant value. In practice this value is produced by the
resultant flux which can vary over a wide range.
Therefore in contrast to the induction motor, it is mainly the rotor and air gap
conditions which dominate the performance of the synchronous motor.
Rotor andfluxconditions
The air gap MMF circumstances explained in Fig. 1.20 are best expressed in
vector form so that a wider range of operating conditions can be explored.
Fig. 1.22 shows the vector diagrams which correspond to Fig. 1.20(a), (b) and
(c) respectively. The starting point in each of thefiguresis the appliedfluxwhich
is assumed to be constant and the reference vector. The armature reaction MMF
produced by the stator current is shown in phase with the current producing it.
The induced voltage in the winding is shown in quadrature to and proportional
to the resultant MMF and the flux so produced.
The induced voltage and current vectors have been shown in dashed lines to
differentiate between the stator vector diagram and that of the rotor MMFs.
The rotor MMFs are single phase vectors which apply to all three stator phases.
The resultant flux is a single entity which rotates in the air gap inducing three
voltages into the three phase displaced windings.
The practical understanding of the motor is, however, greatly enhanced by
including these two diagrams, the air gap MMF diagram and the stator electrical
diagram onto one composite picture and this is what has been done on the
complete motor 'vector diagram', Fig. 1.23. Because the most frequent method
of use involves control of applied field MMF (Mf) so as to ensure a constant
value of resultant MMF (MM) and henceflux(Flm) the diagram has been drawn
AC motors
39
with the flux as the horizontal reference vector. The armature reaction MMF
(Ma) is shown in phase with the stator current and the resultant MMF in phase
with the flux.
Operation under variable frequency conditions
Most practical circumstances involve the control of applied MMF (by controlling
thefieldcurrent) so that the resultantfluxremains at a constant value at all times
with the value of induced voltage El then being proportional to the applied
frequency. With the synchronous motor this can be achieved at the same time
as controlling the power factor of the input current. vi
11X1
Fig. 1.23 Combined stator/rotor vector diagram of a synchronous motor
If the stator current is controlled to remain at the same angle to El as shown
in Fig. 1.23 then, as load changes, the current vector would move up II and the
applied MMF (Ma) would move along the line ABC.
If the current was to be kept in phase with El then the applied MMF would
have to follow the line AD. A constant leading power factor condition can be
40
AC motors
achieved by allowing the applied MMF vector to follow the line AF as the load
changes.
1.3.3 Synchronous motor equations and relationships
Again the steady state relationships between the motor currents and voltages at
varying applied frequencies and loads is carried out by writing down the
equations from the equivalent circuits and vector diagrams.
One fact which is always true about a synchronous motor is that the speed
and the frequency are directly related at all times irrespective of load or the
method of control being applied, i.e. rotor speed is always given by:
S = 120 x F/P RPM
(1)
where F is the applied frequency in hertz and P is the number of rotor and stator
winding poles.
Because the optimum conditions of operation are usually associated with the
use of a constant resultant air gap flux this condition will be studied now.
Relationships at constant flux
The induced voltage El will always be directly proportional to the applied
frequency if the flux is maintained constant. Therefore
El = Elr x F/Fr
(2)
where Elr is the value of the induced voltage at rated frequency Fr.
The current drawn by the motor will be affected by the amount of torque
being generated and then the shaft torque is given by:
T oc 12 x COS (An2) x Flm
(3)
if a constant flux is being assumed then
T oc 12 x COS (An2)
To solve the system further it is necessary to relate 12 and An2 to the air gap
MMF and the applied field.
The resultant MMF required to produce this flux value will be related to the
induced voltage by a curve similar to that of Fig. 1.13, for the present let us
assume that the rated flux value requires an MMF value of Mmr, a constant
value, independent of frequency and load. As shown in Fig. 1.23 this value will
be related to the appliedfieldMMF (Mf) and the armature reaction MMF (Ma)
of the stator winding and its current.
The next object is therefore to find the value of Mf and the angle relating it
to the resultant flux.
The armature reaction MMF (Ma) is proportional to the armature current
and dependent on the detailed motor design
Ma = Ka x 12
(4)
where Ka is the design constant.
AC motors
41
Also the appliedfieldMMF (Mf) is proportional to the currentflowingin the
field winding, therefore:
Mf = Kf x If
(5)
where Kf is again the appropriate motor design constant.
From Fig. 1.23
Mm = Mf x COS (Bnl) + Ma x SIN (An2)
also
(6)
Mf x SIN (Bnl) = M a x COS (An2)
= Ka x 12 x COS (An2)
(7)
These equations allow the relationship between Mf, Bnl, 12 and An2 to be
decided once a further statement about An2 is made. If the value of An2
corresponding to a specific torque and current condition is stated then the values
of all the parameters can be deduced for all other conditions.
Once 12 and An2 have been established then the input current, II, and the
stator resistance and reactance voltage drops can be evaluated and the terminal
voltage VI calculated in the same way as with the induction motor (refer to
Section 1.2.5).
In this case the power losses have to be calculated separately as the total
power losses equal
Stator copper and eddy current losses = 3 x (II)2 x Rl
+ Core losses = 3 x (E1)2/RL
+ Excitation losses = (If)2 x Rf
+ Bearing friction losses
+ Windage losses
The mechanical power being transferred to the rotor can be calculated from the
vector diagram as
Mechanical power = 3 x 12 x El x COS (An2)
(8)
and the power output can be found by subtracting the mechanical bearing,
friction and windage losses from this value.
Summarising therefore, the most difficult aspect of the synchronous motor
relationship is the effect of a variation infieldcurrent on the stator current and
its power factor.
Figure 1.24 shows a typical case.
The stator current has a minimum value at a specificfieldcurrent value when
the motor is operating at its unity power factor point. This point varies with the
torque applied to the drive. This demonstrates clearly how essential it is to
maintain continuous control over field current if a satisfactory and optimum
variable speed drive is to be produced.
42
AC motors
The solutions to other conditions are carried out in a similar way to the above;
in general, the relationships in the synchronous motor depend on the magnitude
of the armature reaction field, the initial set up conditions and the field control
strategy adopted.
unstable
region
leading
current
lagging
current
field current or MMF
•
Fig. 1.24 Typical field current/stator current relationships
13.4 Examples of synchronous motor calculations
1) Speed voltage and flux
Question
A 100KW, 6 pole synchronous motor is designed to operate from a 415 volt,
50 hertz supply under rated conditions. At what speed would the motor have to
be operated to ensure approximate ratedfluxconditions when 200 volts line was
applied to its terminals.
Answer
Constant flux is indicated if the voltage per hertz is constant, therefore
F/Fr = 200/415
F = 200 x Fr/415
F = 241 hertz.
AC motors
43
The speed of this frequency will be equal to
S = 120 x 241/6
= 482 RPM
Question
If the motor was required to operate at a speed of 1400 RPM and the rated
supply voltage could not be exceeded. What would be the level of air gap flux
in relation to the rated level.
Answer
Rated V/F = 415/50 = 8-3
The speed of 1400 RPM requires a frequency of
F = 1400 x 6/120
= 70 hertz.
Operating V/F at 1400 RPM therefore
= 415/70 = 5-93
Therefore flux will be
5-93/8-3 = -714
i.e. flux at 1400 RPM equals 71-4 per cent of the rated flux.
2) Armature reaction and power factor
Question
The above motor operates at 0-9 power factor lagging under its rated load
condition. Ignoring losses and stator impedance how much will the applied field
have to be increased to make the motor operate at unity power factor, at rated
load, if the applied field MMF is initially equal to the resultant air gap MMF.
Answer
Referring to Fig. 1.25 which is a simplified vector diagram of this condition
II is the rated current at 0-9 pf lagging.
Triangle OAB represents the rated MMF diagram with
OB representing the applied MMF,
OA the resultant MMF and
AB the armature reaction MMF
OA = OB initially
Angle An2 = ACS(0-9) = 25-84
Ani
Mf1
Fig.1.25
AC motors
45
AB = 2 x OA x COS (90 - 25-84)
= -872 x OA
AC = AB x COS (An2)
= -872 x OA x -9
= -785 x OA
OC equals the value of applied MMF needed to cause the motor to run at
unity power factor when AC will then represent the armature reaction MMF.
OC = ^OA 2 + AC2
= OA x Jl
+ (-785)2
= OA x 1-27
= OB x 1-27
Mf2 = Mfl x 1-27
i.e. a 27 per cent increase in applied field.
1.4 Harmonics and AC motors
Most variable frequency drive systems use switching circuits which do not
inherently produce sine waves, they normally produce square or pulsating
voltages or currents. If such waveforms are to be supplied directly to the AC
motor then some understanding of their effects on the motor is essential.
The most satisfactory way of considering such complex waveforms is to split
them into a sunusoidal fundamental wave of the desired frequency and then
consider the effect of the remaining harmonics separately. The assessments
which have been made in this chapter up to now are applicable to sinusoidal
waveforms and they can be applied to the fundamental values extracted from
the complex waves being generated.
The unwanted harmonics will normally be at frequencies higher than the
fundamental and many of the drive systems are specially arranged to avoid
frequencies below or near to the fundamental values. The frequencies of the
harmonics may be directly related to the fundamental frequency as in the case
of the six step systems described in Chapters 4, 6 and 7 (where the main
harmonics are at six times the fundamental operating frequency) or they may
be unrelated to the fundamental as is the case in the cycloconvertor and some
of the voltage source PWM systems.
The type of drive system also decides whether the harmonics occur predominantly in the voltage or the current. Voltage source inverters usually produce
complex voltage waveforms containing large amounts of harmonics and the
46
AC motors
resultant current waveforms tend to contain a lesser value of harmonics.
Conversely, current source systems produce currents with high harmonic
contents and in many cases the voltage waveforms tend to be more sinusoidal.
It is not usual to find similarly large harmonic contents in both the current and
voltage of any system.
In general, harmonics do not produce any useful torque in any of the systems
discussed in this book, they produce additional losses and harmonic pulsations
in the output torque from the motor shaft.
1.4.1 Harmonic power losses
Additional power losses can be produced in any part of the motor and the
following gives some general points on these:
1) The presence of harmonics in the currents in motor stators and rotors
means that the total RMS value of the currents will be larger than the
fundamental value which actually does the work. This increase in
RMS current obviously produces an increase in losses.
2) Additional harmonics in the current will be at higher than normal
frequencies and due to skin effects the resistance of the windings at
these frequencies may be higher than at fundamental frequency. Thei
stator windings will not be affected too much by this effect but it may
be more significant in the rotor losses in induction motors. The use of
deep rotor bars in some designs means that the rotor resistance will
increase considerably with harmonic frequency.
3) Harmonics in the applied voltage waveforms to the motor can cause
an increase in the core and stray losses in the motor.
The actual amount of the extra losses caused by harmonics can only be accurately
assessed by the motor designer after he has received specific information from
the designer of the variable frequency controller. The individual drives chapters
give some details in this respect and the information below will also help in the
overall understanding. However whatever the magnitude of the extra loss turns
out to be it is usually dealt with by derating the motor's output performance,
i.e. by using a larger motor than would be the case if the waveforms were
sinusoidal. It is very unusual if an allowance of more than 10 per cent is needed
with any practical drive designs and very often the allowance can be much less
than this.
1.4.2 Torque pulsations
The torque is generated in AC motors by the reaction between the air gap flux
and the rotor current in the case of an induction motor, and the stator current
in the case of a synchronous motor.
It will be found from the discussion following this that the air gap flux is in
general relatively unaffected by harmonics, because the magnetic circuit is
basically inductive higher harmonics do not cause a significant value of harmonic
AC motors
47
flux. The torque pulsations which may be generated are therefore the result of
the harmonic currents in the appropriate motor windings.
If the rotor current of an induction motor contains a significant content of
harmonics then the result will be significant torque pulsations. The induction
motor acts in a similar way to a transformer in that once an allowance has been
made for the necessary magnetising current then the stator and rotor currents
(or more correctly amp turns) balance and this includes the harmonics. Therefore if harmonics are contained in the stator currents, then they will occur in
similar magnitudes in the rotor currents and cause torque pulsations as a result.
In a synchronous motor the substantial sinusoidal resultantfluxreacts with the
stator current directly and therefore the resulting pulsations are dependent on
the harmonic content in the stator current.
In general harmonics in the voltages applied to AC motors only affect the
level of torque pulsations if they cause direct results in the flow of harmonic
currents.
The drives which produce the largest value of torque pulsations are therefore
the current source systems where the current waveforms are clearly far from
sinusoidal. More details regarding these pulsations are given in the current
source drive chapters following.
1.4.3 Harmonic equivalent circuits
Further understanding of these effects can be made by considering the equivalent
circuits of motors under harmonic conditions. This can enable an assessment of
the likely levels of harmonic currents to be made.
In an induction motor the magnitude of the equivalent circuit components
will alter when the harmonic frequencies are considered. The impedance values
of the circuit leakage inductances will increase; they are effectively air cored
inductances, their values will increase roughly in proportion to frequency. The
stator resistance value will remain approximately constant, frequencies up to
1000 hertz are not likely to cause significant change due to skin effects. The rotor
resistance however, will change and it could rise to twice the standstill value due
to the influence of the deep bar designs used to improve starting performance.
The use of an equivalent resistance for the core loss is not entirely accurate for
harmonic considerations but this item is not significant to the overall flow of
harmonic currents.
The effective value of slip increases considerably when the harmonics are
considered. Anyfifth,harmonic, for example, produces a rotating magnetic field
which rotates atfivetimes the fundamental MMF wave speed, the result is that
the slip is approximately six per unit, taking account of the rotor speed.
Fig. 1.26 shows these effects in graphical form.
Let us firstly consider the magnetising circuit and the harmonic currents
flowing in it. At higher frequencies the value of Xm increases in proportion and
in addition the likely magnitude of the harmonic current or voltage reduces.
Xm is therefore, in general, a high impedance to harmonic current flow. For
48
AC motors
example, if a 50 hertz, quasi-square wave is applied to a typical 55 KW, 415 volt
motor, the magnetising current of the motor would contain approximately
30 amps of fundamental current with a total of harmonic current of less than
0-75 amps RMS, i.e. less than 2-5 per cent. Because the applied voltage normally
alters with the frequency this relationship exists throughout the normal speed
range. Even when harmonic currents are injected into the motor from a current
source type drive the impedance of the magnetic circuit is such that the harmonics
tend toflowelsewhere, i.e. in the rotor rather than into the magnetising system.
R1
X1
nR2
harmonic number =
/st
harmonic frequency
rated frequency
Fig. 1.26 Harmonic values of equivalent circuit parameters of an induction motor
The very important conclusion to the above is that whatever waveform you
apply to or feed into an induction motor the induced voltage is always very close
to sinusoidal and for theoretical study the assumption of a sinusoidal fundamental
induced voltage is valid and does not introduce significant inaccuracies.
AC motors
49
The follow-up point to this is that if harmonic currents do not flow in the
magnetising circuit then they must flow in the rotor circuit. It is the interaction
between these rotor harmonic currents and the sinusoidal fundamental flux
waveform which causes pulsations in the output torque.
In the synchronous motor the induced voltage in the stator windings also turns
out to be sinusoidal whatever the waveshape of the stator current. Even in the
severe case of the six step current source drive described in Chapter 7 where
currents with 25 per cent harmonic distortion are fed into the motor the induced
voltage is still sinusoidal at fundamental frequency. The only reasonable
explanation for this is that the harmonic effects caused by the stator MMF
waveform induce equal and opposite changes in the rotor field current to take
place so that the resultant flux remains sinusoidal. The presence of harmonic
currents in the stator still produce pulsations in the torque whose value is related
to the proportions of the harmonics, but they produce negligible harmonics in
the magnetic circuit.
1.5 Motor power losses
When motors are used at variable speed and variable frequency the power losses
which occur in them cannot be assessed simply, because the many components
of the total loss vary in different ways as the speed, frequency, voltage and
current are altered. The total losses under any specific operating condition can
only be found by studying the individual components on their own and assessing
the likely value of each and then adding them up to arrive at the total. This
section is intended to identify the relationships which occur between the individual
components of loss and the variable parameters of the system. Firstly we will
discuss the individual components of the total power loss.
Copper losses
The electrical windings of a motor will always have a finite resistance, however
small, and this will cause power losses which are generally proportional to
I2 x R, where I is the currentflowingthrough it. As long as I is the total RMS
value of the current this will represent the total power loss in either an AC
or a DC circuit and it will automatically take into account any harmonic
component in the current.
The value of the winding resistance will increase with temperature and this
will depend on the current loading and the effectiveness of the overall cooling
of the machine. To be on the safe side the resistance at rated temperature should
be used in loss calculations.
The above approach of using the total RMS value of the current will
be satisfactory in most circumstances but if more precision is required it is
necessary to consider any harmonic content in the current more thoroughly.
Due to skin effects where the current tends to concentrate in the outer layers of
50
AC motors
the conductor, the resistance of the winding will not be the same to each
harmonic frequency, it will increase as the frequency rises. The more correct way
of assessing the losses in this case is to sum up the individual I2 R values for each
harmonic contained in the total current.
The skin effects can be more noticeable in the rotor windings of induction
motors where deep bars are often employed as a means of increasing the mains
operated starting torque. However, the skin effect only affects the losses caused
by harmonic components in the rotor current because in normal variable
frequency operation the motor slip speed is small and approximately proportional to torque. The one exception to this is in the case of the slip power
recovery system where the slip speed increases as motor speed is reduced.
However, the slip power recovery system always uses a wound rotor and skin
effects are not so pronounced as they are in cast cage windings, etc.
In general, due to harmonics, the copper losses in AC motors will be higher
than those occurring when the motor is used under sinusoidal operating
conditions at the same power. The amount of the increase will depend on the
harmonic content of the current waveform and hence in the type of drive being
employed.
Except for the skin effects the copper losses in motors do not alter significantly
with speed, frequency or voltage changes.
Iron losses
In every cycle of operation, thefluxin the core of the motor will be reversed and
this causes a small loss of energy usually called hysteresis loss. Its value depends
on the quality of iron being used, the value of the flux density over which it is
being used and the frequency of operation.
For a particular iron circuit detailed study shows that this loss is proportional
to:
frequency x (flux)*
where x is dependent on the quality of the iron and which normally varies
between 1-5 and 2-5. Traditionally x equals two has been used for the general
unspecified case for AC magnetic circuits.
The total iron losses also contain another component, an eddy current loss
due to induced currents flowing in the iron. In order to reduce this loss to
relatively small proportions AC magnetic circuits are laminated using thin
insulated sheets clamped together. There will still be a small eddy current loss
and for a specific iron circuit this loss will be proportional to:
(frequency)2 x (flux)2
The total iron loss will therefore vary in a relatively complex way with flux
and frequency depending on the relative sizes of these two components. Typically
the eddy current loss at rated conditions will be only one third of the hysteresis
AC motors
51
loss and hence
Iron loss = KH x F x $ 2 + KE x F x (j>2
at rated frequency
KH x Fr x ft = 3 x KE x Fr2 x ft
KH = 3 x KE x Fr
Iron loss = KE x F x cj)2 x (3 x Fr + F)
as flux is given by induced voltage divided by frequency
Iron loss = KE x El 2 x (3 x Fr + F)/F.
where KE and KH are loss constants.
Harmonics in the supply voltage do not in general increase the peakfluxlevel
in the iron and the above losses are not significantly affected. However, there
may be an increase in stray losses resulting from the harmonics.
Friction losses
In general the power losses in bearings will vary in direct proportion to the speed
of the motor.
Windage losses
These are caused by the fans which may be mounted on the rotor for cooling
purposes and by the rotation of the motor itself. The power losses caused by
windage will be proportional to the third power of the rotor speed and hence
they will drop to very low levels under low speed operating conditions.
Excitation losses in synchronous motors
The power required to provide the necessary excitation in a synchronous
machine is normally included in the total power losses of the motor. As most
motors are operated at a constant flux the amount of power required for this
purpose does not change significantly as the speed or frequency of the motor is
changed. However, the amount of excitation current required does vary significantly as the motor stator current changes due to the effects of armature
reaction. It is not unusual for the excitation current to have to double between
no load and full load current in the stator and this means that the excitation
power required is increased by four times. The required excitation power will
therefore vary with stator current and its power factor.
There may also be another machine providing the excitation power and some
power loss will occur in this exciter.
Damper cage losses in synchronous motors
Most synchronous motors, even those used with variable speed controllers, have
a cage type winding on the rotor to stabilise the operation of the motor and to
52
AC motors
enable an appropriate value of reactance to be obtained. This cage does not
carry any fundamental current except when the rotor or position is changing,
when it has a current induced in it which causes a steadying torque to be
generated.
This cage will however carry induced harmonic currents and these will
produce some heat losses. Such currents and losses will be most significant when
the motor is used as a current source drive where the motor stator current has
a significant harmonic content.
1.6 Motor voltages to earth
The conventional use of A.C. motors and generators is for them to be connected
directly to A.C. distribution mains supply systems and such systems are
normally arranged to be positively related to earth potential. It is usual for A.C.
Systems to have their neutral points earthed directly or for them to be earthed
through a resistance, an inductance or via a transformer. The result is that under
normal operating conditions the mains neutral point is near to earth potential
and it is only during fault conditions that the neutral point can move substantially away from this point. The windings of the motor are therefore normally
only exposed to the peak of the mains phase voltage with respect to earth and
occasionally perhaps, to the peak of the line voltage if there is an earth fault on
one phase terminal of the motor.
The motor windings are housed in slots in the earthed metal core of the motor
and the insulation surrounding the windings is normally chosen in relation to
the operating voltage and the mains supply earthing arrangements as described
above.
When a motor is fed from an electronic variable speed drive convertor its
windings are not so closely related to the mains network and it is possible for
the potentials of the motor windings to rise significantly higher with respect to
earth if the supply side mains supply has an earthed neutral as is normally the
case. This situation is mainly related to d.c. link type convertor systems where
the mains power is converted to d.c. using a set of electronic switches and then
inverted to variable frequency A.C. to supply the motor. In these systems there
are two separate voltage generators, the mains supply and the motor itself and
they will be connected together via switches which are capable of being closed
at any time during the cycle of both mains and motor voltages. The result is that
under some circumstances the time of operation of the switches will lead to the
two voltages being added together for short periods of time and hence the peak
voltage between the motor and supply side of the systems can be significantly
increased. If the supply side of the system is tied down to earth then the result
is that the motor side of the system will rise to higher voltage levels with respect
to earth. Alternatively, if the motor side is connected to earth then the higher
voltages with respect to earth will appear on the supply mains side of the system.
AC motors
53
In general these higher voltages are sections of the supply and motor sine
waves as dictated by the operation of the switches, they are not sudden short
transients. The waveforms of the voltages to earth will contain harmonics
related to the frequency of the switching of the supply and motor convertors and
they can be very complex in detail. The peak value of these waveforms with
respect to earth varies considerably depending on the«type of drive system being
used and the particular operating condition, but in general it is possible for the
maximum value of this peak to be between 2 and 2.5 times the normal peak
voltage which would occur in a sinusoidal earthed neutral system. These
voltages will occur throughout the convertor which is more remote from the
earth point and they must be allowed for in the design of the equipment and the
motor.
When using d.c. link type drive systems with existing motors it is necessary
to make sure that the motor winding insulation is capable of accepting the extra
voltage to earth, or alternatively the motor side of the system should be tied
down to earth, while making sure that the supply side can cope with the extra
voltage. Double wound transformers sometimes need to be ased to isolate the
supply and motor circuits from each other to allow connections to earth to be
safely established.
Chapter 2
Power switching devices
2.1 Introduction
There is a very wide range and variety of semiconductor switching devices which
can be considered for use in AC variable speed drives and it is the purpose of
this chapter to explain the principles, the capabilities and the performance
characteristics of those devices which are most likely to be used for this purpose.
The chapter is intended to deal with semiconductor devices specifically from
an AC variable speed drive point of view; it is not in any way an exhaustive
1
study of all semiconductor devices available. As such I have decided to split the
devices into three broad classifications, namely:
Thyristors — the well established high power switching device which
can be easily turned ON but which is inherently difficult
to turn OFF.
Transistors — which are fully controllable amplifying devices, the power
equivalent of the heart of all electronic systems.
Gate turn off thyristor (the GTO) the latest power switching device which has all the
attributes of the thyristor plus the ability to turn it
OFF using gate control.
The early AC variable speed drives, the cycloconvertor and the slip power
recovery drive, used naturally commutated, slow speed switching devices such
as mercury arc rectifiers and subsequently, thyristors. Thyristors are still used
widely in such drives. As devices with faster switching speeds became available,
forced commutated thyristors were used in inverter circuits and the DC link
inverter type drives came into being. In due course it was found that the
switching speed of conventional thyristors could be improved no further and
development in the field of transistors came up with the much faster devices
needed for high frequency pulse width modulated systems. Unfortunately it was
not found to be possible to produce the required high switching speeds at the
same time as high blocking voltages and power ratings in transistors and again
work was done in other directions. The gate turn off thyristor is a device which
is now becoming available in high power and voltage ratings to replace the
Power switching devices
55
forced commutated thyristor in the higher power DC link inverter drive
systems.
In general the development of new and more advanced AC motor drive
systems has gone hand in hand with the improved semiconductor switching
devices which have become available. There is no doubt that this trend will
continue and that higher quality semiconductor devices will be developed in the
future and that appropriate improvements in drive systems will occur as a result.
In all power circuits the semiconductor devices are used as switches, i.e. they
are either operated in the OFF (no current condition) or the ON (high current
condition) and control is obtained by choosing the timing for switching between
these two conditions. This is because under these two conditions the power
losses in the semiconductors are at their lowest value; under the OFF state no
current isflowingand under the ON state the forward voltage drop across the
devices is at its minimum value. It is rarely practical to use devices under
conditions where a significant portion of the circuit voltage occurs across the
device while current is flowing in it because the very high internal dissipation
which results. This chapter therefore concentrates on the switching devices and
the switching capabilities of the devices.
2.2 The thyristor
The thyristor is a three terminal semiconductor switch capable of carrying
current from its anode to cathode, in one direction only and capable of blocking
voltage in either direction. It has two stable states of operation as shown in
Fig. 2.1, the OFF state with no current flowing, this occurs with no current in
the gate connection, and the ON state where current flows from anode to
cathode. It is possible to switch the device from the OFF state to the ON state
simply by passing a small current between the gate and cathode connections. It
is only possible to switch the device from the ON state to the OFF state by
reducing the currentflowingin the anode/cathode to zero for sufficient time for
the device to regain its blocking capability.
As seen in Fig. 2.1 the thyristor is a four layer semiconductor device with
alternative doped positive and negative layers, each containing free charge
carriers of the appropriate polarity, the current is carried through the device by
these charge carriers. When the anode is negative with respect to the cathode the
junctions Jl and J3 are reverse biased and J3 normally accepts most of the
circuit voltage across it. Whatever happens to the gate only affects the junction
Jl and J3 continues to block the flow of current.
When the anode is positive with respect to the cathode then the junction J2
is reverse biased and Jl and J3 offer no resistance to the flow of current. With
no gate current flowing J2 will continue to block the voltage. If under this
forward blocking condition, a current is passed from the gate to the cathode the
electrons flowing from the cathode to the gate are arranged to 'spill over' and
56
Power switching devices
prevent the J2 junction from continuing to block, hence initiating current flow
between the anode and cathode. Once this process has commenced, electron
multiplication takes place due to atomic collisions and the silicon triggers itself
into a highly conducting state, the ON state. Once this self sustaining process
is under way, the gate current ceases to have any influence.
anode
anode
gate
+*P2+ + + J3
- _ - N 2 = - J2
current
flow
direction
+
I I N1 I I
J1
gate
cathode
cathode
off-state
anode /cathode voltage positive or
negative on - s t a t e
high resistance
low resistance
anode positive w.r.t. cathode current flow •
Fig. 2.1 Thyristor principles
All present day thyristors are produced from doped silicon material and the
switching action takes place completely in the solid silicon slice which is mounted
in a sealed enclosure usually containing an inert gas. Very high capability
thyristors can be produced; most have OFF state leakage currents from 1 to
100 milliamps even though they may be capable of carrying anode to cathode
currents of 5000 amps and blocking peak voltages of 4000 volts. Although the
majority of thyristors have turn-off times of around 500 microseconds enabling
them to be used at frequencies of up to 200 hertz, fast turn-off devices only
taking 10 to 20 microseconds to turn-off are available and these are used for
operation at frequencies up to 1000 hertz.
Like all semiconductor devices they need to be fully understood if safe and
reliable operation is to be obtained. They have very specific and absolute limits
Power switching devices
57
of capability and it is essential that they are fully protected against all the
possible situations which could damage them.
2.2.1 Capabilities and performance
The thyristor is a one way switch, it is either ON or OFF and it can only carry
current in one direction. It can be switched from the OFF to the ON state by
passing a small current through its gate connection. It can only be returned to
its blocking OFF state by removing the anode/cathode current for the turn-off
time.
Voltage capabilities
The thyristor is capable of blocking voltages in either direction and most can
block similar values of voltage in the two directions. They are sold with stated
voltage capabilities and devices having to withstand capabilities from 50 volts
to 5000 volts are available.
If the reverse voltage (cathode positive) applied to a thyristor exceeds its
capability it can be irreparably damaged due to the excessive leakage current
which would flow. If an excessive forward voltage (anode positive) is applied to
the device while in its OFF state it can be made to switch into its ON state and
in so doing it may be damaged. It is therefore essential to make sure that
thyristors are used well within their assigned voltage capabilities.
Fig. 2.2 shows the voltage characteristics of a thyristor. In the blocking state
the leakage current flowing is almost independent of voltage and is a very low
value, once the voltage capability is exceeded the reverse leakage current very
quickly increases and in the forward condition it switches into the ON state.
Although forward breakover itself does not immediately damage the thyristor,
the current which may flow in the device as a result of its breaking over may
cause it to be damaged or its capabilities to deteriorate.
A thyristor's maximum repetitive forward voltage capability (VDRM) is the
maximum value which it can withstand except for the occasional slightly higher
transient. Its repetitive reverse voltage capability is designated its VRRM value.
In addition to the maximum values of voltage which can be applied to the
thyristor it is also necessary to limit the rate of rise of the forward voltage while
in the blocking state. If a critical value is exceeded then, due to capacitive effects
in the silicon material, it may be caused to incorrectly switch into its ON state
even though no gate current has been applied to it. Most power thyristors are
capable of accepting rates of change of voltage of between 100 and 300 volts per
microsecond but special fast devices may accept over 1000 volts per microsecond.
Unfortunately in many circuits in which thyristors are used the switching ON
of one thyristor can often cause a high dv/dt to be applied to other thyristors
in the circuit.
Semiconductors are particularly sensitive to the application of excessive
voltages and it is essential to ensure that a thyristor's stated voltage capability
is not exceeded even for fractions of a microsecond. When choosing thyristors
58
Power switching devices
for use in practical circuits therefore, it is necessary to allow for the absolute
peak conditions which can be experienced under normal, abnormal and faulty
operating conditions. It is essential to know what the peak voltage applied
across the thyristor will be and it is necessary to allow for variations in supply
and circuit voltage and for the possibility of transient overvoltages. In addition
it is usual to include overvoltage suppression circuits near to thyristors to ensure
that transient peaks are attenuated. When thyristors are used in inverter circuits
where there are large capacitors to ensure that the voltage levels cannot change
quickly and are predictable, then they can be used near to their repetitive
ratings. However, if the thyristor is to be connected to a mains supply which
may be exposed to sudden variations and lightning strikes, etc. it is important
to include an appropriate safety margin in choosing which thyristor to use. It
is not unusual for a thyristor only to be used at a working peak voltage of
40 to 50 per cent of its repetitive voltage rating.
anode
cathode
current
forward
characteristics
on -state
VD.R.M
i
y
VR.R.M
t
breakover
point
I I blocking
_ * / off-state
anode/cathode
voltage
reverse
leakage
current
reverse
characteristics
Fig. 2.2 Thyristor voltage characteristics
Current capabilities
Current flowing through the thyristor in the ON state causes a small voltage
drop across the device and this is usually between 1 and 2 volts at the nominal
current rating of the device. Fig. 2.3 shows typical shapes of the current/voltage
curves of thyristors and from this you can see that the voltage drop varies with
temperature and from device to device. In general there is quite a wide variation
in forward characteristics even between the devices of a particular manufacturing
Power switching devices
59
batch and it may be necessary to select them into narrower bands particularly
if a number of thyristors are to be operated in parallel. This voltage drop causes
power losses in the thyristor and it is normal to mount them onto suitable
heatsinks to dissipate this heat and to keep the temperature of the thyristor
down to acceptable levels. Most silicon thyristors are capable of operating at
junction temperatures of up to 125 degrees Centigrade without any reduction in
capability, but if this value is exceeded it is possible for the thyristor to fail to
block forward voltage and it can switch into the ON state even without any gate
current flowing.
range of spread
between individual
thyristors
1.0
forward voltage
2.0
Fig. 2.3 Forward on-state characteristics
You will note that the forward characteristic of the thyristor is non-linear and
so the value of the power losses generated by the current will depend on the
waveshape of the current and on the duty cycle of the load. Fig. 2.4(a) shows
a typical set of power loss curves produced by most manufacturers. If thyristors
are to be used correctly it is necessary to allow for the highest value of forward
voltage drop for the specific type of device and also to take the current waveform
and duty cycle into account. It is also necessary to ensure that the internal
temperature of the device is not exceeded and hence the temperature differential
from the heatsink surface to the junction also has to be allowed for.
The thyristor has a relatively low thermal mass and its junction temperature
can rise rapidly with increased current and power losses. If overloads lasting
more than a few seconds can occur it is necessary to take these into account in
60
Power switching devices
arriving at the useable current rating of the thyristor to ensure that the critical
125 degree Centigrade junction temperature is not exceeded.
Under fault conditions it is possible for the thyristor current to exceed the
above figures as long as it is not necessary to retain blocking ability after the
fault. In such cases the thyristor junction temperature can be allowed to rise to
very high values without permanently damaging the thyristor. Fig. 2.4(b) shows
typical curves which will describe a particular thyristor in this respect, the peak
current curve will be used to assess the fault conditions if circuit breakers are to
be used for protection and the I2t curve will be used if fuse protection is being
employed.
conduction angle
60* 12 0° 180° clc
dc ,
50 75
100 125
maximum cooling surface
temperature
mean on-state current
(a) temperature /power loss curves
0
In
c
o
Ia
3
O"
U)
1
4>
0
^
.
(am
5)
a
\
3
•
1
TJ
sec
?ea
:urr
f 6
"a C
> o
2 3A5
10 20 30A050
numberof cycles at 50Hz
(3
2
A
6
8
10
time(milliseconds)
(b)surge on-state current capapability of thyristors
Fig. 2.4 Thyristor thermal and overload capabilities
In general, current and surge capability in thyristors does not cost very much
money and it is usual to use a slightly larger thyristor than is really necessary
to ensure against occasional excessive conditions and guarantee very reliable
operation.
Power switching devices
61
Thyristors are also only able to accept a limited rate of rise of anode current
di/dt, when they are switched into the ON state. When gate current is initially
applied the anode current starts to flow near to the gate area and it takes some
time to spread throughout the thyristor, hence it is necessary to restrict the level
of current which flows immediately after switch ON. If the safe rate of rise of
current is exceeded immediate failure takes place due to excessive local heating
near to the gate region. Most thyristors will accept di/dt values of between 100
and 200 amps per microsecond but those specially made for fast turn off
applications can have allowable values of up to 1000 amps per microsecond. In
all cases it is necessary to ensure that the limiting values are not exceeded by
including small inductances into the circuit to control the di/dt.
Switching characteristics
Turn on
A thyristor can be switched into its ON state by the application of an appropriate pulse of gate current while forward voltage (anode positive) is being
applied to the thyristor. Turn on does not happen instantly but takes a finite
time made up of a delay time (when little appears to happen) and a rise time
during which the anode to cathode voltage falls. In general the total turn on time
will vary up to 10 microseconds in length and it will be dependent on the amount
of gate current used and the rate at which it rises. The higher the gate current
and the faster it rises, the shorter the delay time and the shorter the turn on time.
If, once the thyristor is conducting, the gate current is removed, the thyristor
will remain in the ON state as long as the anode to cthode current is above the
latching current for the thyristor. If the current is below this level it will switch
back into the non-conducting OFF state.
Turn off
The thyristor will stay in the ON state as long as the anode current remains
above the holding current. If the current reduces below this level then the device
will attempt to turn off. It will do so as long as the anode to cathode voltage
remains in the reverse direction for a specific time to allow the thyristor to
recover its forward blocking ability. If the current happens to be reducing
relatively rapidly the turn off process becomes a little more complicated because
the presence of free charge carriers in the silicon allows the current in the
thyristor to reverse as shown in Fig. 2.5. The reverse current cuts off rather
rapidly once the free carriers have been absorbed and a high reverse voltage
'spike' is produced. It is necessary to ensure that the forward voltage does not
occur until at least the turn off time has expired and when it is applied it should
not be applied at higher than the critical rate or the device may switch on again.
Gate firing
The gate to cathode circuit of a thyristor is a p-n junction and looks like a diode
62
Power switching devices
from the external point of view. It is only able to carry current in one direction
and it only has a very low reverse voltage capability of between 10 and 20 volts.
The value of gate current necessary to fire the thyristors will vary from thyristor
to thyristor and with temperature.
It is most satisfactory to use relatively high levels of gate current in single
short pulses or as pulse trains to minimise the overall gate dissipation produced.
High levels of gate current ensure that all thyristors willfireand also give the
maximum di/dt capability to the thyristor. But this level cannot normally be
applied continuously or else it may damage the gate junction.
specified di/dt
anode current
reverse recovery
current
turn-off time tq
specified
dv/dt
anode/cathode
voltage
t i me (miroseconds)
Fig. 2.5 Thyristor turn off
2.2,2 The available thyristors
A full range of thyristors is now available from the majority of the world's
established suppliers, it is the intention here to summarise details of those
thyristors which are readily available from a variety of sources. In general the
range spans from 1 amp to 10,000 amps through a single thyristor, and covers
repetitive voltage ratings from 50 to 5000 volts. Devices are available using
silicon discs from a few mm to 150 mm in diameter and they can weigh from a
few grams to nearly 2000 grams for the largest devices.
Many of the characteristics of thyristors are interdependent, in that to obtain
Power switching devices
63
good capability in one respect may adversely affect the achievable capabilities
in another. Hence the thyristors which are marketed widely tend to be those
devices having the most suitable balance of characteristics to meet a reasonable
wide range of applications. Within a certain size of slice higher voltage ratings
tend to correspond with lower current ratings. In order to obtain a fast turn off
capability it is usually necessary to accept a lower peak voltage capability and
sometimes a lower current rating due to the higher forward voltage drop value.
These interdependent factors have led to two basic groups of thyristor being
available: those for lower speed, mains frequency, naturally commutated applications and secondly fast turn off thyristors suitable for forced commutation,
high speed switching, inverter type applications. Both of these groups are used
in AC motor drives, the converter grade devices for the supply side converters
of DC link systems, for cyclo-converters and slip power recovery, inverter grade
thyristors are used in quasi-square wave and pulse width modulated inverters.
Fig. 2.6 This picture shows a full range of thyristors from the small stud mounted device to the
large disc encapsulated ones. It also shows the silicon disc slices which are the active
part of the devices. {Marconi Electronic Devices, Ltd.)
Converter grade thyristors
These thyristors are optimised for current and voltage rating and for high surge
current capability, other parameters take second place. They will normally have
a di/dt capability of 100 amps per microsecond and a dv/dt value up to 200 or
64
Power switching devices
so may be acceptable. They are usually available in selected forward voltage
drop bands for parallel operation and selections for operation in series are
obtainable. Turn off time is not normally declared but it would be expected to
be in the region of 200 to 500 microseconds.
Table 2.1 gives details of a typical manufacturers range of convertor grade
thyristors.
Inverter grade thyristors
These are usually for forced commutation applications where the critical parameter is turn off time. In order to obtain a low value of this, the manufacturers
have to use special resistivity silicon and particular charge carrier doping in the
manufacture of the slice. The result is that it is difficult to obtain high voltage
blocking capability at the same time. In general, therefore, fast turn off thyristors
are only available over a reduced voltage range compared to convertor grade
devices.
en
o
2000
I
C
J
o
o
1 000
J
A
J
J
J
J
J
>
- r a n g e of a v a i l a b l e devices
S
1
0
10
/
20
30
40
turn-off time-microseconds
50
Fig. 2.7 Available fast turn off thyristors
Inverter circuits also tend to produce high values of di/dt and dv/dt so the
device manufacturers have usually managed to design their devices to include
values up to 1000 amp per microsecond and 1000 volts per microsecond.
Turn off times as low as 10 microseconds can be obtained but the achievement
of very short turn off times usually means other parameters are limited. A range
of different selections are therefore usually made so that the user can optimise
between the circuit voltage and the cost of commutating components. Fig. 2.7
Power switching devices
65
shows a typical range of available fast turn off thyristors from the voltage point
of view — fast turn off means lower voltage capability.
Table 2.2 shows a typical manufacturers range of inverter grade, fast switching thyristors.
Assymmetric thyristors
Fast turn off characteristics can be optimised more readily if the reverse voltage
capability of the device is allowed to be very low. Many of the circuits which use
these thyristors have a reverse diode connected across every thyristor so preventing the application of reverse voltage, therefore it is often unnecessary for the
thyristor to have any reverse voltage capability. Assymmetrical thyristors are
devices with very low reverse voltage capability and they usually have a lower
turn off time than their bi-directional counterpart.
Reverse conducting thyristors
When fast switching thyristors are used in voltage source inverters a reverse
connected diode is usually connected directly across the thyristor. In order to
minimise inductive voltage transients it is necessary to use fast diodes (diodes
with fast recovery of blocking capability) and to mount these very close to the
thyristor to minimise the circuit inductance. Some manufacturers have decided
that the best solution to the difficulties caused by the required close proximity
between these two devices is to mount them both in the same encapsulation, i.e.
a reverse diode in parallel with a thyristor, both in the same housing. These are
called reverse conducting thyristors and using this technique it is possible to
obtain an optimum choice of parameters for voltage source inverter switching
use.
Amplifying gate thyristors
One technique to improve the dynamic performance of a thyristor, particularly
its di/dt capability, is to increase the gate current to a much higher value so that
the device is switched into conduction much quicker. If this is done by normal
means the gate power requirement is much increased and there is a serious
danger of damage to the gate of the thyristor. A more satisfactory method is to
make two thyristors on the same silicon slice and to arrange for the small one
to be fired and for its anode/cathode current to form the gate current for the
main thyristors. The small firing thyristor effectively amplifies the gate pulse to
the main thyristor, hence its name, the amplifying gate thyristor.
Many of the fast turn off thyristors and some of the convertor grade thyristors
are of the amplifying gate type.
2.2.3 Using thyristors in AC motor drive circuits
When using thyristors in naturally commutated circuits where the reversing sine
wave voltages allow natural switch-over to take place, it is only necessary to
correctly protect and fire the thyristors.
VDSM> V R S M
Voltage
rating
MDRM
VRRM
Surge
current
ITSM
On-stage
voltage
vT
I2t
1-2-1-5
1-5-2-5
1-5-2-0
1-5-2-0
-1-4
-*l-6
-2-4
->2-4
-3-6
-3-2
-•40
-+1-5
-1-8
-2-5
-2-5
-4-0
-3-5
-•4-4
10-50
50-100
100-300
300-600
600-1000
1000-1500
2000-2500
1-5-2-0
1-3-2-0
2-0-2-5
1-5-2-5
-1-2
-1-2
1-10
8-5-20-0
20-30
40-60
4-0-12-0
2-0-5-0
1-2-2-2
0-2-1-5
•01-0-2
360-2000
1000-4000
5000-18000
100-700
30-100
8-5-24
•24-11-25
•0005-0-2
•05-025
•015
•0075
01-05
0-2-0-1
0-5-0-2
2-4
50-2
°C/watt
Max
volts/^s
dv/dt
Max
amps//is
di/dt
or
module
Flat base 300-1000 100
or disc
Flat base 300-1000 150
or disc
Disc
300-1000 100
Disc
300-500 150
Disc
1000-2000 150-300
100
100
Screw
base or
module
Screw
200
200
base or
module
Flat or
200-1000 100
screwbase
Type of
Thermal
housing
resistance
junction/base
200-300 300-1000
200-400 400-1000
300-500 300-1000
200-300 300-1000
200
400
400
400
100-200 300-1000
200
50-150 150
200
40
20
Typical
mA
60-150
20-50
mA
200
100
Max
/xsecs
Turn
Gate Latching
off time current
and
to fire holding
Tq
current
§•
11
to
I
Po\
Amps mean
Peak
Peak
Volts at 10 ms peak lOmS, 125°C
at 85°C volts x 103 volts x 103 3 x IT amps x 103 amp2 sec x 103
base temp
Nominal
current
rating IT
Table 2.1 Range of available convenor grade thyristors
66
•1-15
•2-11
1-0-2-0
2-0-4-0
1-5-3-0
2-3
2-2-5
1-5-2-5
1-5-2-0
20-2-5
1-1-5
1-5-20
2-0-2-5
2-0-2-5
20-2-5
1-5-2-0
2-2-5
01-1.0
01-10
01-1-2
0-1-0-6
0-6-1-4
0-1-0-6
0-6-1-3
1-3-2-0
-1.2
1-2-2-0
-1-2
1-2-2-0
0-10
10-50
50-100
100-200
100-200
200-400
200-400
200-400
400-600
400-600
600-800
600-800
4-12
3-10
3-12
10-15
10-15
15-20
15-20
Amps/103
Volts
Volts/103
Amps (mean)
Surge
current
capability
10 mS
vT
On-stage
voltage
Repetitive
voltage
rating
max
Nominal
current
rating IT
85°C base
80-720
45-500
45-720
500-1125
500-1125
1125-2000
1125-2000
11-25-45
•05-12
0-2-6-5
5-20
20-80
A2sec/103
I2t
lOmsecs at
125°C
Table 2.2 Range of available fast turn-off thyristors
01-04
01-04
01-04
•04-03
•04-03
•03-02
•03-02
0-2-01
50-2
2-0-0-5
0-5-0-2
0-2-0-1
°C/watt
Thermal
resistance
junction/base
Screw base
Screw base
Screw base
Screw base
or disc
Screw base
or disc
Disc
Disc
Disc
Disc
Disc
Disc
Disc
Type of
housing
200-800
200-1000
200-1000
200-1000
200-1000
500-800
500-800
200-1000
200-500
200-500
200-500
200-500
Volts/^s
dv/dt
max
-1000
-1000
-1000
-1000
-1000
-1000
-1000
200-800
100-100
100-200
100-500
100-500
Amp///s
di/dt
max
10-25
10-35
20-50
20-40
30-60
20-40
30-70
10-25
5-25
5-30
10-25
5-20
/isecs
Turn-off
time Tq
300
300
300
400
400
400
400
350
100
100
150
150
mA
IGT
Gate
current
to fire
3'
68
Power switching devices
A naturally commutated thyristor switch
Fig. 2.8 shows a typical complete thyristbr switch for use in mains frequency
naturally commutated circuits. The fuse may or may not be included depending
on the degree of overcurrent protection required and on the other methods
which may be included to limit fault currents. If a fuse is used it will usually be
one designed specially for use with semiconductors, having a low peak voltage
during arcing.
The series reactor and the snubber circuit are for voltage protection, any high
transients occurring across the switch will be dropped across the reactor and the
R/C snubber will prevent them occurring directly across the thyristor. In some
cases non linear metrosil or varistor suppressors may be used instead of the R/C
snubber. The series reactor may be a specific item in the circuit, it may be ferrite
cores surrounding the cable or busbar or it may be the cables connecting the
switches together. If thyristor switches are used in parallel to increase the power
rating, the reactor may assist in ensuring equal sharing of the total current
between the parallel switches.
series
reactor
thyristor
pulse
transformer
D1
311
V
R1
snubber
circuit
R2
1
fuse
Fig. 2.8 A typical thyristor switch for use in a naturally commutated convertor
Gate firing
The thyristor will usually befiredby pulses of gate current and these are usually
passed to the thyristor via isolating pulse transformers so that the electronic
circuits can operate at a low potential to earth. Single pulses are suitable to fire
thyristors as long as it is possible to ensure that there will be a forward voltage
across the thyristor at the time when the pulse is applied. If there is any doubt
then a train of pulses will be needed to ensure that the thyristor willfireas soon
as the voltage across it is in the positive (forward) direction. It is not usual for
gate pulse to be applied during the reverse period because the reverse leakage
current can increase when gate current is applied. This is only of serious concern
Power switching devices
69
if thyristors are connected in series to increase the total voltage capability of the
switch.
The resistor Rl is used to decide the level of gate current to be fed to the
thyristor. Resistor R2 can be useful in increasing the dv/dt capability of the
thyristor, however if the thyristor is of the 'shorted emitter' type then the resistor
will be unnecessary as this feature carries out the same function.
The components Cl, C2 and Dl are included in the gate circuit to prevent
interference from causing misfiring of the thyristor. They are to prevent low
level interference pulses from firing the thyristor and to prevent interference
caused by the switching of the thyristor getting back into the firing electronics.
The earthed screen on the transformer also helps in this respect and this item is
particularly important with high voltage power circuits and sensitive electronics.
Fig. 2.9 This is a complete naturally commutated thyristor switch containing the thyristors on
the heatsinks, a series reactor at the back and snubber components and firing pulse
transformer items in the front of the module. These modules are designed for parallel
operation with other similar modules to achieve high powers. (G.E.C. Industrial
Controls, Ltd.)
The only item of significance not shown in thisfigureis the heatsink on which
the thyristor is mounted and which is used to remove the heat losses from it.
A switch of this type is normally used in circuits where the voltage across it
will be alternating positive to negative on a cyclic basis. The gate pulse will be
70
Power switching devices
applied at some time while the voltage across it is positive and the anode/cathode
current will naturally come to zero at some time during the negative half cycle
of voltage, maybe due to the switching on of other thyristor switches in the
circuit.
If a switch is required to operate in a circuit where the voltage does not
naturally reverse then some other means of bringing the anode current to zero
to turn-off is required. The process of forcing the switch-off of the thyristor is
known as forced commutation. Switches of the forced commutated type are
required in voltage and current source motor inverter circuits where the circuits
are only exposed to a DC source of power.
load current
/swi
load
thyristor current
capacitor
current
diode
current
thyristor voltage
point of closure of SW1
Fig. 2.10 Forced commutation switching
Power switching devices
71
Forced commutated thyristor switches
The principle of a forced commutated thyristor switch is for the anode/cathode
current to be temporarily by-passed through a capacitor while the thyristor is
allowed to regain its blocking ability. This principle is shown in Fig. 2.10 where
a previously charged capacitor is suddenly switched across the thyristor which
is carrying anode to cathode current in the inductive load circuit. The closure
of the switch causes the current to be diverted out of the thyristor, through the
capacitor. Initially the reverse voltage of the capacitor appears across the
thyristor until the flow of load current through it causes its charge to reverse.
If the time Tl is larger than the turn off time of the thyristor then the thyristor
will regain its blocking ability before the capacitor voltage reverses.
firing components
1
off signal
Fig. 2.11 A forced commutation thyristor switch
The diode across the load is to allow the load current to continue toflowwhile
the load inductive energy is dissipated, otherwise a very high voltage would be
induced in the load and this would cause breakover of the thyristor switch.
This is only a one shot switch because once the capacitor has been charged
72
Power switching devices
up to the supply voltage it is unable to repeat the turn off process. The complication in forced commutated switches is to reverse the capacitor charge so that
repetitive ON-OFF switchings of the thyristor can be done.
There are numerous circuits which will do this and it is not my intention here
to detail them because they are not directly relevant to AC motor drives. One
typical example will be explained to demonstrate the principles.
Fig. 2.11 shows such a switch designed for use from afixedvoltage DC supply
and for feeding an inductive or motor load. The main thyristor switch is shown
in the centre and the components to the right of it are those items needed to
protect and fire it and those on the left are the forced commutation switch off
circuits. In order to explain the principles the load is shown as inductive but with
an alternative path for its current through the parallel diode.
The components LI, Rl and Cl are for voltage protection as with the
naturally commutated thyristor switch and the gatefiringarrangements are also
similar. The area which needs explanation is the forced switching off.
T2 fired
•-+-/-
i-x
recharge
commutating
capacitor |
Fig. 2.12 Forced commutated voltages and currents
The capacitor C2 is the commutating capacitor and when charged as shown,
the switch on of thyristor T2 will initiate the turn off of thyristor Tl. When
thyristor T2 isfiredthe current previouslyflowingin Tl will be diverted through
L3, T2 and C2 (L3 is only a small inductance to limit the initial rate of rise of
this current, it may be just ferrite cores around the cable). The current will
Power switching devices
73
continue to flow into C2 until its voltage has risen to the same value as the DC
supply, when diode Dl will take over the load current.
The components D2 and L2 are included to allow the capacitor C2 to be
recharged back to the correct polarity for switch off to be repeated and this
recharging occurs when the main thyristor Tl is switched back on. Because the
charge on C2 is now reversed, the switching on of Tl causes the circuit C2, D2,
L2, Tl to be a closed circuit and C2 will circulate a current through D2 and L2
via Tl, this will be a half sine wave resonant current which will stop automatically
when C2 has fully reversed so as to be ready to again turn off Tl. The voltages
and currents which occur during this sequence are shown in Fig. 2.12. In this
case the time during which the capacitor C2 diverts the current from Tl and
maintains reverse voltage across it is shown. It varies with the level of current
flowing with the shortest turn off time being when the load current is high. The
thyristor Tl must recover its blocking ability during this time.
Such a switch as this can be opened and closed rapidly at a frequency decided
by the size of the commutating components C2 and L2.
When such switches are used in motor drive circuits they may not be fitted
with independent commutating components. It sometimes can be more economic
to use the same commutating components for the two switches in one phase of
an inverter bridge circuit.
2.3 The transistor
The transistor is a three terminal semiconductor device capable of carrying
current from its collector to its emitter only and the value of this current can be
controlled by the amount of current passed between its base and emitter
connections. It is not a switch like the thyristor but it is a continually controllable
device whereby significant current can beflowingthrough it at the same time as
a forward voltage is occurring across it. The voltage occurring between collector
and emitter is dependent on the amount of base to emitter current flowing and
the load impedance. When no base current isflowingthen collector to emitter
current will be negligible and the circuit voltage will occur across the transistor
(collector to emitter). As base current is increased the collector current increases
thus causing some of the circuit voltage to occur across the load and the
remainder across the transistor.
In its simplest form the transistor is a three layer semiconductor device with
alternate positive and negative charged semiconductor materials, Fig. 2.13
shows the NPN version most common for power switching duties, where the
voltage of the circuit occurs across junction Jl.
When used in inverter and variable speed drive circuits however, the transistor
is never used in its controllable mode with significant voltage and current
occurring in it at the same time. It is used as a switch in order to reduce the
power losses in the transistor itself. By using it in this way it is possible to control
74
Power switching devices
much higher levels of load power with particular transistors. It is therefore used
in either its switched OFF state whereby negligible current is flowing in the
transistor and it is blocking the current voltage, or in its ON state where a high
level of current isflowingthrough it and as low a voltage as possible is occurring
across it. These two conditions are shown in the figure.
collector
base
C
N
current
flow
direction
P
N
emitter
off-state
high
resistance
on-state
low
resistance
Fig. 2.13 Transistor principles
Transistors do not normally have any inherent reverse voltage withstand
capabilities and they are usually used in such a way that reverse voltage does not
occur. There is also one other very important difference compared to thyristor
switches; the level of base current necessary to achieve the ON state is large.
Base currents of at least one tenth of the collector current are often required in
transistors suitable for significant power switching applications. Hence very
much larger levels of base current and power are therefore needed to secure
good switching performance. The ratio of collector current to base current is
Power switching devices
75
known as the current gain and with bipolar switching transistors this may have
a value of between 5 and 50 under rated operating conditions.
Transistors are inherently fast switching devices which are capable of being
switched on and off in only a few microseconds with correct circuit design. They
can therefore be used at operating and switching frequencies much higher than
thyristors. The transistors I have described up to now are more commonly
known as silicon bipolar transistors and this type of device is usually capable of
operating at frequencies of tens of kilohertz.
Bipolar transistors have advanced considerably in recent years but their
power capabilities are still well below those of thyristors. Peak voltage capabilities
are limited to the order of 1200 volts and maximum continuous collector current
ratings of up to 1000 Amps can be obtained. In general this means that
transistors can be used in AC motor drives of ratings of up to a few hundred
kilowatts operating at mains voltage of up to 500 volts RMS line. Whether this
range will be increased during coming years depends on the progress that is
made in the area of gate turn off thyristors which are at present seen to be able
to satisfy the higher power drive rating needs. There will clearly be progress
in power transistors because of their considerable superiority in switching
frequency but whether this will cause increases in power, current and voltage
capability cannot be predicted with accuracy.
Although the bipolar transistor is the most significant device used today for
motor drive applications there is another one which is gaining interest. It is the
power metal oxidefieldeffect transistor or MOSFET which has come about due
to the relatively low current gain of bipolar transistors and the wish to reduce
the power of base drive circuits. This device is the power version of the field
effect transistor and the current in it can be varied by changing the voltage
applied to its gate control connection. The result is a very high gain device which
can be switched very quickly so that it can be used at frequencies in the
megahertz region. The main factor which has limited its use in the inverter drive
field has been its lower voltage capability and the relatively high value of ON
state resistance and therefore power loss.
From the physical point of view power transistors in general look very similar
to thyristors. It is necessary to mount them onto heatsinks to dissipate the
internally generated heat and hence the type of sealed enclosure used will depend
on the power rating of the device, screw base, TO3, flat base and double sided
cooled capsule designs are all obtainable.
As with all semiconductor devices, they have absolute limits of capability
which must not be exceeded or else failure occurs. It is essential to understand
them fully and to know how to protect them if safe and reliable operation is to
be obtained.
2.3.1 Capabilities and performance of transistors
Voltage capabilities
Transistors are only able to block voltage in one direction, with the collector
76
Power switching devices
positive with respect to the emitter in the case of NPN devices. The highest value
of forward voltage can be withstood in the OFF state if a small reverse voltage
is applied between the base and emitter, i.e. emitter positive with respect to the
base. Under the off state condition a small leakage current willflowthrough the
transistor and its value varies significantly with temperature. Such leakage
current values will vary from fractions of a milliamp to 10 milliamps for the
larger higher voltage devices.
Fig. 2.14 This shows a full range of transistor silicon slices and completed devices. {Marconi
Electronic Devices, Ltd.)
If the maximum collector-emitter voltage (VCEX) is exceeded even for very
short periods of time then the transistor will be damaged.
When used for switching purposes the voltage which can be applied across the
transistor particularly immediately after current flow has to be restricted to a
lower level known as the collector emitter sustaining voltage (VCE(sus)), the value
Power switching devices
77
which can be accepted for identifiable periods of time. This value may be
between 15 and 40 per cent below the maximum possible VCEX value and it is
restricted to this value because of the heat dissipation which can be caused by
the residual currentflowingagainst this blocking voltage soon after conduction.
The circuit voltage must be kept within this VCE(sus) maximum value if failure
during switching is to be prevented.
The base to emitter junction of a transistor is a low voltage one and it is
usually only capable of sustaining a reverse voltage of between 5 and 10 volts.
This has a direct effect on the design of the base drive circuitry.
Due to the variation in switching characteristics and leakage currents between
transistors it is not practical to consider the use of transistors in series in order
to sustain higher levels of voltage. Therefore the maximum circuit voltages in
which transistors can be used is limited by the capabilities of the transistors
themselves.
Current capabilities
The transistor is a current controlling device in the sense that variation of the
base current can directly alter the collector current. When used as a switch the
principle is to drive the base with a relatively high current so that the maximum
collector current flows, so dropping the whole of the circuit voltage across the
load which is effectively in series with the transistor. The value of voltage which
then occurs across the transistor — the ON state voltate drop or collector to
emitter saturation voltage — will then vary with the level of collector current
flowing in the device and with the junction temperature. There is also significant
variation between different transistors. This is shown on the upper graph of
Fig. 2.15 which shows the saturation voltage characteristics of the transistors of
a particular type reference. From this you can see that there can be a two to one
spread in forward voltage drop (VCE(sat)) between different thyristors and
operation at junction temperatures in excess of 100 degrees Centigrade can
cause another doubling of the value.
Whenever transistors are used in motor drives they are operated in the
saturated region with the minimum forward voltage drop. From Fig. 2.15 it can
be seen that in deciding the rating of a transistor the maximum value of VCE(sat)
will have to be used to ensure that all transistors will be within their maximum
temperature rating. As a consequence some transistors will run a lot cooler than
the limiting volt drop device.
Fig. 2.15 also shows the other important feature of the ON state transistor,
namely the fact that the current gain, the ratio between collector current and
base current, reduces as the collector current increases. In this case the current
gain has reduced to seven at the rated collector current. For this reason it is
conventional for transistors to be rated at their peak current values rather than
with thyristors where the mean current is usually referred to as the rated value.
So a transistor with a rated current of 100 amps can usually only be used at
mean currents of 30 to 50 amps when used in the three phase bridge circuits
needed for variable speed drives.
78
Power switching devices
Fig. 2.16 shows the importance of the base current and the changing gain to
the operation of transistors. In motor drive circuits the current demanded by the
motor and hence the current which passes through the switches depends on the
level of motor torque, etc. If the collector current ever happens to exceed the
level dictated by the base current then the transistor will come out of saturation
and a considerable voltage will appear across the collector to emitter and the
result will be a sudden large increase in heat in the transistor. The consequence
2.0
3
3
10
20
30
40
col lector current , I c - amps
50
30
20
10
10
Fig. 2.15
20
30
50
Transistor on-state curves
can often be sudden failure due to over temperature. For example, in Fig. 2.16,
if the collector current tries to rise above 23 amps when a 1 amp current is being
injected into the base then the collector/emitter voltage will rise to a very high
value. If the base current is 5 amps then a collector current up to 52 amps would
be acceptable with the transistor remaining in saturation up to this level. In
Power switching devices
79
other words, the base current has to be chosen to correspond to the maximum
value of current which is ever expected to occur. It also has to correspond to this
current flowing through the transistor with the lowest gain value. This means
that to achieve a currentflowof 50 amps using the transistor with characteristics
as Figs. 2.15 and 2.16 it is in fact necessary to input a base current of over 8 amps
to make sure that it never sees more than the saturation voltage of approximately
two volts at say a junction temperature of 120 degrees Centigrade.
I c /I B curve
saturation curve
r
r
50
40
IB= Samps
'B=
3
/
s^
I B = iamp
20
10
f
collector/emitter voltage VCE
I
I
I
'
I
I
I
I
I
[
I
1_
to
1
2
3
base current I g
A
5
Fig. 2.16 Base current needed to ensure saturation
Because of this base to collector current relationship the transistor cannot be
seen to have any significant overload capacity above the design ratings. If the
current goes too high then a large energy loss will occur in the device and it will
cause failure. The only sensible way of protecting against overloads is to arrange
for the transistor to turn the current off when it reaches a limiting level.
Like all semiconductor devices transistors have a limiting junction temperature above which they will fail to work correctly, 150 degrees Centigrade is a
typical maximumfigure.The maximum power dissipation is therefore dependent
on the device thermal resistance between the junction and the heatsink surface
and the effectiveness of the heatsink. The allowable temperature of the transistor
to heatsink surface therefore has to be reduced as the wattage dissipation
increases as shown in Fig. 2.17 for a range of different sized transistors.
80
Power switching devices
When assessing the thermal circumstances it is necessary to take account of
all the losses which occur in the transistor.
1) The collector current losses caused by the ON state saturated collector
vrjltage VCE(sat) as discussed above.
2) The energy loss caused by theflowof base current. The base to emitter
junction causes a voltage drop of typically 1 to 2 volts maximum
resulting in an additional power loss. Most manufacturers provide
curves of the base to emitter saturation voltage VBE(sat) for use in
estimating these losses.
3) The switching losses caused by the voltage and current transitions
from the OFF to the ON states and vice versa.
1000 r
device junction/base
thermal resistance
c
o
a
a.
1 500
•6
100
heatsink or case temperature *C
150
Fig. 2.17 Thermal ratings
Darlington transistors
The low gain and the high base currents required with transistors has led to
the use of cascaded transistors in the Darlington configuration as shown in
Fig. 2.18.
With this approach the total gain of the pair of transistors can be approximately equal to the individual gains multiplied together and hence minimum
gains of 30 to 50 are possible. Both transistors may be made on the same slice
or incorporated together in one housing but it is possible to use two individual
transistors in the same way to achieve the same objective.
Power switching devices
81
Fig. 2.18 The Darlington connection
Switching characteristics
TURN ON
A transistor can be turned on by applying a current to the base sufficient to
cause the transistor to become saturated. To do this the base current must be
larger than that required to match the likely collector current on switch on (see
Fig. 2.15). The transistor will not however turn on instantly even when the base
current is applied very quickly and there will be a short period of time while the
collector current is rising and the voltage VCE is falling. During this time a
significant energy loss will be caused by the switching action. The period of the
turn on will depend on the level of the applied base current, a high level reducing
the time to turn on and consequently the energy loss occurring.
TURN OFF
If the base current is removed the transistor will switch from its saturated ON
state into the OFF state. Again it does not do this instantly and there will be a
period during which the current is falling and the voltage is rising. It is preferable to allow the base current to reverse during this switching time.
Fig. 2.19 shows these two conditions in a typical transistor.
The application of the base current is followed by a short delay time and then
the current rises rapidly to the value dictated by the circuit. When the base
current is removed initially some of the collector current flows as a reverse
current in the base to clear out free carriers from the silicon material. This is
followed by a period when the current falls rapidly and the collector emitter
voltage rises. The peak losses to occur during the switching ON and OFF can
be very large compared to the normal saturated losses and they can damage the
82
Power switching devices
transistor. For example the peak losses during switching can often be ten times
the level of ON state saturated losses caused by the normal circuit current. It is
therefore essential that the switching times are short so that that energy and
therefore heat caused by these switching losses are minimised. It is also essential
that the voltage collapses quickly on turn ON and that the current reduces
quickly on turn OFF.
on
off
off
collector /
emitter
voltage
power
dissipation
^V^rise time
collector
current
delay t ime
base
current
\
(
!
V
c1
fall time
8.
\
\
f
Ir
\
/
Fig. 2.19 Transistor switching waveforms
The transistors capability during switching is usually expressed in the form of
a collector voltage/collector current graph which indicates the safe operating
area (SOA) of the transistor. Fig. 2.20 is typical of power switching transistors
SOA curves. All points remote from the horizontal VCE axis and the vertical Ic
axis will refer to points of high loss (power loss being volts times current) and
the times for which the transistor can accept these losses are shown by the one
millisecond, 100 microsecond and one microsecond curves.
These curves are used by showing on them the locus of collector current and
voltage during the transition from the OFF point A, to the ON point C and vice
versa. The dotted A—B—C curve shows a good turn ON curve because the
voltage reduces before the current rises. The curve C—B'—A' is not such a
good turn OFF curve because the current does not reduce quickly enough. It is
possible with poor circuitry design for the transistor to traverse through very
high loss areas. For example, if on switching OFF the circuit current cannot be
reduced quickly the transistor voltage may rise to the full circuit value before the
Power switching devices
83
current has reduced significantly giving the curve C—P—A. It is even possible,
owing to switching voltage transients and the discharge of capacitors, for the
current/voltage locus to move completely outside the SOA and immediate
damage would be the result.
1000
IMS
10
collector/emitter voltage
100
OFF
1000
(SUS)
Fig. 2.20 Transistor safe operating area
2.3.2 The available transistors
Transistors can be obtained with voltage capabilities of up to 1200 volts and
with continuous current ratings of up to 1000 amps. The higher current ratings
are not available above 400 to 500 volts. There are a limited number of suppliers
and in general only a part of the available range can be purchased from one
supplier who may specialise in the lower power ratings, the high current ratings,
the high voltage ratings, etc. Although there has been steady improvements in
the ratings available over the past ten years the range of available sizes has not
yet fully stabilised.
Table 2.3 shows approximately what is available in transistors suitable for AC
motor drive use in 1986/7. The typical switching times show that these devices
are capable of being switched ON in up to three microseconds and being
switched OFF in between 5 and 12 microseconds.
1000
1200
550
400
-120
-360
-600
-1000
-1200
50-100
100-300
300-500
500-800
800-1000
1200
1000
-60
500
300
1000
1000
850
850
5-8
5-8
6-10
4-8
5-10
5-10
5-10
700
1000
10-50
HFE
min value
Gain at
rated I c
Volts
V CE
(sus)
Volts
-12
Amps
Amps
continuous
VCE
max
1-10
Peak
collector
current
Nominal
collector
current
1-25
10
1 25
1-5
20
20
20
Max value
at rated I c
VCE (sat)
1-75
20
1-75
1-75
•1-05
•1-05
•15—05
•2-08
•2-1
20
•8-1-5
1-2
3-7
3-7
5-8
6-10
2-3
2-3
2-3
1-2
1-2
1-2
1-1-5
3-7
1-2
1-1-5
3-7
0-5-1-5
10-65
20
•5-1
2-5
0-5-1-5
10—65
20
fiS
/IS
TF max
T s max
TON max
Fall time
Storage
time
Turn on
time
/iS
Screw base
orTO3
Screw base
orTO3
Flat base
or disc
Flat base
or disc
Flat base
or disc
Disc
Disc
Type of
housing
°C/watt
Thermal
resistance
junction/case
Max value
at rated IB
VBE (sat)
Table 2.3 Range of available NPN silicon bipolar transistors (1986)
I
I"
CO
I
i
Power switching devices
85
Darlington transistors
Complete Darlington transistors in one package are available from some sources
and such units can operate at up to 100 amps continuous and up to 850 volts
VCE(sus). Characteristics vary but gains can be in the range 30 to 50 and
switching times are approximately double the values applicable to equivalent
single transistors. Some manufacturers will include in the package the additional
components which are found to be beneficial in stabilising the transistors and
minimising the switching times. The two most popular arrangements are shown
in Fig. 2.21. Circuit (a) gives good overall stable performance and circuit (b)
usually has a significantly shorter turn off time due to the speed up diode.
-nFig. 2.21 Improved Darlington circuits
MOSFET transistors
Power MOSFET's are only available from a limited number of suppliers and in
a limited range of ratings. 20 amps rating can be obtained at up to 100 or so volts
86
Power switching devices
and 500 volt ratings can be obtained with a few amps. Turn off times are very
short ranging from 005 to 0-2 microseconds making them suitable for use at
frequencies of more than 100 times those applicable to bipolar transistors.
2.3.3 Using transistors in AC motor drive circuits
Transistor switches are used mainly in pulse width modulated inverter systems
as described in Chapter 5. Their power ratings are such that they can be used
for drives of up to say 200 KW, above this rating gate turn off or forced
commutated thyristors are found to be more suitable. They are now used instead
of forced commutated thyristors in this lower power range and their use is
restricted to the ON-OFF switches required in DC link series choppers or in the
motor inverters. They are therefore mainly used as the switches in the three
phase inverter bridge circuits which are fed from a DC link supply and which
directly feed the motor stator windings. They are only used in voltage source
systems and hence the switches always have a reverse diode across them to take
the reactive current and any regenerative energy.
Switching protection
When transistors are used in motor drive circuits it is essential to ensure that
they can be fully protected at all times. One important area is during switching
when it is possible for the transistor to be forced to move outside of its safe
operating area. During turn on of the transistor the current is transferred from
one of the bypass diodes into the transistor and the characteristics of the diode
can cause problems to the transistor; when the diode turns off, its current
reverses and this causes a peak of current to flow through the transistor. If
the diode is slow in turning off it can cause transistor SOA failure. Reference to
Fig. 2.20 shows that for minimum switching losses and to keep within the safe
operating area it is preferable to ensure that
at turn on the VCE collapses quickly and the collector current rises slowly,
at turn off the current collapses quickly and the voltage rises slowly.
The switching aid circuits shown in Fig. 2.22 enables these objectives to be
achieved and such circuits are a small price to pay to ensure safe and reliable
operation at economic transistor power ratings.
Circuit (a) causes the turn off current to reduce quickly by diverting it through
diode Dl into capacitor Cl as the voltage rises. The resistance Rl prevents
discharge of Cl into the transistor on switch on.
In circuit (b) the inductance L reduces the rate of rise of current on switch on
and R2 and D2 prevent high transient voltages due to the snapping off of the
current in Dl and L on switch off. Dl and Cl assist turn off as in (a).
Transistors can be used singly if their rating is appropriate to the system, if
higher power is required then they can be operated in parallel to increase the
circuit current. Due to the great variability in switching performance it is not
practical to connect them in series to increase the system voltage and hence
transistors are limited to use in circuits operated at no more than 600 to
800 volts DC.
Power switching devices
87
Fig. 2.22 Transistor switching aid circuits
Parallel operation
Transistors can be operated in parallel to increase the circuit current rating and
when doing this it is necessary to take steps to ensure that the total current is
shared reasonably between the individual transistors during switching and
during normal conduction.
The simplest method is to closely select the transistors so that they all have
similar parameters and then allow for the small imbalance which may occur.
From the steady state point of view the VCE(sat)/Ic characteristic would need to
88
Power switching devices
be matched and the VBE(sat)/Ic characteristic if all the transistor connections are
to be directly paralleled — this also implies that the gain characteristics would
also need to be matched. Clearly the degree of matching necessary, will depend
on how close to their maximum current ratings the individual transistors are to
be used.
select:VCE(SAT)/IC
VBE(SAT)/IC
storage time
AI =
Fig. 2.23 Parallel operation of transistors
AV B E
Power switching devices
89
When parallel transistors are switched they must all switch ON and OFF in
similar times if the current balance is to be maintained transiently. Hence
matching of the turn on times and storage times would be required. Again the
degree of selection of these parameters depends on how closely the transistors
are being used, if the imbalance during switching is significant it can cause
individual transistors to have to operate outside their safe operating areas so
causing them to fail.
If the extra power losses can be accepted then correct sharing of current can
be ensured by connecting a resistance in the emitter connection to each transistor.
This will directly share the collector currents and also tend to equalise the base
impedances. This method is not often used because of the extra losses involved:
it is preferable to derate the transistors. An alternative is to accept the VCE(sat)
variations but ensure base current sharing by putting series resistors in the base
circuits. It is usual to fit switching aid circuits to the paralleled transistors to
ensure that all transistors are always kept within the safe operating areas during
switching. Fig. 2.23 shows the methods normally used to parallel transistors.
Base drive circuits
The drive circuit to the base of the transistor switch is very important to the
performance of the switch, it enables the transistor to be used at its optimum
rating and it can be used to protect the transistor against excessive load current.
To achieve the most satisfactory switching performance from the transistor a
high base current
during turn -on
to minimise turn-on losses
3to5uS I
turn -off
reverse base
current reduces
storage time
turn _on
Fig. 2.24 Ideal base current waveform
90
Power switching devices
base drive current waveform as shown in Fig. 2.24 is needed. The base drive
current as explained earlier is required during the whole of the conduction
period to ensure saturation of the transistor and an initial higher peak of current
can help the transistor accept any switch on peak caused by bypass diode
recovery or capacitor discharge. To turn the transistor off it is necessary to
reverse the base voltage and allow the initial flow of reverse current while the
transistor is recovering its OFF state. After recovery the voltage blocking ability
of the transistor can be enhanced if a few volts negative are kept on the base
during the OFF period. Fig. 2.25 shows a typical circuit to do this.
Fig. 2.25 Typical base drive system
The switching ON of transistor Tl will apply the base current to the transistor
via resistor Rl to switch it on; T2 is held off during this period. When turn off
is required, Tl is cut off and T2 switched on, this allows the reverse current to
flow and when the current in T has stopped, the reverse current willflowthrough
the diodes Dl to keep a reverse bias voltage on the main transistor T. The
Power switching devices
91
transistor T3 is switched on and off to turn the main transistor on and off
respectively.
The base drive current therefore needs to have positive and negative supply
voltages available with respect to the emitter of the main transistor and the
whole of the base circuit will be at the potential of the main transistor. The
base circuit will therefore need to be fully isolated and provided with its own
independent power supplies.
A typical transistor switch
The complete transistor switch for use in a motor drive circuit will therefore
consist of a combination of the above mentioned items. Fig. 2.26 is such a switch
showing the transistor which may be a simple transistor, a parallel group of
transistors or a Darlington arrangement. A switching aid circuit may be included
to enable optimum transistor operation. The base drive circuit as shown includes
inputs from the main transistor collector and emitter, the purpose of these is
likely to be to protect the transistor against overcurrents. If the transistor tries
to come out of saturation while it is conducting, the base drive circuit is likely
to be arranged so that the transistor will be immediately turned off to avoid
damage to it. A measurement of the collector/emitter voltage during the ON
period enables this condition to be detected.
on/off
signal
base drive
system
see 2-25
switching
aid circuits
see 2-22
Fig. 2.26 A typical transistor switch
2.4 Gate turn off thyristors
The Gate Turn Off Thyristor (GTO) has many similarities to the thyristor as
already described but the achievement of turn off from the gate has led to
compromises on other parameters. It is also a device which is still undergoing
92
Power switching devices
development and improvements in its capabilities are to be expected during the
next few years.
Like the thyristor, it has two stables states, the ON state and the OFF state.
However, in the GTO these two states can best be maintained by the application
of a small gate current and a reverse gate voltage respectively. The other
significant difference is that the majority of GTO's available at present only have
a very small reverse voltage capability; this is one of the factors which may
change in the near future.
As with the thyristor the GTO can be switched into the ON state by the
application of a relatively small gate current (usually larger than needed with a
thyristor) which triggers the device into conduction. Once the device is conducting the presence or not, of forward gate current has only a secondary influence
on its performance.
The principle additional feature of the GTO is that if the gate voltage is
reversed and a significant reverse current is allowed to flow in the gate then it
is possible to alter theflowof charge carriers in the silicon and allow the device
to revert to its OFF state. A substantial level of reverse gate current is needed
to achieve this but the energy level required is very small compared to that
involved in forced commutation of ordinary thyristors. When correctly turned
off, GTO turn off times can be in the order of 10 to 50 microseconds.
The GTO is still a four layer semiconductor like the thyristor, able to carry
current only in one direction, but to achieve turn off each device is made up of
many small GTO thyristors in parallel on the same silicon slice. Fig. 2.27 shows
the comparison between the cathode surface of a normal thyristor and a GTO.
The normal thyristor has a central gate area and a large portion of the cathode
area is remote from the gate. If an attempt is made to turn such a device off using
the gate only the area very near to the gate would be affected. In the GTO
therefore, the gate is made to surround many small cathode 'islands' so that it
is capable of affecting all areas of the cathode quickly and effectively. Clearly
this leads to more complicated and accurate manufacturing methods and to a
reduction in the effective cathode area available on a particular size silicon slice.
This construction also leads to an alteration in the performance parameters, for
example, a higher positive gate current is required to make sure all the 'islands'
turn on and steps have to be taken to make sure that some of them do not turn
off at low current levels.
The achievement of good GTO performance is now even more dependent on
the peripheral components used with them. A high quality snubber circuit is
essential because during turn off time the anode current is diverted into the
parallel connected capacitor. It is also essential to use fast diodes in association
with the GTO's to ensure optimum performance and protection.
The range of available GTO's is extending all the time, at present, units
capable of blocking over 4000 volts and also capable of turning off anode
currents of over 1000 amps are available from a number of sources. There is no
doubt that the range of devices available will continue to be expanded.
Power switching devices
93
centralgate
normal thyristor
gate
cathode 'islands'
gate turn off thyristor
Fig. 2.27 GTO internal design
2.4.1 The capabilities and performance of gate turn off thyristors
Voltage capabilities
All GTO thyristors are capable of blocking high forward voltages and some are
also able to block reverse voltages at similar levels. There are in fact two
different methods of making GTO's with the necessary turn off qualities. One
method is the use of anode emitter short circuits which allow the free carriers
in the N base to discharge quickly; unfortunately this method prevents one of
the junctions from blocking the reverse voltage and such devices have very little
reverse blocking ability. The alternative is to achieve control over the free
carriers by doping the silicon with heavy metal; in this case the junction retains
its reverse blocking ability and such devices have high levels of reverse blocking
ability.
94
Power switching devices
Reverse blocking GTO's can be used in all circuit arrangements but those
with only forward blocking ability can only be used in systems which allow a
reverse diode to be connected across the GTO or with an additional series diode
to take the reverse volts.
At present only a limited range of reverse blocking devices are available but
no doubt more will become available in due course.
As with thyristors, GTO's can be switched into the ON state if an excessive
forward voltage is applied to it; but a GTO is much more likely to be directly
damaged by doing this as the initial current flow is likely to be concentrated on
only one of the cathode 'islands'.
The GTO is also susceptable to high dv/dt and again if excessive dv/dt is
applied causing the device to switch on, it is likely to be irreparably damaged.
However, in general, a higher dv/dt capability is essential for correct switch off
and dv/dt ratings of GTO's tend to be higher than those for normal thyristors.
From other voltage points of view GTO's have similar characteristics to
thyristors and have to be treated accordingly.
Current capabilities
Due to its design and construction GTO's will have a larger voltage drop while
carrying current in the ON state than normal thyristors: values of twice those
of thyristors are not unusual. There is still a wide variation of voltage drop
between individual devices and the value changes with temperature and anode
current. As the junction temperature limits are similar to normal thyristors,
GTO's are therefore more critical from the thermal point of view and to achieve
the maximum current ratings a higher level of cooling is needed. As GTO's are
in general made for relatively high current levels double sided cooling and more
effective cooling are more common than with thyristors.
Thermal conditions are not the only concern with GTO's; all GTO's have a
specific maximum value of current which can be turned off using the gate. If an
attempt is made to turn off a higher level of anode current than this critical level
then the GTO will be damaged permanently.
On the other hand, the GTO does have the ability to accept quite high levels
of fault current without being damaged as is the case with normal thyristors, as
long as no attempt is made to turn this higher level off using the gate. Single half
cycle peak current capabilities (ITSM) often times the maximum turn off current
are typical with GTO's.
The important parameter with a GTO, therefore, is the maximum level of
anode current which can be turned off using the gate and as long as the junction
temperature is kept below the critical value (usually 125 degrees Centigrade)
control over the device can be reliably maintained.
The switching power losses in a GTO are however more significant and they
have to be allowed for in the thermal calculations.
Because of the construction previously described the speed of turn on of the
GTO is much quicker than with conventional centre gate thyristors. The current
Power switching devices
95
only has to spread for a short distance from the gate before the whole of the
device is turned on. The result is that GTO's can normally accept higher levels
of initial di/dt without damage. Their capabilities are in general comparable
with fast turn off thyristors. However, as always, if their capabilities are
exceeded damage will result.
Switching characteristics
Turn on
This is very similar in a GTO to that in a conventional* thyristor. However the
amount of gate current required is increased due to the much larger gate area
but this is to some extent compensated for by the higher di/dt capability. The
GTO turns on quicker due to the 'island' structure and the rise time is shorter.
Once the device is fully in the ON state it is possible to remove the gate current
and the device will stay in that condition. However, because the GTO is made
up of a lot of small GTO's in parallel, if the anode current is not high some of
the GTO's may turn off due to too low a holding current. This may not matter
if the current stays low but if it could increase due to circuit conditions the
delay
time
anode / cat hode^voltage
^
anode
current
L
storage total turn-off
time
time
gate current
initiation
of turn -on
turn-off
Fig. 2.28 GTO switching
96
Power switching devices
remaining parts in conduction could be overheated. It is therefore worthwhile
continuing to pass a small forward gate current during the conduction period
to make sure that the whole of the device remains in the ON state.
Fig. 2.28 shows the switching conditions of a GTO and this shows that at turn
on the current is likely to overshoot above the nominal level due to the effects
of bypass diodes and the discharge of snubber capacitors.
The turn on of the GTO initiates the discharge of the snubber capacitor
(which is essential to GTO turn off) and to avoid problems during turn off, it
is necessary to ensure that the capacitor is fully discharged before turn off is
initiated. Hence the minimum ON time allowed depends on the design of the
snubber circuit. The losses during turn on will depend on the rate of rise of the
current; they can be reduced by using a di/dt reactor in series with GTO anode
to slow up the rate of rise of current.
TURN OFF
The application of a reverse gate voltage will cause the GTO to turn off and
initially it is necessary to remove the charge carriers from the junction. This is
done by drawing reverse current out of the gate as shown in Fig. 2.28, the rate
of rise of this gate current is important to achieve clearing of the charge carriers
by the time the reverse gate current is at a sufficient level to turn off the device.
Once this has been done the device quickly turns off as long as there is an
alternative path for the anode current to flow in. This alternative path is the
snubber capacitor and it is arranged so that the current can quickly transfer by
having a direct path into a low inductance, low resistance capacitor. This
capacitor will also decide the rate of build up of the forward voltage during this
turn off period.
GTO
Fig. 2.29 GTO snubber circuit
Power switching devices
97
It is normal to use a polarised snubber circuit with a GTO and that shown
in Fig. 2.29 is the most usual arrangement. When the GTO turns off the circuit
current is temporarily diverted into the capacitor via the diode, and the capacitor
charges at a rate decided by the current flowing and the size of the capacitor.
The value of C is therefore decided by the allowable dv/dt across the GTO
during turn off.
GTO snubber circuit
The energy thus stored in the capacitor is discharged into the GTO when it
switches on and R is included to limit the di/dt occurring. For satisfactory
overall operation all the energy in C has to be discharged via R during the on
time of the GTO and this will decide the minimum on time. The presence of C
reduces the turn off losses in the GTO but the necessity to discharge it during
the on time causes significant losses in the resistor R.
The current in the GTO will not immediately come to zero, there will be a
small level of tail current which will take some time to disappear. Once the
anode current has reduced the reverse gate current will also reduce to a low level
during the tail period. A voltage overshoot on switch off is likely due to the
circuit inductances and capacitors.
Gate voltages of between 15 and 30 volts are needed during the initial turn
off period when reverse gate current is high but once the device has turned off,
the application of a few volts negative to the gate can ensure the optimum
voltage blocking capabilities.
Gate drive requirements
The performance of the gate drive circuit is crucial to the achievement of
optimum GTO performance. A suitable gate drive needs to take account of:
1) A high forward gate current of 10 to 25 amps may be needed to turn
the GTO on quickly. It has to rise to this level in a short time, say,
one to two microseconds.
2) During the remainder of the on period the gate current must be
reduced to a lower level to ensure the device stays on and to minimise
gate losses.
3) During turn off the reverse gate current has to be made to rise up
steadily to the necessary level to turn the device off during the storage
time of the device.
4) As the anode current falls quickly the gate current has to be allowed
to collapse quickly without high voltages being induced into the
system.
5) A negative voltage of up to 10 volts should be applied for the remainder
of the off period to ensure optimum blocking capability. Only a small
gate current will flow.
6) The signals of the gate drive circuit will be low level electronic signals
whereas the GTO will be at the power circuit potential. Isolation is
98
Power switching devices
required between these two. The gate power requirements may be fed
to the gate via transformers or direct feeding of the gate current via
transistors operating at the GTO potential may be used with isolated
DC power supplies to the individual gate drives.
Fig. 2.30 This photograph shows a wide range of gate turn off thyristors and the si/icon s/ices
which go into them. (Marconi Electronic Devices, Ltd.)
2.4.2 Available gate turn off thyristors
As GTO's are still developing it is better to refer to manufacturers' published
literature for the latest in this respect.
However, Table 2.4 gives details of a typical range of GTO's in order to show
the range of capabilities available and to give some indication of the typical
values of the many parameters.
350
600
1400
800
800
1200
200
450
600
1600
150
600
1800
2500
18
70
270
125
250
500
400
400
700
70
230
420
800
25
250
600
800
IT
ITGQ
50
200
600
1200
1200
4500
4500
1200
1200
2500
1200
1300
2500
2500
1200
1600
2500
1200
1200
1600
180
500
6000
1500
2500
4000
6000
5000
10000
31
3-8
2-5
2-8
3-2
3-2
20
2-3
2-5
3-8
2-5
3-2
30
3-5
2-5
2-5
30
Low
Low
Low
1000
1400
2000
650
1250
100
15
15
15
15
Low
Low
Low
Low
400
4000
10000
16000
500
2500
5000
8000
X
"RRM
TSM
Max
On state Surge current
RMS
Max
Max
current turn off forward reverse voltage
10 ms
current blocking blocking
voltage voltage
Table 2.4 Some of the available gate turn off thyristors (1987)
dv/dt
250
500
250
250
200
200
200
200
100
100
100
400
600
500
200
200
200
1000
500
500
500
500
500
500
500
350
350
350
600
1000
1000
1000
1000
600
Amps//is Volts//is
di/dt
40
90
120
320
180
180
220
70
120
280
Amps
8
10
100
50
6
15
20
20
18
18
20
8
8
20
6
6
15
TGQ /is
4
10
10
10
12
12
15
5
6
10
4
4
10
TG T
Gate current Turn off Turn on
to turn off
time
time
ITGQ
CO
CO
|
Co
CD
I"
s.
1
to
1
100
Power switching devices
2.4.3 Using GTO's in AC motor drive circuits
Gate turn off thyristors are mainly used in high power pulse width modulated
voltage source inverter drives as described in Chapter 5. This circuit does not
require any reverse voltage capability in the switch because of the presence of
the bypass diode.
on/off
signal
Fig. 2.31 A typical GTO switch
A typical switch for such an application is shown in Fig. 2.31. The GTO will
be provided with a substantial snubber circuit to assist turn off and a series di/dt
reactor is shown to limit the rate of rise of current originating from the remainder
of the circuit. Each GTO will be provided with its own gate drive circuit which
will provide the necessary positive and negative current pulses and voltages to
the gate as well as isolating the electronic ON/OFF signal from the high
potential of the GTO.
Fig. 2.32 shows the basis of a direct drive GTO gate drive circuit. The ON
drive circuit provides an initial high positive pulse from Cl when transistor Tl
is switched ON. After the initial discharge of Cl, R2 will control the level of gate
current during the remainder of the ON period. It will also control the recharge
of Cl before the next ON pulse is required.
The OFF circuit is initiated by switching T2 into its conducting state; the
charge on Cl is then available to produce the necessary reverse current via LI.
Once the GTO turns off the reverse gate current suddenly drops and the current
in LI circulates via the diodes Dl which maintain a reverse voltage on the gate
during the OFF period.
With this type of arrangement the signals controlling Tl and T2 have to be
isolated usually with opto-isolators and the DC energy sources shown as
batteries would be transformer isolated power packs.
Alternative arrangements where the gate is connected to similar ON and OFF
circuits via transformers are possible so that the transistor switch circuits can be
Power switching devices
101
fed from common power supplies. The transformers needed are difficult to
design due to the high rates of change of current required and the circuits are
more difficult to understand at this level. However the objective and basic
operating principles are very close to those described here.
One factor of the utmost importance in GTO switches is the mechanical
arrangement and the necessity for all the items of the switch, i.e. snubber and
gate drive circuits, to be situated very close to the GTO. Because of the high
currents which flow in most of the components of the switch and the fast rate
the current is diverted even small values of stray inductance can have very
serious consequences, causing high transient voltages and preventing the snubber
and gate drive from performing satisfactorily.
on signal
4^
off signal
—
4=ci ° N (1 R I
supply
u
GTO
1 opto I — | T 2
C2
off
D1
supply ±
Fig. 2.32 A typical GTO gate drive arrangement
Overcurrent protection
As mentioned previously GTO's cannot be turn off if the anode current has risen
above the controllable turn-off level. There are therefore potentially two
methods of ensuring satisfactory overcurrent protection of GTO's, these are:
1) If an overcurrent is detected, then inhibit the gate drive so that it does
not try to turn the GTO off, and then rely on fuses or crowbar systems
to remove the fault before the current in the GTO has reached its
short time overload capability indicated by its ITSM rating.
2(a) Use the GTO at a maximum normal current significantly lower
than its maximum controllable turn off level.
(b) Include inductance in the power system so that the fault current
cannot rise too quickly.
(c) Turn the GTO off as soon as the excessive current is detected.
If correct design is used this latter method can turn the current off within the
102
Power switching devices
turn off time of the GTO, i.e. 10 to 50 microseconds from the point where
the overcurrent is detected and the gate drive circuits initiated to turn the GTO's
off.
Fig. 2.33 This is a complete phase assembly for use in a PWM inverter. It uses gate turn off
thyristors and all the associated components required are mounted on this assembly.
The di/dt reactors are at the bottom and the gate drive circuits are at the top. Air is
blown through the central duct to cool the components. (G.E.C. Industrial Controls,
Ltd.)
Parallel and series operation of GTO's
Multiple GTO switches can be used to produce higher power switches but when
doing so it is necessary to carefully match the GTO's and circuits if correct
sharing of the current and voltage is to take place.
For parallel operation it is necessary to select GTO's for:
Forward voltage drop
Turn on delay time, and
Turn off storage time
as well as to add parallel sharing reactors in series with the GTO's and make sure
that gate drive and snubber circuits are the same.
Power switching devices
103
For series operation the GTO's will need to be matched for:
Forward leakage current and
Turn off storage time
and the snubber circuits will also be expected to perform the job of ensuring
transient and steady state sharing of the total circuit voltage between the
individual GTO's.
Chapter 3
Power switching circuits for
variable speed drives
3.1 Introduction
We have already studied the motors used and the types of semiconductor
switches. In this, the last preliminary chapter before getting into the drive
systems themselves we will be considering the circuits in which the switches
already discussed will be used.
The three phase bridge or double way circuit is now almost universally used
in variable speed drive systems but its operation varies with the type of switches (
being used and overall characteristics of the remainder of the system. In its
naturally commutated form it can operate in its rectifying or regenerative mode
depending on whether the power flow is from the AC to the DC or vice versa.
The bridge circuit is also used for motor convertors to direct the DC link power
to the correct motor windings. In this case its operation depends on whether the
circuit in total has a high or a low impedance, i.e., whether it is a current source
or voltage source system.
Other circuits are used in variable drive systems. For example, some small
DC link inverters can be operated from a single phase mains supply and in
such cases a single phase mains side convertor will be used. As will be seen in
Chapter 9 some cycloconvertors can use three pulse convertors as an alternative
to the six pulse, 3 phase bridge. However, these arrangements are relatively
unusual in practice and so in this chapter we will be concentrating on the circuit
most widely used in all systems, the three phase bridge.
3.2 The 3 phase naturally commutated bridge convertor circuit
3.2.1 As a rectifier
When six switches are connected together as shown in Fig. 3.1 it is possible for
them to convert the three phase AC fixed voltage mains supply into variable
voltage DC power. This is done by closing the positive switches in sequence
when the mains sinewaves become positive and closing the three negative
switches when the mains sinewaves are negative. If the switches are closed at the
Power switching circuits for variable speed drives
105
correct points in the cycle it is possible for the current to naturally pass from one
switch to the next under the influence of the reversing AC voltage sinewaves.
To achieve this the three positive switches have to be closed at intervals of
120 electrical degrees referred to the supply frequency and the three negative
switches closed at 120 degree intervals to each other but 60 degrees displaced
from the positive ones.
DC positive
ATT3
STI
AT4
AT6
ti
t3
t2
I
A
\
Fig. 3.1 The 3 phase bridge convertor
I
I
lcurrent
Mlow
AT5
DC negative
t4
t5
I
I
t6
I
/ C
\
\
/
\
.••'•
\
/
V \ /
\c
V
\"A
.. B
Fig. 3.2 DC positive voltage
If the three mains sinewaves are as shown in Fig. 3.2, these being the supply
phase to neutral voltages and the three positive switches are closed at the points
shown then the three sections of the AC sinewaves will be transferred to the DC
106
Power switching circuits for variable speed drives
positive output terminal and this would take up the voltage shown by the heavy
line.
If we assume that a steady DC current was flowing at this time then this
current would flow through Tl during the time tx to t3, through T3 during the
time t3 to t5 and so on. During the period tj to t3 the A phase voltage is the most
positive and even if all three switches Tl, T3 and T5 had been closed the current
would only flow through Tl because T3 and T5 would have a reverse voltage
, cx= 30°
Fig. 3.3 Varying the firing points
across them. When T3 is closed at t3, its voltage (phase B) will now become the
most positive of the three and the current will automatically transfer into the T3
switch, with Tl ceasing to carry current. In other words as long as the switches
Power switching circuits for variable speed drives
107
are closed at the correct times the current will naturally commutate into the
correct switches. These switches can therefore be normal thyristors which are
capable of being switched ON but will only revert to their blocking state if the
anode current is brought to zero.
If now instead offiringthe thyristors at the above points, we delay their firing,
then we will transfer a different 120 degree section of the mains supply voltages
to the DC positive connection. As long as the delay is no more than 180 degrees
then natural commutation will still take place. This can be proved by considering the switch over of current from Tl to T3, if T3 is fired at t3 then as the B
voltage exceeds the A voltage T3 will take up the current. If thefiringof T3 was
delayed the B voltage will still be higher than the A voltage and so T3 will still
take up the current; this situation still occurs as long as the firing of T3 is not
delayed past t6, i.e. 180 degrees from t3.
Fig. 3.3 shows the results of delaying thefiringof all positive thyristors. From
this you will see that the average voltage occurring on the DC positive output
connection with respect to the supply neutral will reduce as the firing point is
delayed. You will also see that the average voltage will become negative if the
delay is greater than 90 electrical degrees. In fact this is only true if something
else in the circuit causes or allows the current to continue flowing in the DC
circuit. For the present let us assume the current isflowingin an inductive load
as Fig. 3.1.
The operation as explained above for the positive side of the bridge can also
happen in reverse on the negative side, but now thyristors T2, T4 and T6 must
be fired at or later than points t2, t4 and t6 respectively.
In practice both sides of the bridge are normally operated together with all
6 thyristorfiringpoints being delayed by the same amounts at all times and with
the six thyristors beingfiredat 60 degree intervals in the sequence designated by
their numbers. The total voltage occurring across the DC terminals will then be
equal to the (positive side to neutral voltage) minus (the negative side to neutral
voltage) and this is shown for delay angles of up to 90 degrees in Fig. 3.4. The
total voltage has a six pulse ripple component and the DC current is always
flowing in one positive thyristor and one negative thyristor at the same time.
Commutation
In practice the current does not immediately transfer from one thyristor to the
next, due to inductance in the mains supply system the two thyristors both carry
current for a short period while it is transferring from one to the next. During
this period the DC voltage takes up a mean value between the two appropriate
sine waves, as shown in Fig. 3.5, this voltage drop occurring across the reactance
of the supply. The overlap angle will vary with the value of the current flowing,
the inductance of the supply and the value of the delay angle. The result of this
overlap period is that the mean value of the DC voltage is reduced. Further
study of the operation of this circuit can show that the mean DC voltage
108
Power switching circuits for variable speed drives
occurring across the load is given by the equation:
Vd =
x Vac x (cos a - Xt/2) - IdR - 2VT
where Vac equals the RMS AC supply line voltage
a is the delay angle
Xt is the per unit supply reactance
Id is the DC mean current
\|/of=9O°/' I
f \ /^.
a v e r d g e v a l u e ' - - z e r o •'•.
\ \ /
VV • y \ y/ \y/ \y. V\ y//
\
/\
/\ / \ / \
Fig. 3.4 Varying DC voltages
Power switching circuits for variable speed drives
109
R is the circuit resistance (excluding the load)
VT is the forward voltage drop of a thyristor switch.
This equation assumes that the DC current is continuous, i.e. it does not come
to zero at any time. Fortunately most loads are inductive and discontinuous
current usually only occurs at low values of load current.
Fig. 3.5 Natural commutation
This equation can alternatively be expressed as:
Vd =
3V2
x Vac x cos a — 6 x Id x f x Ls - IdR — 2VT
110
Power switching circuits for variab/e speed drives
where f is the supply frequency and Ls is the total effective inductance of the AC
line connections and supply system. The first term in this equation is the DC
voltage neglecting overlap and switch voltage drops. The second term is the
voltage drop due to supply system reactance and caused by overlapping as one
Fig. 3.6 This picture shows a complete 3 phase naturally commutated thyristor bridge containing both forward and reverse thyristors for providing full four quadrant motoring
and regenerative operation when fed from a 3 phase mains supply at up to 500 volts
AC line. (G.E.C. Industrial Controls, Ltd.)
Power switching
circuits for variable speed drives
111
current drops and the other rises. IdR is the resistance voltage drop and the final
term, 2 x VT, is the voltage drop in the positive plus the negative switches
through which the current is flowing.
3.2.2 As an inverter ~ regeneration
You will have noticed from Fig. 3.4 that with a delay angle of 90 degrees the
voltage waveform on the DC side oscillates above and below zero at six times
the mains frequency. If the circuit is such that the DC current is continuous so
that each thyristor carries current (and therefore passes the AC voltage to the
DC) for the full 120 degrees then the mean value of this DC voltage will be zero.
If this continuous current flow can still be maintained it is possible for delay
angles greater than 90 degrees to lead to a negative average voltage across the
DC terminals. We now have a positive DC current and a negative DC voltage
and the result is that the power has reversed and it is nowflowingfrom the DC
load to the AC supply. This is the regenerative condition of this circuit when the
bridge is inverting the load DC power into AC.
current
load
voltage
Fig. 3.7 Regeneration
In order that continuous DC current will flow it has to be the load which is
forcing its flow around the circuit and Fig. 3.7 shows this condition. The mean
voltage from the load has to be slightly higher than the bridge voltage in order
that the current is forced around the circuit overcoming the voltage drops in the
circuit resistance and inductance. The inductance LL is shown in the circuit and
this makes sure that the voltage ripple coming from the bridge does not allow
the current to drop to zero at any time.
The circuit is shown with a DC machine as the load, and in this condition it
would be operating as a generator. However, in AC drive systems the load is
likely to be other convertor/inverter bridges feeding AC motors, and there may
be large capacitors across the DC link or relatively large reactors in series.
Whatever the circuit consists of the load will be equivalent to the DC generator
shown in Fig. 3.7 as far as its effect on regeneration of the supply bridge is
concerned. It may not produce as smooth a back emf as the DC generator but
the load circuit will be serving to keep the flow of current continuous.
On this basis the control angle of the mains convertor can be increased up to
/12
Power switching circuits for variable speed drives
at least 150 degrees to achieve high negative DC voltages as required. The limit
to the delay angle is caused by the overlap period as previously explained, the
transfer of tfie current between a pair of thyristors has to be completed and the
thyristor has to have retained its blocking ability before the a = 180 degree
point is reached. If commutation is not completed by this point the current will
Fig. 3.8 DC voltages during regeneration
not transfer but will revert back to the previous thyristor causing a very high
fault current to subsequentlyflowin the system due to the bridge output voltage
which will reverse to the positive side upsetting the balance of voltages in the
load circuit.
Power switching circuits for variable speed drives
113
Fig. 3.8 shows the DC output voltages produced under regenerative conditions
and it should be noted that:
a) The angle of overlap increases as the delay angle increases; in fact, the
shaded area of voltage drop due to overlap remains approximately
constant at all delay angles and hence the angle alters with the voltage
difference occurring at the firing point.
b) The other point is that this voltage drop now causes the magnitude
of the negative DC voltage to increase, whereas, in the rectifying
condition, it reduces it.
3.2.3 Switch voltages
The thyristor switches used in the naturally commutated 3 phase bridge circuit
need to be able to accept the full peak of the supply line voltage in either the
forward or reverse directions. Under rectifying conditions the voltages across
these are predominately reverse voltages and an increase in delay angle causes
the forward voltage to increase and this dominates in the regenerative mode.
In addition, it is necessary for the thyristor to be able to cope with any
excessive steady state or transient conditions which may occur on the mains
supply. Lightning strikes and the switching on and off of other large loads on
to the power supply system can lead to significant changes in the supply voltage
and very high transient peaks are possible. To cope with lightning most power
apparatus has to be tested at very high levels for tens of microseconds and most
transmission systems have to be fitted with lightning surge arrestors to keep the
voltage transients to acceptable levels.
Semiconductor switches cannot accept excessive voltages even for very short
periods of time without failure and it is not usually practical to rate them to
accept the maximum supply voltage which could occur. It is more economic to
add special surge voltage suppression components to the input AC terminals
and then to use a modest safety margin on the thyristors by choosing their
forward and reverse repetitive voltage capability to be approximately twice the
peak of the nominal AC mains line voltage sine wave. The choice of safety
margin clearly depends on the effectiveness of suppression components
provided.
Fig. 3.9 shows the theoretical voltage to occur across the switches under
different delay angle and load conditions, showing that it is the peak of the line
voltage which is important. This also shows the importance of the commutation
overlap periods which considerably distort the waveform. The 'notches' in these
waveforms occur due to the sudden switching ON of other thyristors in the
circuit and the rates of rise and fall of the voltages is directly dependent on the
speed of their switching and the effectiveness of circuit inductances and snubber
components. It is the rate of application of forward voltage (dv/dt) which is
most important and the 120 degree delay angle condition is the most serious
from the thyristor point of view because the high dv/dt occurs just as the voltage
crosses the zero value. At this point the voltage rises to approximately 50% of
114
Power switching circuits for variable speed drives
the peak value, very suddenly, and it is this condition which often decides the
degree of dv/dt protection included in the circuit.
The practical waveforms differ slightly from these theoretical ones due to the
presence of the other components in the circuit, surge suppression capacitors
Fig. 3.9 Thyristor anode/cathode voltages
and snubber circuits. These usually introduce oscillatory resonant 'ringing' into
the voltage waveforms and Fig. 3.10 shows the typical results occurring across
the thyristors in this circuit. Clearly these oscillatory peaks also have to be taken
into account in choosing the voltage and dv/dt capabilities of the thyristors.
Power switching circuits for variable speed drives
115
delay angle = 35°
Fig. 3.10 Oscillogram of voltage across thyristor
3.2.4 DC voltage harmonics
Clearly the DC voltages produced by this circuit are far from smooth and
steady, a considerable content of six pulse harmonics occurs and this will have
an important influence on the flow of harmonic currents in the circuit. It is easier
to appreciate the effects of these harmonics if the waveform is analysed and split
into its component characteristic harmonics. A Fourier analysis shows that the
output voltage waveforms of this circuit contain 6th, 12th, 18th, 24th, etc.
harmonics in reducing magnitude, approximately the maximum magnitude of
the harmonic is inversely proportional to its harmonic number, e.g. the twelfth
harmonic is approximately half the magnitude of the 6th, etc. Fig. 3.11 shows
the RMS value of the DC harmonics at different delay angles showing that, as
would be expected, the worst case occurs at zero mean voltage, i.e. 90 degrees
delay angle. At this point the voltage distortion level is approaching 30 per cent
total harmonic distortion. It improves at high levels of voltage whether in the
rectifying or inverting regions.
The curves in this figure have been drawn assuming negligible overlap. In
practice, the presence of overlap does alter the higher harmonics significantly
but the sixth is relatively unaffected. In practice it is usually the lower harmonics
which are most important and it is unnecessary to delve further into this subject
here.
If the harmonic impedance of the DC circuit can be estimated it is possible
to calculate the approximate values of the DC current ripples from these voltage
figures.
3.2.5 AC current harmonics
The situation regarding the current in this circuit can best be appreciated by the
study of diagrams of the type shown in Fig. 3.12. This shows the total current
flowing in the DC positive output connection and that flowing in the DC
negative connection, shown on either side of a zero line. Obviously the same
current flows in both these connections so that these currents are equal and
116
Power switching circuits for variable speed drives
25 -
20
40
60
80
100
120
delay angle oc
140
160
180
Fig. 3.11 DC output voltage harmonics for the 3 phase, fully controlled bridge
opposite as shown. The thyristors chop these DC currents up so that it is time
shared between the thyristors feeding the appropriate connection, i.e. referring
to Fig. 3.1, the positive side current is shared between thyristors Tl, T3 and T5
and the negative current between T2, T4 and T6. This is shown in Fig. 3.12
where each thyristor carries the current for 120 degrees plus the overlap angle.
I have shown some ripple on the DC current so that the diagram is a more
realistic representation of practical circumstances.
The current which flows in the AC lines is a combination of the appropriate
positive thyristor current and the negative thyristor current and thatflowingin
the A phase connection is shown in heavy lines. In the ideal case with infinite
DC inductance and negligible overlap this becomes a quasi-square wave with
two 120 degree current blocks, one positive and one negative per cycle. In this
ideal form the AC current waveform will contain a harmonic spectrum as shown
in Fig. 3.13 with the magnitudes of the harmonics reducing as frequency
increases. You will notice that the magnitude of the harmonics is directly related
to the level of fundamental (or mean) currentflowingso that the harmonics will
be high when the fundamental current is high and vice versa.
Power switching circuits for variable speed drives
117
DC positive current
rVr\ A A
T5
T3
,'i
B*
C*
B-
B-
T6
T2
I V/ \
J
DC negative
current
I
T6
/j
M /
\
A1
/
A phase
current
Fig. 3.12 Current flow in the 3 phase bridge
20r
variation due to
circuit reactance
g I 10
I
5th
7th
11th
13th
harmonic number
Fig. 3.13 Harmonic
spectrum
of the AC
current
This ideal case rarely applies in practice due to overlap and the presence of
ripple on the DC current. However, the ideal approach is relevant to many cases
because the highest harmonic currents usually occur at the highest fundamental
current and under high load current conditions the DC current ripple is at a
minimum.
The shaded area on the tops of the columns of Fig. 3.13 indicate a typical
118
Power switching circuits for variable speed drives
variation in harmonics due to overlap and delay angle changes. The highest
values correspond to low current or low DC voltage conditions and the lowest
values to high current and high DC voltage conditions. The variation due to
these changes is small.
The changes due to the variation in DC ripple current can be significantly
larger. Increased DC current ripple does not necessarily increase the total level
of harmonics however, but it does tend to increase some, while at the same time
reducing others. It is not easy to generalise because of the many possible DC
T -A/WW
Ai
Idc (mean)
10
DC ripple factor
- Ai
Idc (mean)
Fig. 3.14 The effect of ripple on the DC current on the AC input harmonics
current waveshapes depending on DC load resistance, inductance and back emf
combinations but the chart on Fig. 3.14 gives approximate values of these effects
under the conditions likely in AC variable speed drive systems. This shows that
ripple in the DC current causes the fifth harmonic in the AC supply current to
Power switching circuits for variable speed drives
119
increase whereas the other harmonics reduce from the nominal smooth DC
current value. The amount of ripple in the DC current and hence the value of
the DC ripple factor will depend on the firing angle of the bridge and the load
impedances, etc; the DC ripple factor of two corresponds approximately to the
onset of discontinuous current.
The DC current harmonics in AC variable speed drive systems may also
include harmonics related to the motor frequency and in general they will not
be directly synchronised with the mains frequency. The result is a continuous
changing of the AC current waveshapes and the introduction of frequencies
completely unrelated to mains frequency and varying With the motor frequency
and speed. Fortunately the levels of these harmonics are not usually large or
important.
3.3 The 3 phase bridge inverter
Many of the DC link inverter systems require motor convertors which can
convert the DC link power into variable frequency AC to the motor and the
most usual circuit used for this is again the 3 phase bridge. In this case however,
the bridge needs to be self-commutated because it is not possible to rely on
induced voltages coming back from the motor. It is usually shown as in Fig. 3.15
as the power normally passes from the DC side to the AC motor and in this each
of the switches has to be capable of being turned ON or OFF itself.
Idc
DC
Fig. 3.15 The 3 phase inverter bridge
The switches can be any of the self-commutated types discussed in Chapter
2, i.e.
Normal thyristors with forced commutation capacitors, reactors and
switches.
Transistors
Gate turn off thyristors
120
Power switching circuits for variable speed drives
and the choice will normally be made based on the rating and performance
required from the circuit.
The principle of operation of this circuit is that the switches are opened and
closed in such a way that the DC voltage or current appears on the AC output
as alternate positive and negative polarity. There are in fact two ways that this
circuit can be used and the circuit operates differently in the two cases. Which
method is appropriate depends on the circuits peripheral to the bridge. If the DC
source is a low impedance voltage which is capable of allowing any load current
to flow, then the closing of the switches will cause the voltage on the DC
connections to be transferred to the output AC terminals of the motor. If the
impedance of the DC supply is relatively high so that the DC current is
smoothed and unable to change rapidly then the switches are used to direct
this current into the appropriate phase windings of the motor; the voltages on
the DC terminals and the motor terminals will then depend principally on the
motor induced voltage rather than on the original DC link voltage. The low
impedance DC source operation is known as a voltage source system and the
high impedance case is known as a current source system. These two will be dealt
with separately.
3.3.1 The voltage source bridge inverter
In this case the closure of a switch will transfer the DC voltage to the appropriate AC output terminal and the value of the DC voltage will be relatively
unaffected by the flow of current which results. The waveshape of the output
voltage depends on the time of operation of the switches as shown in Fig. 3.16
which shows the voltage occurring on output terminal A due to the operation
of switches SI and S4. If each switch is closed for half of the full cycle as in
Fig. 3.16(a) a square wave output results. Fig. 3.16(b) and (c) show that the
magnitude and waveshape of the output can be changed by reducing the periods
when the switches are closed. If the switches can be opened and closed at will
then arrangements like Fig. 3.16(d) can be readily produced.
The problem shown up by this figure is what happens to the current; in this
simple circuit the current can only flow through a switch when it is closed so
arrangements have to be made for the load current to be allowed to flow elsewhere during the periods while the switches are off. It is not possible to switch
the load current ON and OFF directly with the switches due to the inductive
nature of the load and the circuit voltages which would result from trying to
switch the current off.
All voltage source circuits therefore include alternative paths for the flow of
the load current and Fig. 3.17 shows the normal bridge arrangements with
reverse connected diodes across each switch. Now, when positive current is
flowing into the motor, say through the A phase, and SI is closed the current
willflowthrough it, when SI is open the current willflowthrough D4. Negative
current flowing from the motor will either flow in S4 or Dl. Clearly when the
current is flowing in the diodes it will be reverse current into the DC link and
Power switching circuits for variable speed drives
121
S4 closed
SI
closed
SI
closed
Si open
SA open
„
SV
close
SA open
.
SA closed
\
\
\
SI closed
Fig. 3.16 Output voltage waveforms possible with a voltage source inverter bridge
di
D3
D5
1 SI
-i-DI
S5
V
dc
—- reactors
dt
S3
B AC
output
*
r
S2
,DA
Fig. 3.17 The voltage source inverter circuit
122
Power switching circuits for variable speed drives
the DC link has to be able to accept the current. Fig. 3.18 shows the general case
where the switches are ON for part of the cycle and due to the influence of load
motor inductance the output current is relatively sinusoidal and continuous at
a power factor of less than unity. Now whenever SI is closed and positive
current is flowing, it will flow through SI; otherwise the remainder of the
positive current will flow through D4, this is shown shaded.
SI
closed
current in D1
power factor
angle
c
S4/
L\.
closed '
Fig. 3.18 Current flow in the voltage source inverter
Similarly with the negative current, the dotted portions willflowin the diode
Dl. The result is that the motor current is chopped up into sections by the
operation of the switches. The overall result is that the DC positive connection
current is a complex sum of the currents flowing in the switches SI, S3 and S5
and the diodes Dl, D3 and D5. Another important conclusion is that the current
has to switch from one switch into a diode and vice versa very quickly. It is
normal for a large capacitor to be connected across the DC link to allow the
reverse diode currents toflow.It is also normal for the mechanical construction
to be relatively compact to minimise the stray inductances which would hamper
the fast alterations in the current flow paths.
Study of Fig. 3.18 will also show that the diode currents have two components,
the current when the main switch is opened and secondly the current still flowing
at the end of the appropriate half cycle of voltage pulses. This latter component
is caused by the power factor of the load current, when the load power factor
is high the diode current is low. When the load power factor is low most of the
current flows in the diodes.
The presence of the diodes means that no reverse voltage can ever occur
across the switches and this makes it possible for transistors and GTO's to be
employed in this circuit.
Power switching circuits for variable speed drives
123
The necessity for fast diodes
Reference to Fig. 3.18 shows that the current has to switch from diode D4 into
SI at points marked X. At the point where SI is switched on it immediately
applies reverse volts to D4 and the current in D4 drops quickly, limited only by
any di/dt limiting reactors in the circuit. Due to the stored charge effects in the
diode the current in it will temporarily reverse before the diode recovers and as
the full DC link voltage is behind the flow of current it can rise to quite a high
value particularly if the diode has a high stored charge value. Once the free
carriers have been removed from the diode junction the current ceases to flow
and it can do this very quickly causing very high voltages in the circuit. This is
shown in Fig. 3.19 which shows that the reverse diode current causes the switch
current to overshoot and this has to be taken into account in rating the switch.
over shoot in
switch current
only limited
by circuit
components
slow
diode
diode
stored charge
area
Fig. 3.19 Diode recovery
From this point of view the preferred diode, chosen to minimise adverse effects
on the main switches, is one with a low value of stored charge and with a slow
recovery of blocking capability. Such a diode will have a recovery reverse
current of the type shown in Fig. 3.19, the reverse current being restricted to a
low value (to limit the switch current overshoot) and with a slow tail off of
reverse current to prevent high induced voltages in circuit and stray inductance.
These are usually referred to as fast turn off diodes and they are essential when
high speed switching is required.
124
Power switching circuits for variable speed drives
Regeneration
The condition of powerflowfrom the motor to the DC link in this system means
that the majority of current flow now goes through the diodes and is therefore
fed back as negative current into the DC link. The result is that the DC link
capacitor gets charged from the load energy and if this condition is to be
prolonged it is necessary to remove the energy from the DC link to prevent the
capacitor voltage from rising out of control and damaging the circuit switches.
3.3.2 The current source bridge inverter
During the discussion on the naturally commutated rectifier bridge the assumption was regularly made that the DC current would be smooth and continuous
due to the presence of inductance in the DC circuit. When the load for such a
convertor bridge is an inverter bridge the DC link reactor has the same effect on
the motor inverter bridge by preventing the circuit current from changing
rapidly. The operation of the inverter is then quite different to the voltage source
situation in that the switches now perform the job of directing the steady and
continuous DC current into the appropriate motor windings. It is therefore the
circuit current which now dominates the operation of the bridge rather than the
voltage. It is now therefore essential that one of the three positive switches and
one of the three negative switches in the bridge are always closed to provide a
path for the current. On this basis the current will always have a path to flow
in and there is normally no need for additional components to provide alternative paths. In addition, because the voltage on the convertor side of the DC
link reactor does not have to be identical to that on the input side, it is possible
to accommodate for any motor power factor and any angular displacement
between motor current and voltage.
Idc
Vdc 2
Vdci
Idc
Fig. 3.20 The current source inverter circuit
Fig. 3.20 shows the current source circuit and Fig. 3.21 shows the way the
current is split up by the convertor into three AC output currents to the motor.
The currents will be basically quasi-square wave in shape; this being dictated
Power switching circuits for variable speed drives
125
solely by the DC link reactor and the switches, the motor voltage has hardly any
influence on the shape of the current waveform.
The voltages occurring in this circuit are dependent on the induced voltages
in the motor and the load on the system. It is only possible to be certain of one
thing, i.e. when a switch is closed then the DC connection is directly connected
• 1 cycle of motor frequencyDC t current
55
5A
switch 1
closed
switch 6
closed
switch 3
closed
switch 2
closed
switch 5
closed
switch A
closed
SI
zero.
S6
DC - current
Fig. 3.21 Current flow in the current source inverter circuit
to the appropriate AC motor terminal. The voltage Vdc2 in Fig. 3.20 is therefore
the rectified value of the motor terminal voltages and its value depends on the
phase relationship between the closure of the switches and the induced voltages
in the motor windings. What happens is that the value of Vdc2 varies with the
load and the motor power factor reduces with the load so that the multiplication
of the DC link voltage and current equals the power being drawn by the load.
126
Power switching circuits for variable speed drives
Regeneration
The condition of zero power being fed to the motor is usually represented by
zero DC volts and a significant value of DC current. The reversal of the power
flow, i.e. power from the motor to the DC is represented by a reversal of the DC
link volts, the current continuing to flow in the same direction as when the
motor was being driven.
In order for the system to balance in the steady state condition the input mean
DC voltage Vdcl will always be equal to the mean value of Vdc2.
In this circuit there is nothing to prevent reverse voltages occurring across the
switches and this condition will occur regularly under most loading circumstances.
The switches therefore have to be capable of accepting similar levels of forward
and reverse voltage.
As will be seen in the later chapters on current source systems the switching
over of the current from one switch to the next is accompanied by high transient
voltages being generated in the motor leakage inductance.
Although it is theoretically possible to consider more complicated patterns of
firing of the switches this is rarely done in practical designs because of the
additional switching transients likely to be introduced. It is also theoretically
possible to allow the current to flow straight through the inverter from the DC
positive to negative, by passing the motor if this is beneficial to the system
operation.
3.4 Isolation of electronics
In all the circuits discussed in this chapter the power semiconductor switches will
be operating at the mains or motor voltages and it is found that the many
switches in the circuits all operate at different potentials from each other at any
one instant. In most circumstances it is required to control these switches by
means of relatively complex electronic systems which will be being fed by
regulated power supplies in the 5 to 20 volt region and with one rail of the power
supplies at or very close to earth potential. It is therefore necessary to isolate the
electronics from the semiconductor switches, sometimes to quite high voltage
levels.
At the same time the semiconductor switches require a significant amount of
power to be available to feed the gate or base control terminals. It is not
normally practical to obtain this power from the anode and cathode circuit
connections of the switches; it is preferable to use auxiliary power supplies for
this purpose. If this power is to be obtained from a common source, e.g. an
auxiliary mains transformer or from the DC link, it will be necessary to isolate
the feeds of power to the individual switches because they are operating at
different potentials and the gate/base connections will also be at the potentials
of the switch anodes and cathodes.
Power switching circuits for variable speed drives
127
If the switches are naturally commutated thyristors as in most mains rectifier
bridges, the solution to these isolation requirements can be relatively simple.
Because the gate power required by normal thyristors is low, and because the
thyristors can be turned on with single pulses or trains of pulses having a low
on to off ratio, it is possible to pass this power to the gate using small voltage
or current type pulse transformers. It is possible to make such transformers with
large voltage capabilities between the secondary and primary and of sufficient
quality to ensure the passage of good pulses to the gate. It is also possible to
control the flow of pulses to the gate from the primary low voltage side of the
control signals
from earthed
electronics
power supplies
to pulse amplifiers
pulse
amplifers
isolation pulse
transformers to isolate
gate power and
controlling signals
non - active
pulse shaping
and interference
protection
components
DC
voltages to the load
AC
mains
voltages -
to isolation circuits as above
Fig. 3.22 Isolation of naturally commutated
bridges
pulse transformer. Fig. 3.22 shows such a typical arrangement applicable to a
naturally commutated thyristor rectifier. In this case the pulse transformers will
only be a few watts rating and all the switching will be carried out on the low
voltage primary side of the pulse amplifiers. The circuits shown on the secondary
side of the pulse transformers will consist only of non active components,
e.g. resistors/capacitors/diodes, included to control pulse shape or to reduce
interference effects.
A similar approach can be adopted for forced commutated thyristor circuits
because only low level turn on pulses are required by both main and auxiliary
thyristors.
The requirements for transistor and gate turn off thyristor circuits, however,
have to be more complex from this point of view because of the much higher
power required to control them. With the relatively low gains of power transistors,
even the Darlington cascade types, relatively large control currents have to be
used. It is also necessary to have positive and negative sources available to
achieve optimum transistor switching. It is a similar story with GTO's, although
high voltage
isolation
barrier
Fig. 3.23 Typical isolation of a transistor inverter
DC-
DC*
transformers for
power feed to base
drives
electronic controls
output
opto-isolators
for signal feeds
to base drives
B
i
CO
1
I
I
5
00
Power switching circuits for variable speed drives
129
the initial turn on power is not particularly large the fact that continuous gate
current is required during the ON period increases the ON power very significantly. The switch off gate current is much larger still, often being up to 20 per
cent of the anode current and negative bias voltages are also required to increase
blocking capability. The result is much more control power required and more
difficulties with isolation.
The base and gate drives required for transistors and GTO's often require
positive and negative power supplies and this means that a separate regulated
power supply is required for each of the switches in the circuit. Two methods
of feeding and isolating these power regulators are regularly used in variable
speed AC drives. They are:
1) To have a mains fed transformer with an individual secondary for
each of the base or gate drive circuits.
2) To have a high frequency chopper type switched mode power supply
fed off the DC link with an output transformer or transformers with
individual isolated secondaries for each switch.
mains power
supply transformer
with isolated
secondaries
L-
nr
barrier
\
UJJ__UU__UJ
nn
electronic controls
tor signal feed
nn
Fig. 3.24 Typical isolation of a GTO inverter
In this latter case it is possible to operate with a short break in the mains
supply because in many cases the DC link capacitor can hold up the DC link
for a reasonable period of time.
In order to isolate the electronics it is necessary to feed the control signals
to the base or gate drive circuits either via isolating transformers or via optoisolators and both methods are in regular use.
130
Power switching circuits for variable speed drives
Figs. 3.23 and 3.24 show typical arrangements. The transistor inverter circuit
is shown with a switch mode DC fed power supply to feed the power to the base
drives and opto-isolated control signals from the electronics. The typical GTO
circuit includes a mains fed transformer to feed the auxiliary power and small
pulse transformers to isolate the control signals.
In both cases the high voltage isolating barrier in the transformers and
opto-couplers have to be rated for the full peak voltage of the power circuit
including any transient allowances. Testing normally has to be carried out at AC
sinusoidal levels of twice the normal circuit voltage plus 1000 volts.
Chapter 4
The six step voltage source inverter
for induction motors
4.1 Introduction
This system was one of the earliest DC link induction motor drive systems to
be developed and it came into use soon after the principles of forced commutation
of thyristors became established in the 1960's.
Like many of the systems to follow it consists of a convertor to change a fixed
frequency, fixed voltage mains supply into variable voltage DC, followed by a
forced commutated inverter to convert the DC to a variable frequency AC
output. In this case, the output is a quasi-square voltage waveform which is very
well suited to supply the most reliable and robust of motors, the induction
motor.
Although there are now very few new designs of this type being generated,
there are a large number of these drives in service and an understanding of this
drive is essential as a basis for those to follow.
In general this type of drive has been manufactured in sizes from less than one
kilowatt up to hundreds of KW and at motor voltage levels up to 500 volts.
Because of its simplicity of switching it is suitable for relatively high frequencies
of operation and it is in the high frequency area that this drive has a long term
future. It has been used extensively for drives operating at mains frequencies
of 50 to 60 hertz but such drives are now using the pulse width modulated
techniques described in Chapter 5 where improved performance can be
achieved.
4.2 Principles of operation
The elements of this system can best be explained with reference to Fig. 4.2
which shows the naturally commutated mains supply convertor which rectifies
the mains to produce variable voltage DC into the DC link. The dominant
feature of the DC link is a large capacitor which is capable of ensuring that the
DC link voltage can only change relatively slowly and which is able to provide
whatever current is demanded by the following inverter. The DC link may also
132
The six step voltage source inverter for induction motors
contain a small reactor to limit fault currents and to help to isolate the two
convertors from each other.
The inverter bridge consists of six switches each of which is capable of
switching the currents on and off itself. They may consist of power transistors,
forced commutated thyristors complete with their commutation circuits or gate
turn off thyristors with their gate drive systems. These six switches connect the
Fig. 4.1 This cubicle contains the complete six step voltage source drive for a 200 HP
induction motor. The thyristor rectifier and DC link capacitors are shown in the lower
right section and the forced commutated thyristor inverter is mounted above the
central coo/ing fan. Auxiliary power circuit components are in the left hand section
and the electronic board is mounted on the left hand door. (G.E. C. Industrial Controls,
Ltd.)
The six step voltage source inverter for induction motors
133
DC link voltage onto the motor terminals in sequence to produce a square wave
motor terminal voltage of any frequency from zero to the switching limits of the
inverter.
The supply side convenor controls the level of voltage occurring on the DC
link and therefore the value of the voltage applied to the motor. It is the voltage
which dominates this system and the currents which flow will take up the levels
and waveforms dictated by the circuit voltages and impedances.
DC link
reactor
supply side
convertor
DC
capacitor
reverse
diodes
motor side
inverter
Fig. 4 . 2 The six step voltage source inverter drive
In general the motor currents will be out of phase with the output voltage due
to the motor magnetising current and it is necessary to provide a path for the
flow of this reactive current. Reverse diodes are included across the inverter
switches for this purpose and they return the reactive currents back into the DC
link capacitor.
The current whichflowsin the supply side convertor is related to the in phase
value of the motor current.
The frequency of the output to the motor is controlled by the rate of switching
of the motor convertor switches and this is usually decided by a variable
frequency oscillator in the electronic circuits controlling the inverter.
The system usually operates without any feedback of the motor rotor conditions e.g. speed, reliance being placed on only electrical stator measurements.
The principle is to apply the appropriate voltage and frequency to the stator of
the motor and then to leave the motor to look after itself. The optimum
situation is usually achieved by controlling the voltage to frequency ratio so as
to maintain a constant flux in the motor air gap.
The normal method of inverter switching is to close the two switches of each
phase alternately so that the connection to the motor winding is alternatively
switched to the positive and negative rails of the DC link. The three phases are
operated sequentially, 120 electrical degrees apart and the result is a quasisquare wave voltage appearing across the motor terminals as shown in Fig. 4.3.
The currentflowingin the motor windings is dictated by this voltage waveform
134
The six step voltage source inverter for induction motors
and by the effective leakage inductance of the motor stator windings. Over most
of the frequency range the current waveform is more sinusoidal than the voltage
and reasonably smooth motor operation is achievable. At low speeds the current
waveform more closely follows the voltage and the stepping of the motor
rotating fields leads to the motor cogging round. Usually operation down to
four or five hertz is practical.
I
switch
A
closed
switch
1
closed
switch
Vdc
closed
S6
S3
S5
S2
B
S3
S2
C
S5
T
Vdc
I
voltage
A-B
line
voltage
I3-C
li ne
voltage
C--A
Fig. 4.3 Motor voltage waveforms
The direction of rotation of the motor can be selected at will electronically just
by changing the sequence of closing the inverter switches, clearly change of
direction is normally only carried out with the motor at rest.
The six step voltage source inverter for induction motors
135
Although the inverter itself is inherently capable of operating in its rectifying
mode to take power from the motor and feed it into the DC link these drives
are not usually regenerative because it is necessary to add additional equipment
in order to feed the regenerated power back to the mains supply. With this drive
the reversal of power is brought about by reversing the DC link current, keeping
the voltage in the same direction as when motoring. This current can only be
returned to the supply if a reverse connected supply side convertor is included
in the drive.
Control is usually arranged so that the motor operates at its designed air gap
flux level and at a low slip value so that the current-demanded by the motor is
at a minimum level. This is because most inverters are limited in the amount of
current which can be switched, so that it is most economical to achieve the
maximum torque from the current available from the inverter. If the convertor
is notfittedwith a means of feeding back or absorbing regenerated energy it may
be necessary for the control system to prevent regeneration occurring at all.
Normally, whenever the inverter frequency is reduced below that dictated by the
motor rotor speed, the motor will immediately become a generator and regenerate
the load inertial energy. When this occurs the current becomes in antiphase to
the voltage and the majority of the current flows via the feedback diodes onto
the DC link where it causes the capacitor to be charged up to a high voltage very
quickly. As this could easily damage the circuit switches this condition has to
be prevented. A method often used to do this is to allow the inverter frequency
to be dictated by the DC link voltage so that if the voltage rises the frequency
is automatically raised to prevent regeneration.
The DC link reactor helps to smooth the DC voltage and also assists in the
circuit protection by limiting the rate of rise of current being fed into a fault
from the supply convertor.
The general characteristics of this system are that the motor voltage is varied
to match the frequency so as to keep the motor flux relatively constant and the
result is that the DC link voltage also varies in proportion to the motor
frequency and speed. In this case the supply current is approximately proportional to the motor torque and the motor's magnetising current circulates from
the DC link capacitor to the motor. With the mains commutated supply side
convertor the input power factor varies with the DC link voltage and hence it
will be approximately proportional to the speed of the motor.
4.3 Detailed analysis of the system
This system is a voltage source design with a large DC capacitor which prevents
fast changes of DC voltage. As such it is the voltage which dominates the system
conditions and from which the other circuit parameters can be derived. The
capacitor and feedback diodes effectively allow the currents in the circuit to be
decided directly by the value of the DC link voltage and the frequency of
operation of the motor convertor.
136
The six step voltage source inverter for induction motors
4.3.1 Circuit waveforms
Voltage waveforms
As will be seen from Chapter 3 the supply side convertor will produce a DC
voltage containing a significant amount of harmonic ripple at six times the
supply frequency. However the link reactor and capacitor acts as a filter to this
ripple and the majority of it appears across the reactor making the voltage
across the capacitor relatively smooth. This smooth voltage is then chopped up
by the motor convertor to produce a quasi-square voltage waveform across the
motor terminals. This motor voltage waveform keeps the same shape at all
frequencies and it is in fact the optimum shape because it does not contain any
even or third harmonics and the lowest harmonic it contains is thefifth.It is also
a satisfactory waveform because the fundamental value of it is quite high,
approximately 95% of the RMS value, and it is this value which produces power
transfer in the motor.
If the motor is Delta connected then the quasi-square waveform of Fig. 4.3
appear across each of the motor windings. If however the motor is Star connected
then the phase winding will see a different voltage. Study of Fig. 4.3 will show
that there are always two of the three motor terminals connected to one DC rail
with the third terminal connected to the other DC rail. The neutral therefore
takes up a point between the DC rails with a two to one voltage split. Further
study will show that the neutral actually oscillates at third harmonic frequency
about the mid-point of the DC link as shown in Fig. 4.4. The phase voltage with
a star connected motor is therefore shown now to have six steps per cycle each
of one third of the DC voltage. This waveform does in fact contain the same
proportion of harmonics as the quasi-square wave voltage.
Motor current waveform
If we assume that the voltage value is correctly chosen in relation to frequency,
a magnetising current will be drawn by the motor and produce a normal level
of air gap flux. This flux will then result in an induced voltage in the motor
windings which will have an almost sinusoidal shape as explained in Chapter 1.
We therefore have a circumstance where the terminal voltage VI as shown on
equivalent circuit Fig. 1.12(c) as a quasi-square wave and the induced voltage
El is sinusoidal. The harmonic content of the terminal voltage must therefore
appear across the leakage inductance and resistance of the stator winding and
the amount of harmonic current to flow will depend on these impedances. This
therefore provides us with a way of establishing the waveshape of the motor
current theoretically and an example is shown on Fig. 4.5 which has been drawn
for the phase current of a delta wound motor. At the top we see that the
quasi-square wave terminal voltage and the sinusoidal induced voltage
waveform. The difference between these is shown as the heavy line and this is
the harmonic voltage which appears across the leakage reactance and resistance.
This is basically a fifth/seventh harmonic voltage and over the working range
The six step voltage source inverter for induction motors
137
of frequencies the value of the leakage reactance to this harmonic will be 10 to
50 times the resistance value and hence the impedance can be assumed to be
inductive. On this assumption Fig. 4.5(b) shows the approximate harmonic
line
voltage
A-B
+Vdc/2
SI
closed
S4
closed
SI
S4
-Vdc/2
voltage of terminal A (with respect to centre point of DC link)
_Vdc
+ Vdc
1- - - —
voltage of motor neutral (w.r.t. centre point of
motor A phase to neutral
DC
link]
voltage
Fig. 4.4 Motor voltage with star connected motor
current which must flow in the motor winding and this will add to the fundamental current which the motor takes for its magnetisation and to generate the
necessary load torque. At this particular frequency therefore (c) and (d) indicate
the approximate motor winding current waveforms under low load and high
load conditions.
In most drives the voltage and frequency applied to the motor are increased
together in order to maintain the flux in the core at an approximately constant
level. Hence as the level of applied harmonic voltage is increased there is also
a corresponding increase in winding reactance so that the magnitude of harmonic
138
The six step voltage source inverter for induction motors
current stays almost constant over the working frequency range. The waveforms
in Fig. 4.5 therefore are applicable to a wide range of frequencies and are typical
of those which could occur in an actual drive.
a
»
terminal and induced
voltages plus difference
r-y
harmonic
voltage
b
c
harmonic current only
winding current at
low load
d
winding current at
high load
Fig. 4.5 Motor winding currents (delta connected motor)
There will, however, be some differences between line and phase currents
and between the current waveforms in star and delta connected motors. Fig. 4.6
shows the winding currents which would occur with a star connected motor and
these waveforms are produced in a similar way to those of Fig. 4.5.
In general therefore the motor current is reasonably sinusoidal at high speeds
and loads. In practice due to the influence of the motor resistance and the
reduced effect of inductance, the waveform does deteriorate at low speeds and
the performance may not be acceptable in the one tofivehertz region.
The six step voltage source inverter for induction motors
139
Reactive current
Now let us consider where this currentflowsin the inverter. In general the motor
power factor will be less than unity due to the required magnetising current (see
Chapter 1) and study of the waveforms in Fig. 4.5 will show that in this
circumstance there will be a period of time when the current continues to flow
in the one direction after the voltage has reversed. This is shown more clearly
a
terminal and induced
voltages, plus difference
i~?"^*ri
;\
A
77
b
A
Vu--
harmonic
voltage
harmonic current only
A
V
c
\"'0~mm
winding current
at low load
V
fundamental
A
winding current
at high load
Fig. 4.6 Motor winding currents (star connected motor)
in Fig. 4.7 which shows the A motor phase from Fig. 4.2. The phase connection
is being alternately switched to the positive and negative rails with switches 1
and 4 being ON in turn. The shaded portions of the current are unable to flow
in the switches which are ON at that time and the diodes Dl and D4 are
provided to allow a path for these currents. The current flowing in Dl will
therefore flow in the reverse direction onto the DC rail; if other phases are
carrying positive current at this time then this diode feedback current will reduce
the amount of current flowing from the DC link to the inverter as a whole.
140
The six step voltage source inverter for induction motors
DC link current
The DC link current on the output side of the capacitor therefore consists of the
sum of the positive currents flowing in the switches while they are ON, minus
the diode currents due to phase delay in the current. If we assume that the
currents to the motor are sinusoidal this is shown in Fig. 4.7 which shows the
DC link current and how it is made up, for a power factor angle of 30 degrees.
The top of the chart shows the current in the positive side of the DC link and
the lower half of the negative side.
DC
link
A phase
connection
to the motor
voltage at A
current in the A
phase connection
to the motor
Fig. 4.7 Reactive current flow
Normally, with a DC link reactor in circuit, the currentflowingfrom the supply
side convertor is the mean value of that shown in Fig. 4.7 with the ripple current
being provided by the capacitor. This ripple current is important to the specification of the capacitor and an analysis of waveforms like that in Fig. 4.8 show
that it contains frequencies related to six times the motor frequency. Fig. 4.9
shows the results of such an analysis showing how these harmonic currents
change with motor power factor. It should be noted that these curves are based
on sinusoidal current to the motor.
Inverter switch waveforms
It is now possible to construct the waveforms showing the conditions of
operation of the switches in the inverter. The conditions during the commutation
The six step voltage source inverter for induction motors
101 j
D6I
SI
S6
|DA|
D3|
D2J
S4
S3
S2
|D1|
D6i
D5
S1
S6
D2JS2
S5
current in DC positive
A:
A
B
mean
current
I i\
/
C
A
current in DC negative
Fig. 4.8 DC link current waveforms
total RMS harmonic
content
40-
\O\li
o
if)
- 30
Z3
cr U
<b
£ —.
:enta
unda
'o 2 20
c
o> E
10
10
09
08
07
06
0-5
0-A
03
motor power factor
Fig. 4.9 Harmonics in the DC link current
0 2
01
0
141
142
The six step voltage source inverter for induction motors
are not included here as these depend on the type of switch being employed and
reference should be made to Chapters 2 and 3 for these effects.
Each of the inverter switches has a reverse diode connected across it and hence
the voltage across the switch can never be more than the diode voltage drop in
the negative direction. When the switch is ON there will be zero voltage across
the switch. When the switch is OFF its complementary switch will be ON so that
the full DC link voltage occurs across the switch during its OFF period. The
voltage and current waveforms typically occurring on the switches in this drive
system are shown in Fig. 4.10. The current waveform changes with the power
factor of the inverter output current. Under high load torque conditions a
substantial portion of the half wave of motor current flows in the inverter
switch. When the motor is unloaded and only magnetising current isflowing,a
half of the half wave will flow in the switch, the other half flowing in the
corresponding diode as shown in Fig. 4.10(e).
a
switch voltage
on
switch
voltage
off
on
1 switch
current
/
off
diode voltage
dr
°P
/ ~ \
/
J
b
\
/
/
/ \
1
I
diode current
\
currents under normal load conditions
switch
current
r\
c
diode current
currents under low load conditions
Fig. 4.10 Switch current and voltage waveforms
4.3.2 Relationships and equations
With this voltage source system the motor conditions are similar to those which
occur when the motor is connected to a mains supply, except in this case the
voltage waveform contains harmonics and the frequency is variable. The motor
The six step voltage source inverter for induction motors
143
torque is produced by the funamental component of the motor current
and the harmonics can be considered as secondary to the motor's normal
operation.
As with a motor connected to the mains, the current drawn by the motor is
directly dependent on the characteristics of the motor i.e. its magnetising
requirements, its leakage reactance and resistance and its load. The power factor
of this current also takes up its natural value. In this system the inverter is a low
impedance source of power which allows the current level and phase angle to be
dependent on the motor's characteristics only.
The vector diagram of the motor is therefore the same as that explained in
Chapter 1 (Fig. 1.15) where the stator current vector can lag the induced voltage
vector by up to 90 degrees under no load conditions. The locus of current with
changing load torque will follow a path which can be approximated to the circle
diagram. However the point to make here is that the motor characteristics
decide the currents flowing in the circuit and the only influence which the
inverter has is in deciding the voltage to frequency relationship which dictates
the motor's flux condition.
Let us now go back to the supply side of the drive system. Referring to
Fig. 4.2 the DC link voltage of the supply side is dictated by the phase angle of
firing of the supply convertor thyristors. If we initially ignore the supply
impedance and the thyristor voltage drop the mean value of the DC voltage will
be given by:
Vdc = 1-35 x Vs x COS (Alpha)
(1)
where Vs is the supply line RMS voltage and Alpha is thefiringdelay angle (see
Section 3.2.1).
In general the DC voltage at this point will contain a substantial amount of
sixth harmonic and this will be smoothed out by the reactor/capacitor filter so
that the voltage across the capacitor will be relatively smooth and equal to the
mean value of Vdc with the ripple voltage now appearing across the DC link
reactor. In fact this is only true if the current in the DC link reactor is flowing
continuously. When the load current in the circuit is low the current in the
reactor will become discontinuous and the capacitor voltage will rise so that
under no load conditions it will take up the peak value of the DC voltage
waveform. The curves of Fig. 4.11 are typical of a DC link reactor/capacitor
filter and these show that the voltage will depend directly on the load drawn
from the filter.
The value of the capacitor voltage is shown for a fully controlled bridge
operating at different firing angles as the load current approaches zero and you
can see that it is necessary for the phase angle of firing to increase still further
when the load current is low. The critical value of load current varies with the
size of the filter components, particularly the inductance of the reactor and if a
large reactor is used this effect will only be of significance at very low currents.
Below this critical level of current the voltage will be related to the firing angle
144
The six step voltage source inverter for induction motors
by an equation of the type
Vc = 1-35 x Vs x COS (Alpha)
+1-35 x Vs x {COS (Alpha - 30)
- COS (Alpha)} x FN(Idc)
(2)
where FN(Idc) is a complex function of the load current and the filter capacitor
and reactor.
In practice there are a number of supply convertor arrangements which can
be used and this relationship may alter depending on the system used — see
Section 4.4.4.
The voltage applied to the motor is directly decided by this level of capacitor
voltage due to the fact that the output voltage waveform is always arranged to
be the same quasi-sqaure wave shape.
The total RMS value of the motor line terminal voltage will therefore be given
by:
Vm = ^
x Vc
= 0-8165 x Vc
(3)
The performance of the motor will be decided not by the RMS value of this
voltage but by the fundamental value, and a Fourier analysis of the quasi-square
wave shape will show that this fundamental value will be approximately 5% less
than the RMS value, i.e.
Vm(fundamental) = 0-778 x Vc
(4)
This voltage, at the chosen frequency, is then applied to the motor and
reference should be made to Chapter 1 to see that the current drawn from the
convertor, and the motor slip speed etc., will depend directly on the load on
the motor and the applied voltage and frequency. Chapter 1 deals with this
calculation in an extensive or simplified way in Section 1.2.5, for the present let
us now use the simplified way which shows us that the motor current will be
made up of two components, a magnetising current, Imag, and a torque
component, It. Imag is found from the magnetisation curve of the motor and
can be derived from the equation:
[l -
Imag = -0-75 x Isat x log. 1 - ^ — — = £
(5)
where Isat and Vsat are shown on Fig. 1.13 and F and Fr are the actual rated
frequencies. The torque component It will be given by:
It = Vm/(R1 4- R27S1)
Then the total fundamental phase current into the motor will be
Im = ^Imag 2 + If
(6)
The six step voltage source inverter for induction motors
145
The motor power factor equals
COS(£ m =
(7)
I t /I m
The power into the motor is equal to
(8)
Pm = 3 x It x Vm
delay angle of
supply convertor
locus of
critical load
current
0
100
15"
90
capacitor
voltage
percentage of
maximum
open circuit
DC voltage
ie 1-35xVs
30"
45"
80
70
60
\
50
60"
40
30
20 \
%
10
vj
85°
o
^sd
I
90°
current
xj
ioo°
sj
HO"
1
120°
^
Fig. 4.11 The effect of discontinuous current at low loads
The power to drive the load will be given by multiplying the motor input power
by the motor's efficiency and the motor torque can be obtained from this and
the speed of the motor. The motor speed will be given by:
S = 120 x F x (1 - Sl)/P
(9)
and the motor torque in Newton metres by:
Torque = (Pm x Efm x 60)/(2 x n x S)
(10)
where P is the number of poles, SI is the slip in per unit and Efm is the per unit
motor efficiency.
146
The six step voltage source inverter for induction motors
Now let us return to the inverter. If the fundamental line current to the motor
is Im then this will be in part carried by the switches and part by the diodes
depending on its power factor. (See Figs. 4.7 and 4.10.)
The peak value of the switch current will be equal to the peak of the inverter
output current and the mean value of the switch current will be equal to:
Im x (1 + C 0 S ( O ( V 2 x 7i)
(11)
where COS 4>m is the motor power factor. The mean diode current will therefore
be equal to:
I m x (1 - COS </>m)l(y/2 x n)
(12)
If there are no losses in the inverter then the mean current in the DC link will
be equal to the difference between the sum of the switch currents which feed the
link busbar and the diode currents i.e.
Idc(mean) = 3 x 2 x Im x COS (j)J{^Jl x n)
= 3 x y/2 x Im x COS 4>Jn
=
1-35 x Im x COS</>m
(13)
and this is also the value which flows in the DC link reactor. From this it can
be shown that the power crossing the DC link will be the same as the power into
the motor because we are neglecting losses.
In practice the inverter does have some losses and these have to be supplied
by the DC link so that correctly
Idc x Vdc = Inverter input power
With most of the inverter switch types used there will usually only be a voltage
drop of two to four volts in the inverter and the remainder of the inverter losses
result in the DC link current having to increase appropriately. Equation (4)
therefore should correctly be
Vm = -778 x (Vc — Inverter switch volt drop)
(4a)
and the DC link current should be found from
Idc = Inverter input power divided by Vc
(13a)
The ripple current in the DC link capacitor will be related to the mean DC
current but it will also be affected by the motor power factor as shown in Fig.
4.9. It can be found by multiplying the motor line current by the values shown
on Fig. 4.9. It should also be noted, however, that any harmonic content in the
motor current with all flow in the capacitor and must be added into the
calculation.
The current into the supply side convertor will, due to the presence of the DC
link reactor, be of quasi-square wave shape and its RMS line value will be equal
The six step voltage source inverter for induction motors
147
to:
Is = Idc x ^= = 0-816 x Idc
(14)
These relationships can be checked against the results taken on a more
rigorous model of this drive system and shown in Figs. 4.12 and 4.13. These were
taken from a 380 volt, 50 hertz, 25 kW, 4 pole motor when running at 25 hz,
715 rpm while being fed from a 415 volt, 3 phase, 50 hertz mains network.
100
motor
percentage
power factor
total system
efficiency
80
DC current (ampsmean)
60
motor RMS
current
20
>
i
i
i
i
i
50
percentage rated torque
i
i
100
Fig. 4.12 Variation with load torque
Fig. 4.12 shows that the DC link current and the input supply current both
vary with the load torque in an almost linear way, whereas the motor current
is still substantial even at very low torques. This is due to the magnetising
current needed by the motor at all times, which causes the motor power factor
to reduce at low torques. The system efficiency also reduces at reduced torque
due to the influence of the fixed inverter and motor losses. The capacitor ripple
current is shown to remain fairly constant as the load torque changes at about
40% of the rated motor current.
Fig. 4.13 shows how the electrical parameters vary with frequency. The DC
volts and the supply power factor are approximated proportional to frequency whereas the circuit currents are relatively unaffected by it. The
148
The six step voltage source inverter for induction motors
upturn in the value of current necessary to generate the same torque at low
frequency as at high frequency is due to the lower overall efficiency at the low
speeds.
100 - 5 0 0
DC
voltage
volts
/
total system
efficiency (%
80
60
40
20
v
-/
/
0
/
XL
/
/-/-
?\ /
. \
DC current at
/ rated torque
*—
"
motor current
/ Q t rated torque
DC current at
50 °/o torque
\
>
firing delay
angle alpha
i
i
I
i
A. supply
power
factor (•/.)
y
i
i
25
frequency hertz
i
\
i
50
Fig. 4.13 Variation with frequency
4.3.3 Examples of calculations
1) Calculation of rated currents and voltages
Question
A 10 kW, 2 pole, 3 phase, centrifuge motor needs a supply of 230 volts line at
a frequency of 300 hertz in order to run at nearly 18000rpm and under rated
conditions it operates at 81% efficiency, a power factor of 0-85 and a slip of 2
per cent. A quasi-square wave transistor inverter is to be used to drive it and this
is to be fed from the 60 hertz mains via a transformer having an output voltage
of 300 volts line. Find the approximate values of following under rated motor
conditions assuming that the inverter efficiency is 90 per cent and the supply side
convertor has no losses:
The mean value of the DC link voltage.
The mean value of the DC link current.
The six step voltage source inverter for induction motors
149
The supply RMS current.
The firing angle of the supply convenor.
Answers
When the motor is operating at 230 volts fundamental line voltage the DC link
voltage will be given by equation (4a) i.e. Vc = Vm/-778 + inverter switch
voltage drop.
Let us assume that the inverter switch voltage drop equals four volts. Then
Vc = 230/-778 + 4 = 300 volts DC
The motor input power = its output power divided by its efficiency,
i.e.
=
10,000/81
= 12346 watts.
The inverter input power equals the motor input power divided by the inverter
efficiency.
i.e.
= 12346/9
=
13718 watts.
Therefore the DC link current from equation (13a) is given by
Idc =
13718/300 = 45-7 amps DC.
From equation (15)
IS = IdC X yjljyji
= 37-3 amp RMS line.
From equation (1)
COS (Alpha) = Vdc/(l-35 x Vs)
= 300/(1-35 x 300)
= -741
Alpha = 42 degrees approximately.
2) Motor magnetisation
Question
In the above example what is the approximate magnetising current under rated
operating conditions and what would it increase to if the DC voltage was
increased to 330 volts? Assume that Vsat equals 300 volts line.
Answer
The total VA into the motor is equal to the input watts divided by the power
factor. Therefore Input VA = 12346/0-85 = 14525.
150
The six step voltage source inverter for induction motors
The input line current to the motor
= 14525/(230 x 1-732) = 36-5 amps line.
Therefore the magnetising current will be approximately equal to:
Imag = 36-5 x SIN (ACS (0-85)) = 19-2 amps line.
Now from equation (5)
19-2 = - 7 5 x Isat x LOG (1 - 230/300) = 109 x Isat
Therefore Isat = 17-6 amps line.
From equation (4a) motor volts with a 330 volt DC link will be equal to 254
volts line.
Therefore the new magnetising current will be given by:
Imag = - 7 5 x 17-6 x LOG (1 - 254/300)
= 24-8 amps line.
Answer: New Magnetising Current = 24-8 amps line.
3) Conditions at reduced speed
Question
If the same motor is operated at 5000 RPM at 50 per cent of rated torque under
constantfluxconditions, what will be the inverter frequency and the DC voltage
and current if the motor efficiency under this condition is 80 per cent, the
inverter efficiency 90 percent and the inverter switch voltage drop is assumed to
be 3 volts.
Answers
Slip speed is approximately proportional to torque. Therefore for half torque
the slip will be 1 per cent of rated synchronous speed i.e. 1 per cent of 18000
RPM = 180 RPM. Therefore for the motor to run at 5000 RPM then the
frequency needs to correspond to 5180 RPM.
ie F
- = 3o°x i S =863hertz
On the basis of constant motor flux the voltage applied to the motor must be
approximately proportional to frequency, therefore under this condition
Vm = 86-3 x 230/300 = 66-2 volts.
From equation (4a)
Vc = Vm/0-778 + inverter switch voltage drop
= 66-2/-778 + 3
= 88-1 volts DC.
The six step voltage source inverter for induction motors
151
The power out of the motor will be proportional to speed and torque and
therefore at this condition
Motor power output = 10,000 x 0-5 x 5000/17640
= 1417 watts.
Power into the motor = power output divided by efficiency
= 1417/-8 = 1772 watts.
Power into the inverter therefore equals
= 1772/-9
= 1969 watts.
From equation (13a)
Idc = 1969/88-1
= 22-3 amps DC.
4.4 Practical circuit design considerations
When this drive is manufactured for commercial sale and for use for a wide
variety of potential applications it is necessary to add auxiliary components to
ensure satisfactory and reliable operation. The components need to be protected
against unusual operating conditions e.g. motor overload, supply power loss,
faulty operation of the circuit, etc. so that unnecessary damage is not caused and
so that the unit is disconnected from the supply quickly.
Some of the components, particularly the semiconductors will dissipate some
heat losses and some means of cooling may be required to ensure that they do
not overheat.
Facilities for automatic control over the drive will be incorporated to ensure
that the motor and the drive are always used in the optimum way — the
principles of control will be described in the next main section. However, the
drive will contain the necessary electronic circuits for automtic control and the
necessary low voltage power supplies and interfacing measurement and relay
circuitry essential for its correct operation.
The drives from different manufacturers may also contain variations in the
circuitry so far described in this chapter, there are always many possible solutions
to the same problem and I intend to deal with the main variations in this section.
This section also contains some information on the factors which decide the
specification and requirements of the main components of the drive power
circuitry. This part is not intended to be an exhaustive study but only to point
to the major principles which decide the size and type of components which are
used in the drive.
152
The six step voltage source inverter for induction motors
The aim of this section is therefore to assist in the practical understanding
of the drives of this type which are in regular use in industry, colleges and
laboratories throughout the world.
4 A.I Over current protection
The current in this circuit is normally kept under close control by using the
supply side convertor as described in section 4.5 but due to the presence of a
large DC link capacitor the supply side convertor is unable to control the
inverter and motor currents in any precise way during transient effects and fault.
The DC link capacitor is a large energy store and it can cause large currents to
flow in the circuit particularly if the inverter malfunctions. The situation will be
studied further with reference to Fig. 4.14 which shows the inverter and motor
part of the system.
follow up
fault current
initial fault
path
I
motor
Fig. 4.14 Fault current paths
During correct and normal operation of the inverter the current flowing is
limited by the reverse voltage generated in the motor with one of the positive
side inverter switches connecting the DC link positive to the appropriate motor
terminal and one of the negative switches connecting a different motor terminal
onto the DC negative. Although the inverter switches are being cycled on and
off at high frequency in response to the changing motor voltages this situation
always occurs when the inverter is working correctly and the motor is correctly
magnetised. It should never be possible for the motor back emf to be lost or for
the inverter switching to be such that the current bypasses the motor, passing,
for example, through switches 1 and 4 or 3 and 6.
Although such faulty circumstances are never supposed to occur they can and
do happen and the result is that the DC link capacitor is effectively short
circuited either straight through the inverter or through the motor windings.
Clearly the worst case condition is if the inverter malfunctions and the two
switches in one phase are switched on together, thus short circuiting the
capacitor instantaneously. From Fig. 4.3 you will see that the switches in one
The six step voltage source inverter for induction motors
153
phase are switched ON and OFF alternately and normally the oncoming switch
is not allowed to turn ON until the outgoing one has fully recovered its blocking
ability. Any excessive current or switch temperature or firing malfunction is
likely to prevent a switch regaining its block capability causing a commutation
failure which immediately results in a short circuit across the DC link through
the offending phase switches. The currents which flow as a result of this fault can
instantaneously be very large and they are only limited by the resistance and
inductance of the fault path, e.g. the capacitor, the DC link connections to the
inverter phase and the impedance of the two switch inverter phase. Because of
the large capacitor this fault can normally damage the switches beyond repair
if it is allowed to persist and it is essential to include arrangements to limit the
size and rate of rise of the current and to cut it off before it does too much
damage, hopefully, before any damage occurs.
The simplest solution to this problem is to put high speed fuses in the fault
path i.e. in the DC link or in the inverter phases to open the fault circuit and
this is done in some designs. However this solution may bring with it other
problems due to the voltages occurring across the fuses when they blow.
In present day designs it is usual to employ more complex means which enable
the fault to be controlled statically using the inverter switches. As there are
always two switches in the fault path the preferred method is to arrange for the
switches to turn off immediately the fault current is detected and this approach
is adopted in some designs. However, whether this approach is possible depends
on the type of switches employed and the margin which is allowed between
normal running currents and the peak current which can be turned off. The fault
usually occurs because one switch has been unable to regain its blocking ability
and this usually occurs just as the complementary switch has been switched ON.
Depending on the details of the switch it may not be possible to turn it OFF
immediately and the fault current will continue to rise rapidly. There is also the
question of the level of fault current one can detect and the speed it can be
detected. Clearly a fault current is not a fault current until it rises above the
normal peak load current which would be expected, and a margin is required
above this level to ensure that incorrect operation of the protection circuit does
not occur. Once a fault has been detected there will then be a time delay before
the switches can be turned off and these factors decide the level of fault current
which may be reached.
Once a fault has been detected it may be possible to reduce its effect by turning
all the inverter switches ON therefore sharing the capacitor discharge current
between the three inverter phases. This is done on some designs but clearly it
would be difficult to mix this approach with that of switching the switches OFF
to cut off the fault.
In all circuits some impedance usually in the form of reactors would be
included to limit the rate of rise of fault so making the fault conditions predictable. In all cases as soon as a fault is detected the supply convertor is instructed
to reduce the supply current to prevent it contributing to the fault and most
drives would be switched off when such a fault occurs.
154
The six step voltage source inverter for induction motors
4.4.2 Overvoltage protection
In its basic form this circuit does not suffer from too many overvoltage problems
because the DC link capacitor normally prevents rapid voltage change occurring.
The normal semiconductor switching voltages do occur and they need to be
limited by snubber circuits across most of the semiconductors. It may also be
necessary to protect against transients from the mains supply system on the
input to the drive by fitting resistor capacitor circuits or overvoltage suppressors.
The only way in which the DC link voltage can normally increase is if
regeneration from the motor occurs. Under this condition the current will be
completely out of phase with the voltage and most of the motor current will flow
in the reactive feedback diodes into the DC link capacitor thus causing it to
overcharge. In practice, either this condition is prevented by the control method
used or the regenerated energy is absorbed by additional circuits thus preventing
the voltage rise.
4.4.3 Factors affecting the specifications ofx the main components
Supply side convenor
The supply side convertor needs to be capable of producing a variable positive
DC link voltage and sufficient power for the systems needs. If it also has a small
ripple voltage content and is capable of reducing the current quickly then these
will be additional but not essential features. Half controlled or fully controlled
thyristor bridge circuits are most common and some of the variations used are
described in Section 4.4.4.
In all cases the convertor needs to be protected against transients and
variations in the mains supply voltage and appropriate voltage safety margins
and surge suppression circuits will be included.
The supply side arrangements may include facilities for slow charging of the
DC link capacitor on initial switch on. This is usually a large capacitor and if
no special measures are taken then a very large inrush current could be caused
and this could damage the supply side convertor.
The DC link capacitor
If we ignore the fact that this capacitor may be affected by the specific inverter
switch commutation methods employed, its main purpose is to allow the motor
to draw whatever current it requires to operate against the connected load. The
capacitor ensures that the inverter is a low impedance source of current for the
motor.
As described earlier it provides a path for theflowof the motor magnetisation
current which appears in the capacitor as a sixth harmonic of the motor
frequency.
The capacitor is there to ensure that the DC link voltage is relatively smooth
in spite of the harmonic voltages coming from the supply side convertor and the
harmonic currents fed into the link by the motor side inverter.
The six step voltage source inverter for induction motors
155
For safety reasons it is necessary to provide some means of discharging the
DC link capacitor when the drive is turned off, otherwise the capacitor voltages
could remain at a dangerous level for a long time.
The DC link reactor
This is provided to assist the DC link capacitor to maintain a smooth DC
voltage and to ensure that the supply convertor current remains continuous
down to quite a low level (refer to Section 4.3.2). It will also assist in limiting
the rate of rise of the fault current which can be contributed by the supply
convertor. The voltage across it will be the harmonic ripple coming from the
supply convertor at three or six times the supply frequency depending on the
supply convertor circuit used. (See Fig. 3.10.)
The motor inverter
The switches in the motor inverter can be transistors, forced commutated
thyristors or gate turn off thyristors and clearly the arrangements made in
the inverter will be decided by which switches are being used, see Chapters 2
and 3. However, in all cases there are common factors which will be dealt
with here.
Every switch will have a reverse diode across it so the switch itself will never
have to support any significant reverse voltage. It will also have to be capable
of being turned off successfully with only the diode voltage drop as a reverse
voltage. The switches have to be capable of supporting the maximum DC
voltage which can occur. Under some circumstances the capacitor voltage can
rise to the peak value of the supply convertor waveform i.e. 1*414 times the
maximum RMS value of the mains supply, and this has to be allowed for. In
addition it may be possible for the capacitor to become overcharged due to
regeneration from the motor and some allowance may need to be made for this.
Because of the presence of this large DC link capacitor, any transient voltage
spikes which occur across the switches will usually be the result of the operation
of the switches themselves and snubber circuits will probably be needed to keep
the voltage margins of the switches reasonable.
The maximum level of current flowing in the switches occurs under unity
power factor load conditions where it has to carry full half cycle of motor
current. As the power factor reduces then more of the currentflowsin the diode
until at zero power factor a quarter of the cycle is carried by both the switch and
the diodes.
Reference to Fig. 4.7 shows that at the end of a switch's conduction period
the current immediately transfers to the opposite phase diode and if no special
arrangements are made this transfer will take place instantaneously causing very
high rates of rise and fall of the current. Reactors may be necessary to limit the
rate of change of currents at this point. Thisfigurealso shows that when a switch
is turned ON the current will not normally switch into it until the diode has
completed its conduction period.
156
The six step voltage source inverter for induction motors
The motor
It is normally quite safe to use standard induction motors on this drive. The
motor current waveforms under high speed, high load conditions are usually
close to sinusoidal and the terminal voltage is usually under control at all times,
having a variable magnitude quasi-square shape.
This voltage waveform does lead to an increase in iron and stray losses in the
motor and some derating may be necessary to allow for this.
4.4.4 Circuit variations
The main differences between the power circuits of different manufacturers is in
the supply convertor arrangements. All drives of this type have bridge type
inverters with six switches and with feedback reactive diodes across the switches.
The supply side convertor is not a very critical item from the design and
specification point of view and therefore it is fulfilled by a variety of arrangements from different designers. The most usual arrangements for the supply
convertor are shown in Fig. 4.15. The six pulse bridge in Fig. 4.15(a) is often
used but this does lead to a wider range of discontinuous current operation or
a larger DC link reactor. Fig. 4.15(b) helps in this respect because it prevents the
DC voltage on the convertor from reversing. The same effect can be produced
in circuit (a) by special flywheel firing of the thyristors. (See Bibliography.) The
half controlled bridge of circuit in Fig. 4.15(c), with a flywheel diode will
produce the correct range of voltage but with third harmonic ripple of substantial magnitude and a larger DC link reactor will be the result. It does however
improve the input power factor albeit at the expense of a low even harmonic in
the supply current. In general this is not such a good circuit as the others.
The system in circuit of Fig. 4.15(d) using a diode bridge rectifier and a series
chopping will give very good overall performance. The voltage ripple to the
chopper is very low and the supply power factor is always very high whatever the
speed of the motor. The chopper is usually a transistor or GTO switch operating
at relatively high frequency in order to reduce the size of the DC link reactor.
The necessity to cope with motor load regeneration and braking may lead to
additional circuit arrangements. Three methods are in regular use with this type
of drive:
1) A 3 phase set of resistors which are switched onto the motor connections when regeneration is detected (usually by a rise in the DC link
capacitor voltage.)
2) A similar arrangement but this time on the DC link itself. A switched
resistor is applied across the capacitor to dissipate the regenerated
energy. If a semiconductor switch is used it may be operated like a
chopper to give control over the amount of regeneration power being
absorbed.
3) The ultimate is to connect a reverse convertor on the supply side to
enable the regenerated power to be fed back into the mains supply
network. Again this is likely to be brought into use by the rise in the
DC link voltage.
The six step voltage source inverter for induction motors
-L
T
fully
controlled
thyristor
bridge
—r-
as a with
flywheel
diode
half controlled
bridge with
flywheel
series chopper
d diode bridge with chopper
Fig. 4.15 Alternative supply convenor arrangements
157
158
The six step voltage source inverter for induction motors
4.5 Overall control methods
There are two parameters only which can be controlled in this six step system.
Normally the control angle of the supply side convertor and the frequency
of inverter switching. These are the only two independently controllable parameters and these must be used together to achieve the necessary degree of
control over the drive.
The phase angle of the supply convertor firing directly controls the level of
voltage on the DC link and applied to the motor. Hence it can be used to control
the level of current in the system if required.
The frequency of the inverter alters the speed of rotation of the motor stator
field and has to be directly related to the rotor speed if proper control over the
motor is to be achieved.
If this system is going to be controlled satisfactorily the following points have
to be considered:
1) The magnetisation of the motor has to be controlled to ensure that
there is sufficient flux to develop the required torque. The most usual
way to achieve this is to keep the ratio of terminal volts to frequency
relatively constant (see Chapter 1) because induced motor volts
are proportional to flux times frequency. If the flux level is allowed
to rise much above the rated motor level the magnetising current
will rise sharply due to saturation, thus increasing the inverter
current.
2) The currents in the circuit must be closely controlled to prevent
overloading the inverter, which usually has a limited current capability.
3) To obtain the best performance the motor should be operated at
a low value of slip. The slip speed being the difference between the
rotor speed and that of the rotating field produced by the inverter
frequency.
4) Sudden changes in slip can cause large torque changes, and even
torque reversal which causes regeneration of the motor and load
inertial energy back through the inverter to the DC link. Such changes
can be caused by a fast change of inverter frequency and the control
system may need to prevent this occurring.
5) Most system of this type do not include any direct measurement of the
motor speed and hence accurate control over slip is not normally
considered. In general, control systems rely on electrical measurements of current, voltage and frequency only.
6) The current in the supply side of the DC link capacitor is related to
the real power drawn by the motor and, if the motor flux is kept
constant over the frequency range, the DC current is proportional to
motor torque. The motor current contains a substantial additional
magnetising component and this causes the inverter current loading
to be increased.
The six step voltage source inverter for induction motors
159
7) Speed control is usually obtained by controlling the frequency and
then compensating for the slip speed by boosting the frequency in
proportion to load torque.
4.5.1 Supply convenor control
The voltage applied to the motor is directly controlled by the supply side
convertor and hence this convertor is usually arranged to vary this voltage
approximately in step with frequency. The level of DC voltage also alters the
current flowing in the circuit and it is usual to combine the control over both
voltage and current into the supply side convertor.
The voltage control can be based on a DC voltage measurement or a measurement of the motor terminal voltage, whichever is the most convenient. In most
drives the DC voltage is used because it is a direct and smooth measurement and
because it is unaffected by changes in inverter frequency.
The most critical current in the system is the inverter current and hence it is
usual for any closed loop current control system to be based on a measurement
of motor current rather than input or DC link current. It is also normal to
include current limiting features to prevent the inverter being overloaded.
The motor current is not however directly proportional to torque and
therefore it is not usually used for the slip compensation circuit which is often
incorporated into the supply convertor control. A DC link or AC input current
measurement is more likely to be used for this purpose.
4.5.2 Inverter control
The frequency of the output supplied to the motor is directly controlled by the
inverter usually via an oscillator based switch firing system. This would usually
be a voltage controlled oscillator in order that some degree of closed loop
control can be included.
The inverter usually also has to play an important part in controlling the
motor flux in order to obtain optimum output torque. It is the ratio between
voltage and frequency which effectively decides the motor flux and it is usual to
keep this ratio sensibly constant during operation over the full frequency range.
One regularly used means of implementing such a control is to employ the DC
or AC output voltage as the reference value for the oscillator frequency, so that
the inverter is always operated at the correct frequency to match the actual
voltage being produced. This approach will always ensure that the correct flux
level is produced and it also serves to protect the system against the adverse
effects of regeneration. If motor energy is fed back into the inverter it will cause
the DC link voltage to increase. If this method of frequency control is used, the
increase of voltage will cause the frequency also to increase so reducing the
likelihood of regeneration. This therefore provides a means of controlling the
amount of braking which is allowed to occur.
The alternative means of maintaining control of flux, namely to set the
frequency and then let the voltage control of the supply convertor ensure that
160
The six step voltage source inverter for induction motors
the correct voltage to frequency relationship exists, is sometimes used but in this
case other additional methods of controlling regeneration will be needed because
the supply convertor will not be able to prevent the DC link voltage rising.
4.5.3 A typical overall control scheme
Fig. 4.16 is an example of the sort of control scheme used for this type of six step
voltage source inverter drive shown in block form.
The supply side thyristor bridge is gated by a set of six gate pulse circuits fed
from a phase shift firing circuit (1) in Fig. 4.16 which is synchronised to the
mains sine waves. An input voltage to this firing circuit controls the phase
position of thefiringpulses, hence changing the output voltage from the bridge.
The signal to thisfiringsystem is produced from a high gain current control loop
amplifier (2) based on a measurement of motor current which serves to ensure
good protection of the inverter against overcurrents. The reference to this
current loop has a preset limit which decides the maximum circuit current
allowed.
inverter switch
drive circuits
signals circuit
speed
setting
current
amp
©
speed
amp
o>
voltage
controlled
slip
© oscillator
compensation
signal
motor
current
v/f
function
Fig. 4.16 A typical six step voltage source drive control scheme
This current reference is obtained from the speed amplifier (3) which has a
speed reference obtained from the drive set up potentiometer (or equivalent
signal) and a measurement signal of DC link voltage; a small signal based on DC
link current is also introduced to compensate for the reduction of speed due to
slip.
The inverter frequency is an open loop control based on the DC link voltage
measurement and a voltage controlled oscillator (4). The V/f function block (5)
is included to ensure that the optimum ratio is used at all frequencies because
The six step voltage source inverter for induction motors
161
at low frequency the stator resistance becomes more dominant and a higher V/f
ratio than at high frequency is required to compensate for this, and to ensure
that maximum motor torque can be produced if required.
4.6 Performance and application
This drive is a general purpose drive suitable for a wide range of straightforward
uses but not capable of the high quality of performance which can be obtained
from some other systems. The main limitation is in the use of the six step square
wave output voltage waveform. Although motors can work quite reasonably at
the higher speeds and frequencies using this arrangement, they do not perform
very well at low frequencies. The level of harmonics in the motor current
waveform will be higher at the lower frequencies because the motor inductance
is not so effective in smoothing them out. The motor MMF waveform therefore
tends to step round the stator with six distinct steps per cycle and the rotor tends
to cog round also. In most normal cases this effect is only important below
about 10 per cent of the drive's rated frequency and the motor is usually only
accelerated through this low speed region.
The system is however quite good at higher frequencies and operation at up
to 500 hertz can readily be obtained with forced commutation thyristor inverter
switches. With transistor and GTO switches much higher frequencies can be
obtained but such frequencies are not normally necessary for motor drives
where the mechanical stresses will limit the motor's speed.
This system is generally used for standard induction motors where the motors
can usually be operated at up to say twice the normal motor frequency i.e. 100
to 120 hertz.
Over the normal range of operating frequency the motor currents tend to be
reasonably sinusoidal with only a small amount of harmonics leading to smooth
and satisfactory performance.
Their use with standard fixed frequency motors rather than with motors
specially designed for them leads to another feature which is often included in
these drives, namely, the ability to operate at higher than normal frequency at
constant voltage and hence reducing motor flux. The economic use of standard
motors dictates that one should use them at their normal rating of say 415 volts
50 hertz or 460 volts 60 hertz, etc., and one should select a drive of appropriate
rating in order to get the most out of the motor. If then it is advantageous to
run the motor at above its normal speed and frequency the inverter can usually
be arranged to run at the higher frequency but nut give any more volts. As a
result constant voltage higher frequency operation with reducing motor flux is
used and is a feature of many such six step drives. This is shown in Fig. 4.17.
It is possible to use this voltage source drive for single motor or multimotor
loads because the motor is generally left to take care of itself. When a number
of motors are connected to the same drive they will all be supplied with the same
162
The six step voltage source inverter for induction motors
values of voltage and frequency but they will be able to draw load and magnetising
current according to their own needs. Also if they are sometimes mechanically
connected together as may be the case on a roller table or conveyor application,
the motors will share the load reasonably well due to the speed droop caused by
the slip.
frequency.
Fig. 4.17 Operation above standard frequencies
Most drives of this type are not regenerative i.e. they are not used to brake
the load and feed power back into the mains. Sometimes dynamic braking
facilities are added by connecting resistance loads to the motor or DC link
terminals via static or electro-magnetic switches.
Most of these drives have a limited overload capability due to the design of
the inverter being current limited and the consequences of excessive current
being inversion failure faults. It is therefore necessary to know the precise
currents which will be demanded by the motor when selecting the size of inverter
to be used.
4.6.1 Torque/speed characteristics
This drive is normally operated over a speed range from 10 per cent of nominal
frequency to say 150 per cent of nominal frequency and operation at very low
speed is only used during starting. The torque capability over the speed range
The six step voltage source inverter for induction motors
163
is usually limited by the current which the drive can provide and this is usually
restricted to a value just above the rated requirement of the motor. Hence the
size of the drive needs to be selected after consideration of the peak torque
required from the motor. If the control system is able to maintain the motor air
gap flux at the rated level at any speed then it is possible to produce high peak
torques over the whole of the speed range.
Within this maximum torque limitation, it is possible to set the motor operating at any value of torque and at any value of speed at will, by the appropriate
choice of frequency.
motor rated
nominal sinusoidal
torque
peak capability
continuous
capability
50
100
150
percent nominal speed
200
Fig. 4.18 Standard induction motor capabilities
If the motor is to be used at higher than its rated frequency it is usually
necessary to reduce motor flux so as to keep the motor voltage within the
capability of the inverter. Fig. 4.18 shows a typical torque/speed of a six step
drive with the upper limit based on a typical inverter current capability of
120 per cent of normal rated motor current. This upper limit is therefore a peak
capability which can only be achieved for short periods of time.
The limitation in peak torque at low speeds is due to the reduction in motor
and inverter efficiency. The inverter is still operating at maximum current but
less of it is available to generate torque.
164
The six step voltage source inverter for induction motors
Due to the increased heating produced by the voltage and current harmonics
generated by the inverter drive the motor is only able to operate at a reduced
continuous torque even at rated speed. At lower speeds the motor cooling may
not be so effective and this limits the continuous rating at reduced speed. The
dotted curve of Fig. 4.18 shows the continuous capability of a standard totally
enclosed fan cooled motor with a six step voltage source inverter of appropriate
size connected to it.
4.6.2 Speed control accuracy
Most drives of this type are frequency controlled with, if necessary, open loop
slip compensation to give a reasonably close control over the speed, in order to
avoid the need tofita tacho-generator to measure the speed. The speed accuracy
with varying load then depends on the precision with which the slip compensation
circuit has been set up. It is usually not possible to obtain an optimum setting
for all speeds and loads and hence some variation will in general occur over the
range. The variation of speed if the drive is set up correctly should not normally
exceed 20 per cent of the rated slip speed of the motor.
More accurate speed control can be achieved by adding a digital or analogue
tacho-generator to the motor shaft to use as a feedback into an overall speed
closed loop control system.
4.6.3 Supply side power factor and harmonics
When set up on a constant motorfluxbasis the DC voltage of this drive will vary
approximately in proportion to the frequency and speed of the drive. This
motor
torque
lines /
1007^ /
M
1007.
/
/
§
607. /
•^ /v
/
/
y \
V
/
/ v
/
/
/
90 7.
/
y
/
s \
zrJ/,',K'
/'*&>
-^i
reactive KVAR
Fig. 4.19 Drive input power factor chart
motor
speed
807.
lines
/ ' 60'/.
/
507.
>^—*—
The six step voltage source inverter for induction motors
165
means that the supply power factor will also vary with the speed in an approximately linear fashion, as can be seen in Fig. 4.13.
The DC voltage under rated conditions has to be set up to match the motor
voltage requirements if optimum torque is to be obtained so that the power
factor under rated conditions will depend on the value of the supply voltage in
relation to that of the motor. In general the optimum condition is when the
supply and motor voltages are the same. If the supply voltage is higher than the
motor's then the rated power factor will reduce and if the supply is below the
rated motor voltage then operation at reduced motor flux will have to be
arranged reducing the motor's torque capability.
Fig. 4.19 shows a composite supply side vector diagram for a drive of this type
showing the likely supply conditions under a wide range of speed and load
torque. From this type of diagram it is possible to plot the vector condition for
any load and speed of the motor. The two conditions shown boldly are, rated
speed and load and 40 per cent speed with 60 per cent motor torque.
Due to the influence of the DC link reactor the DC current will be reasonably
smooth leading to approximately quasi-square waveshapes of input current to
the drive. Hence the input current will contain harmonics related to the supply
frequency only. With a six pulse bridge supply side convertor the harmonics will
be restricted to those at:
6 x fs x (N + 1)
and 6 x fs x (N — 1) frequencies
where N is any integer
and fs is the supply frequency.
The amount of these harmonics will not normally exceed:
1/h times the fundamental current
where h is the harmonic number.
Chapter 5
The pulse width modulated voltage
source inverter system for
induction motors
5.1 Introduction
The difficulty with most six step inverter systems is that their performance at low
speeds is not very good. In fact, in many cases it is unacceptable to dwell in the
low speed region at all. The stepped nature of the stator rotating field onto the
motor causes the torque to be applied in pulsations rather than smoothly. Hence
most six step systems have a limited range over which acceptable performance
is achievable.
This pulse width modulated system is the most widely used method of
improving the low speed performance of DC link inverter systems. The principle
is to use high speed switching to enable the motor current waveshapes at low
speed to be more sinusoidal and hence lead to a smoothly rotating magnetic field
in the motor.
The result can be extremely good performance at low speeds as well as high
and the ability to control the motor accurately around zero speed.
As the technique is basically electronic its cost has been reducing steadily as
large scale integrated circuits and microprocessors have tumbled in price, so that
this system is nowadays often employed for general purpose drives where the
improved low speed performance may not really be needed.
Although this system is by no means new, having been used certainly in the
late 1960's, its use has increased recently because of the availability of faster
switching devices like transistors and gate turn off thyristors. This technique
involves switching the inverter at a rate at least ten times the maximum output
frequency desired and hence good switching performance is essential for this
system.
5.2 Principles of operation
This is a DC link inverter system with the mains power being rectified to
produce DC and a self-commutated inverter to invert the DC into AC of
variable frequency to the motor. However before proceeding further it is necessary
to explain the principle of Pulse Width Modulation in some detail.
The pulse width modulated voltage source inverter system
167
5.2.1 Pulse width modulation
With six step systems the principle is to switch the current or voltage onto the
motor windings once per half cycle so as to produce a square or quasi-square
waveshape. The Pulse Width Modulation (PWM) principle is to switch the
voltage on and off onto the motor many times during each half cycle and to vary
the frequency of the pulses and the width of the ON pulses in relation to the
OFF, so as to simulate a sinusoidal shape for the voltage. With this technique
it is not necessary to change the level of the DC voltage as variation in the
magnitude of the voltage applied to the motor can be obtained again by varying
the width of the applied pulses.
Fig. 5.1 This assembly is the complete drive fora 15 HP, 415 volt, 3 phase, 50 hertz induction
motor. It is a pulse width modulated unit, using transistor switches. These are mounted
directly onto the aluminium chassis for cooling and some of the transistors can be
seen at the lower right hand side. The DC link capacitors are in the lower centre of the
picture. The lower printed circuit card contains the PWM generation electronics and
the card above contains the control electronics. (G.E.C. Industrial Controls, Ltd.)
Fig. 5.2 shows the basic principle. If a complete half cycle is produced by the
application of a large number of pulses — in this case equally spaced — and the
width of the pulses is varied according to a sinusoidal rule then the average value
of the pulses will follow the sinusoidal shape. If this waveform was of the voltage
168
The pulse width modulated voltage source inverter system
applied to an inductive load the current flowing would be basically sinusoidal
with harmonics only related to the high frequency of the pulses.
The situation depicted in Fig. 5.2 shows almost the maximum sinusoidal
shape which could be produced by the level of DC voltage chosen. In fact, the
absolute maximum will be when the peak of the sine wave corresponds to the
DC voltage level.
1 0
i
Fig. 5.2 Pulse width modulation
DC
(ink
voltage
a
107» of maximum output
t
b 50*/. of maximum output
Fig. 5.3 PWM voltage variation
The pulse width modulated voltage source inverter system
169
If a reduced value of voltage is required then the widths of all the pulses must
be reduced in the same proportion i.e. if their ON times are all halved then the
output voltage will be half the maximum value referred to in Fig. 5.2. Fig. 5.3
shows the conditions appropriate to 10 per cent and 50 per cent of the maximum
voltage respectively.
a
80*/o of maximum volts
DC
/
/
\
/
\
s
/
/
>
/
s
\
f
\
s
average value
h
k
7
DC-
U of Tiaximum volts
s
s
Fig. 5.4 PWM voltage variation
The frequency of the output sine wave produced can also be altered at will,
either by altering the time between the high frequency pulses or by altering the
number of high frequency pulses which occur in each half cycle.
In practical circumstances the frequency of the high frequency pulses is
limited by the characteristics of the particular switches and the end result is that
at low inverter output frequencies there will be a large number of high frequency
switching pulses per half cycle, whereas at higher frequencies there will be a
reduced number of pulses. This turns out to be satisfactory because the result
is that the low frequency low speed waveforms are very good with very low
170
The pulse width modulated voltage source inverter system
harmonic contents. The poorer waveforms with only few switching pulses per
half cycle occur at higher speeds where the quality of the waveform is less
important due to the inductance occurring in the motor circuit. In practice,
many pulse width modulated drive systems actually operate in the six pulse,
quasi-square wave mode at top speeds.
a
lowt frequency
-— —-
b
•*>
•^»
hlg h fre que
/
/
-«-
s
\
\
/
s
/
s
s
_
Fig. 5.5 PWM frequency variation
Figs. 5.2 and 5.3 show a situation based on two levels of DC voltages being
available, zero and the DC link voltage. Reference back to Chapter 3 will show
that in most inverter bridges the two levels of voltage available are when the
positive switch is ON, when the positive voltage will be available and when the
negative switch is ON. In most cases the voltage can only be switched between
these two levels and the PWM arrangements have to be related to this. Figs. 5.4
and 5.5 show how this is arranged; an average zero is produced by the positive
switch being ON for the same time as the negative switch. Fig. 5.4 shows how the
voltage is varied under these conditions and Fig. 5.5 how the frequency is varied.
The pulse width modulated voltage source inverter system
171
There are two basic types of PWM wave, those which are not synchronised
with the actual frequency being generated and those which are and these will be
briefly explained.
low frequency
low voltage
b
triangular carrier wave
switching pattern
high frequency , high voltage
Fig. 5.6 The unsychronised triangular wave method
The unsynchronised triangular wave method
The conventional way of producing the necessary firing patterns to produce
these PWM waveforms has been to use the interaction between a sawtooth
shape carrier wave and a low frequency waveform of the desired output shape.
The principle is for the low frequency wave to be identical to the output
waveform required from the inverter i.e. its magnitude and frequency and
772
The pulse width modulated voltage source inverter system
waveshape being those required on the inverter output. The sawtooth wave has
a frequency equal to the desired switching rate and a magnitude in excess of the
maximum size of the low frequency wave.
Fig. 5.6 shows this principle and the firing patterns generated by the intersections of the two waveforms under two different low frequency waves and
hence two different inverter output conditions.
This method can be used very satisfactorily if the frequency of switching can
be set at a value of at least 20 times the output frequency required and if the
actual times to switch are insignificant. This situation can prevail with the use
of transistor switches and with systems only operating with a low output
frequency.
When higher power levels are required the limitations of the switches
themselves become more important to the performance of the PWM system and
more complex methods have been found to be necessary.
The problems are that:
(a) Higher power switches can only be switched at frequencies up to
between 500 and 1000 hertz.
(b) It is not possible to switch them from ON to OFF and back quickly.
Once a switch has been switched ON it has to remain ON for a
specific minimum time. Similarly once it has been switched OFF
it has to stay OFF for a definite time before being switched ON
again.
(c) As a result of (b) continuous high frequency switching at the higher
output frequencies means a considerable reduction in available
voltage output. At high powers this is important and the fact that a
reduced number of switchings means an increase in output power
leads to the necessity to steadily reduce the number of pulses per half
cycle until only one switching per half cycle takes place at maximum
output frequency. Hence it is necessary to drop pulses off as the
voltage and frequency is increased. Unfortunately the unsynchronised
method will result in these pulses being lost in an indiscriminate way
and sudden changes in the circuit currents can be caused when the
switching pattern changes.
With this unsynchronised triangular wave method of producing the firing
pattern, pulse dropping at high outputs is produced by allowing the size of the
modulating sine wave to exceed the size of the triangular wave. As a result, as
shown in Fig. 5.7, there are no crossing points between the two waves at the
peak of the sine wave and hence the number of pulses are reduced. If the
modulating wave is allowed to get very large then only one switching per half
cycle will occur.
The problems caused by pulse dropping and by interference between the
carrier wave and the modulating wave used in this method, have led to the
development of methods where the two waveforms are synchronised together at
all times.
The pulse width modulated voltage source inverter system
173
Fig. 5.7 Pulse dropping
The synchronous gear changing PWM method
The only satisfactory way yet found to overcome the above limitations of the
simple modulated triangular wave method involves keeping the switching
frequency in synchronism with the output voltage wave i.e. keeping the number
of high frequency pulses per half cycle constant. However, because of the
limitations in switching frequency and the need to get good waveforms at low
speed and maximum output (and therefore minimum switchings) at high speed
a method has to be found to suddenly change the number of pulses per half
cycle. This is known as 'gear changing' and in a typical system the number of
pulses per half cycle of output may follow a pattern like that shown in Table 5.1,
which also shows the changes in output and switching frequency which takes
place in each gear. From this you will see that the switching frequency is
maintained between specific limits (in this case 291-6 and 612) and that most of
the gear changes take place at the low output frequencies.
If such a system is going to be beneficial over the simpler unsynchronised
system it is necessary that the sudden transition from one gear to the next should
not produce any change in circuit voltage or current. This means that the
fundamental component of the output voltage waveform should not change on
transition. To do this the width and may be the distribution of the pulses in the
half cycle may need to be instantaneously changed on transition. Such arrangements can only be produced from large scale integrated circuit electronic chips
or from memory based microprocessor systems.
When the final stages of pulse dropping occur from nine pulses per half cycle
down to one some disturbance is inevitable and it can only be reduced by
selecting specific switching points in the output waveform before and after
174
The pulse width modulated voltage source inverter system
transition. Again, memory based microprocessor systems are really the only way
of achieving the necessary performance.
If further details of these systems are required then reference should be made
to the papers referred to in the Bibliography, which will prove most useful.
Table 5.1 Typical 'gear ratios'
Output frequency
Max Hz
MinHz
0-6
1-2
1-9
3-4
5-6
10
17
34
1-2
1-9
3-4
5-6
100
17
34
60
No. of pulses
per half cycle
243
135
81
45
27
15
9
5
Switching frequency
MinHz
291-6
Max Hz
583-2
324
513
307-8
306
302-4
300
306
340
550-8
504
540
510
612
600
5.2.2 The PWM drive system
The elements of PWM drive systems are generally similar to those of the six step
system with the exception that the mains converter can be a diode rectifier only,
and no control is required from the input side of the DC link. PWM systems are,
in general, voltage source DC link systems as shown in Fig. 5.8.
A constant DC link voltage is used and all the control is done via the motor
inverter operating in the pulse width modulated mode.
The inverter uses transistors, GTO thyristors or forced commutated thyristor
switches which have to be able to switch at the PWM carrier frequency which
will be many times the normal output frequency to produce the simulated sine
wave voltage to the motor. As it is a voltage fed system, reverse diodes are
required with each inverter switch to provide a path for reactive currents to flow.
The circuit of Fig. 5.8 includes a DC link reactor as a means of reducing the
level of high frequency currents getting into the input circuit and to force these
currents to flow in the DC link capacitor. The reactor is not needed to smooth
the DC link voltage because the diode rectifier already produces a good and
steady DC level and some manufacturers dispense with this reactor for economy
reasons.
The DC link capacitor is essential to provide a path for the currents which
flow through the feedback diodes in the inverter. As the inverter is in general
operating at high frequency, large AC ripple currentsflowin this capacitor and
it has to be correctly selected for these conditions.
The switching of the inverter circuit is usually arranged so that all switches are
being switched continuously at the high frequency switching or carrier frequency
with each AC output connection to the motor being alternately switched from
The pulse width modulated voltage source inverter system
175
the positive to the negative of the DC link at the carrier frequency. The average
value during this high frequency switching cycle will then represent the value of
the output voltage. Hence the zero point of the output voltage will be produced
when the ON time of the positive switch is equal to the ON time of the negative
switch.
reactor
Fig. 5.8 The PWM voltage source inverter drive
current
switch 1 on
off on
off on off on
off an off\on
off on
off on
SI
SI
"51
P
;V
IsU
switch 4 , off I on | off on | off |on | off |on off
current
carried
by
I
I I
ISA|DI
I I I
I I I
Fig. 5.9 PWM current flow
ISAIDI ISAIDI I'M SI
II
I I
II
II
I/It I
|D«|
i
on. off on off .on,
i i
DA SI
'I
II
DA. SI
I '
II
ii
DA
II
I i
The high speed alternate closing of the two phase switches means that the
current is being continually switched from the main inverter switch into the
complementary diode, as in Fig. 5.9, which shows the conditions as the output
176
The pulse width modulated voltage source inverter system
current crosses from negative to positive. The positive half cycle of current
is shared between the positive switch and the negative diode and vice
versa.
The high frequency switchings are modulated appropriately to produce the
shape of the output waveform and this modulation allows the output voltage
and frequency to be controlled. As indicated above, the modulation just alters
the ratio between the ON times of the complementing switches e.g. (1 and 4).
The three phases are modulated similarly but at 120 electrical degrees (at the
output frequency) to each other and it is clearly practical to alter the phase
sequence electronically so that reverse rotation of the motor can be achieved.
Control over the drive, in all respects, is now carried out via the inverter alone
and most PWM pattern generating systems include inputs to enable independent
setting of voltage, frequency and phase sequence so that the correct conditions
for the motor can be produced.
As with other voltage source systems, if the frequency to the motor is reduced
suddenly the motor can regenerate the load energy into the inverter and the DC
link rises in voltage due to the energy being fed into the capacitor via the
feedback diodes. To guard against this possible increase in DC voltage which
could quickly damage the semiconductors it is usual to include a DC voltage
measurement which will cause increase in inverter frequency if a high DC
voltage is detected. This prevents the motor slowing down too quickly. If fast
slow-down is required then some means of absorbing the regenerated energy on
the DC link is required. Most control systems involve measurements of circuit
currents and it is useful to note here that:
Measurements of current on the input, or on the DC link will indicate the
drive power level because of the diode rectifier and the constant DC link voltage.
Measurements taken at the inverter output will give the motor current and as
its power factor will depend on operating conditions and it is not a clear
indication of motor torque.
Pulse width modulated inverter systems of this type in general, provide
superior performance to the six step alternatives:
1) The range of speed control is much wider and operation at and
around zero speed is quite satisfactory.
2) Low frequency torque pulsations do not occur in the output and
hence there is less chance of exciting mechanical load resonances.
3) The current waveforms in the motor are always very near to sinusoidal
leading to less motor derating.
4) The diode input rectifier means that the input power factor is always
high whatever the speed and load.
5) In multidrive systems it is possible to connect a number of inverters
to the same DC link to allow transfer of regenerated power from some
drives to help feed other motoring drives.
However these advantages are partially balanced by the increased complexity
and by the increased difficulty in protecting these systems.
The pulse width modulated voltage source inverter system
177
Fig. 5.10 This 150 HP drive uses gate turn off thyristors for the inverter switches which are
shown, complete with gate drive circuits, in the top of the cubicle. The diode rectifier
and DC link capacitors are behind the lower panel. The PWM generation and control
electronics are micro-processor based and are on the left hand card. The right hand
card is the switched mode power supply for the drive electronics. (G.E.C. Industrial
Controls, Ltd.)
178
The pulse width modulated voltage source inverter system
5.3 Detailed analysis of the system
In this section I will deal with the waveforms which exist throughout the circuit
and the relationships which occur between the electrical parameters of the
circuit.
In this system the dominant features of circuit operation are the fact that it
is a voltage source system with a constant DC link voltage and the relatively
complex high frequency switching patterns used in the inverter. The large DC
link capacitor means that whatever current the motor requires in response to the
applied voltage and frequency will be able to flow.
As with all AC motor drives the main aim is to apply the appropriate
frequency to the motor to enable it to rotate at the speed desired and to ensure
that the voltage applied is correct to give the correct magnetising conditions in
the motor and the required torque. In this system all the variability rests in the
inverter with both voltage magnitude and frequency being dictated by the high
frequency pulse generation arrangements.
We should therefore start by studying the pulse width modulated voltage
waveforms as applied to the motor in more detail.
5.3.1 Motor waveforms
Voltage waveforms
During normal PWM operation all three phases of the inverter are being
continually pulsed at the high switching frequency similarly to the way shown
in Fig. 5.9 so that the appropriate connection to the motor is being continually
switched from the positive to the negative of the DC link. The ratio of the times
of connection to the positive and negative rails decides the instantaneous
average level of the phase voltage to the motor and this ratio is modulated in
a normally sinusoidal way to obtain the lower frequency fundamental phase
voltage waveform.
The three phases all operate in a similar way and they will all be operating at
the same switching frequency but their modulation waveforms will be displaced
by 120 electrical degrees based on the fundamental output frequency. The phase
and line voltage waveforms therefore become quite complex due to the changing
frequency of the modulation waveform and, when a gear change PWM system
is being used, the wide range of the high switching frequency. However, to help
in the understanding of the principles Fig. 5.11 has been drawn based on the
unsynchronised triangular wave PWM generation method. This shows a
common triangular wave being used by all three phases and onto this the three
modulation waveforms are superimposed. The points where the modulation
waveform crosses the triangular wave decides the points of switching of the
appropriate inverter switches and this is shown for the three phases immediately
below the triangular waveform. The black lines indicate when the positive side
switch (switches 1, 3 or 5 in Fig. 5.8) is switched ON and the spaces show when
The pulse width modulated voltage source inverter system
179
the negative side switch is ON. Fig. 5.12 shows these switching patterns for the
three phases more clearly — now choosing nine pulses per half cycle for clarity.
Fig. 5.13 shows one of the line voltages which result from them, these being the
difference between the waveshapes in Fig. 5.12. The line voltages now clearly
show three levels in the voltage waveform, the DC voltage in a positive direction,
triangular carrier wave
phase A
phase B
phase C
firing
points
Fig. 5.11 3 phase operation
fundamental sine wave
s
y
<
//
sS
y
s
N
S
s
s
s
/ ffl
/
1
Fig. 5.12 3 phase PWM voltages
s
s
s
180
The pulse width modulated voltage source inverter system
the DC voltage in a negative direction and a zero level. Also the line voltages
show twice as many switchings per half cycle.
The figures show the voltage waveform for one specific condition only, one
specific output frequency (in relation to the carrier frequency) and one specific
voltage level. Clearly there are numerous such conditions and the waveforms
produced will be different in all cases. Also thesefiguresshow conditions where
the modulation waveform is synchronous with the triangular carrier waveform
where the result is that the motor waveforms are identical in all cycles. If the
waveforms had not been synchronised then succeeding cycles would have
different pulse patterns to the preceeding ones making them even more complex
to appreciate in detail.
A phase voltage w.r.t centre of DC link
B phase voltage w.r.t centre of DC link
»
4*
— -
AtoB
line voltage
|
V dc
-fundamental
value
Fig. 5.13 PWM motor line voltage
Further complexity occurs due to the need for reducing the switching
frequency by the gear changing and pulse dropping techniques briefly described
in Section 5.2.1 and hence to fully appreciate any specific PWM type drive it is
necessary to study a very wide range of conditions and the specific results will
depend on the particular PWM techniques being employed.
However the principles demonstrated in Fig. 5.9 are true for the majority of
the systems presently in use and these are summarised as:
1) Continuous switching of all phases of the inverter bridge circuit.
The pulse width modulated voltage source inverter system
181
2) Complementary switching of the two switches in each phase with the
switches being ON and OFF alternately without any significant
period with both switches being OFF or open.
3) The phase voltage shows switching from the positive to negative rail
alternately.
4) The line voltage shows switchings from zero to positive during the
positive half cycle and zero to negative during the negative half cycle
and pulses at twice the switching frequency.
If we now relate these principles to their use in variable frequency motor
drives:
At low speed, low output frequency there will always be a large number of
switching pulses in the line voltage waveform. As the voltage magnitude will
also be low the pulses will be relatively narrow even those occurring at the peak
of the fundamental voltage sine wave.
As the frequency output is increased the number of pulses occurring per
output cycle will reduce but their width will increase in order that the fundamental voltage magnitude can be increased.
average of pulses
Fig. 5.14 Line voltage at high output frequencies
At high output frequencies the number of pulses per cycle will reduce still
further and the width of the pulses will increase. The majority of systems will
however allow the pulsing patterns to change at high frequencies with the
central wide pulses all joining together to form a block pulse. So in general the
line voltage waveform to the motor will consist of a few high frequency pulses
either side of a block pulse in each half cycle (see Fig. 5.14). The width of the
block pulse will alter as the voltage magnitude changes. At the maximum
frequency some systems give a final quasi-square single block pulse waveform
as used in the system described in Chapter 4.
182
The pulse width modulated voltage source inverter system
Motor current waveforms
When the switching frequency is high compared to the output fundamental
frequency the motor currents tend to be closer to sinusoidal in shape due to the
smoothing effect of the motor inductance.
As indicated in Chapter 1 the induced voltage in the induction motor will
always be very near to sinusoidal. Therefore the harmonics in the terminal
voltage are all lost across the stator leakage reactance and the value of this at
the switching frequency decides the amount of high frequency contained in the
current waveform. With the improved semiconductors which are continually
being introduced the switching frequencies possible are increasing all the time,
so that the current harmonics in these systems are becoming less and less
significant.
From the motor point of view therefore, its operation can be considered to
be that given by sinusoidal conditions, and the relatively minor effects produced
by the high frequency switching can be ignored.
If the required frequencies demand the use of pulse dropping leading to
quasi-square wave operation at the high speeds then clearly other waveform
conditions will occur. These usually involve a higher degree of harmonic content
in the waveform compared to the lower speed conditions. However as will be
seen from the study of the six step system in Chapter 4 quasi-square operation
at the higher speeds is quite acceptable and typical motor inductances lead to
reasonable current waveforms and only a limited harmonic content.
5.3.2 Inverter circuit waveforms
As will be seen from Fig. 5.9 the inverter conditions are really dictated by the
high frequency operation with the circuit current being switched from the
thyristors to the feedback diodes at the high frequency rate. The lower frequency
output has a rather second order effect on the inverter's operation by just
altering the widths of the high frequency pulses. The operation of the inverter
is therefore akin to a high frequency inverter working with a slowly changing
output current.
Although the detailed waveforms are clearly dependent on the particular
method of PWM generation which is being used the principles are similar for
all methods and hence Fig. 5.15 will be used to illustrate these principles. The
condition represented here is for an inverter giving an output voltage of approximately half of the maximum PWM value, with a sinusoidal motor current (in
a delta connected motor) at a power factor of approximately 0-80 per unit.
The top of thisfigureshows the triangular wave and the three modulation sine
waves with peak values of half the peak of the triangular wave. The intersections
of the two waveforms enable the voltage waveforms of each output terminal to
be decided and these show the switching points from the positive to the negative
side of the DC line. Onto these voltage waveforms are superimposed the
sinusoidal currentsflowingin the output terminals of the inverter. The switching
points from the voltage waves decide how the currents are chopped up so that
The pulse width modulated voltage source inverter system
183
phase A voltage and current
Si
phase B voltage and current
phase C voltage and current
D2
D2
D2
D2
DC link current - inverter input
LJilllJhl I JliIIIIJ••!!JL
voltage across SI inverter switch
n n n n n n n nnnnnn
Fig. 5.15 PWM inverter circuit waveforms
some of the current flows in the diodes and some in the transistor/thyristor or
GTO switches. The references to the switches and diodes refer back to those on
Fig. 5.8.
From these waveforms it can be seen that the currents in the switches and
diodes consist of a series of high frequency pulses with heights which follow the
output sine waves and with pulse widths which vary due to the modulation
184
The pulse width modulated voltage source inverter system
needed to produce the output voltages. If the output voltage required is very low
then the width of the current pulses are equal to the spaces between them and
when the output voltage is at a maximum value (equal to the triangular wave)
then current is flowing in the appropriate switch for most of the time.
The current flow in the DC link is the sum of the switch and diode currents
connected to that link cable and the diode current is always in the reverse
direction to the switch currents. The DC link current figure is obtained by
adding together the currentsflowingin switches 1, 3 and 5 and subtracting from
this the diode currents flowing in diodes 1, 3 and 5. The result is a train of
current pulses of almost identical heights and with an approximately constant
ON to OFF ratio. This ON to OFF ratio is found to be approximately equal
to the size of the modulation waveform in relation to the size of the triangular
wave, and the ratio is therefore approximately proportional to output voltage.
Whatever the level of motor current being carried, at low voltage the DC link
current is a series of very narrow current pulses because the switch and diode
currents cancel each other out for most of the time. At high voltage, the diode
currents flow for only very short periods and most of the current is carried by
the switches.
The current flowing in the DC cable on the motor side of the capacitor is
therefore a high frequency pulsating current and it is the capacitor's job to allow
this current to flow as required. If we assume that there is a significant DC link
reactor on the supply side of this capacitor then all of the pulsations in current
willflowin the capacitor and only the average level of the pulses will be flowing
in the reactor and the supply rectifier.
The graphs in Fig. 5.16 have been drawn to show the magnitudes of the ripple
currentsflowingin the DC link and the capacitor. This shows that, as the motor
voltage varies, the mean and RMS values of the currents in the DC supply to
the inverter vary. If it is assumed that the same value of motor current can flow
at any motor speed and voltage then the mean DC current will vary linearly with
output voltage and the capacitor's ripple current will have a maximum RMS
value of 50 per cent of the peak inverter output current occurring at approximately half volts and speed. From Fig. 5.15 it can be seen that the capacitor
currents will be at a basic frequency of twice the inverter switching frequency.
Fig. 5.15 also shows the voltage which occurs across one of the switches, in
this case, switch 1. The presence of the reverse diode means that the voltage
never reverses and only positive anode to cathode voltage occurs. The voltage
oscillates from zero to the DC link voltage at the high switching frequency.
The very rapid current and voltage changes occurring on the switches and
diodes is a dominant factor in the specification of the inverter. Because switching
cannot take place instantaneously high switching losses can be generated in the
semiconductors and this will usually be the deciding factor in the frequency at
which the inverter can be operated. The high rates of change of current are also
usually unacceptable to the semiconductors and most inverters include small reactors in appropriate places to limit the rate of change of current which can occur.
The pulse width modulated voltage source inverter system
185
approx. max.
RMS ripple
current in
capacitor
05
modulation depth
peak of ref wave
10
peak of sawtooth wave
Fig. 5.16 DC link current conditions
5.3.3 Circuit relationship and equations
As mentioned previously this drive operates with a constant DC link voltage,
with the frequency and voltage control being carried out in the PWM generation
system used to control the inverter. The system is shown in Fig. 5.8 and this
should be referred to during the following explanations.
The DC voltage is always the rectified value of the mains supply voltage and
normally this value will be given by
Vdc = 1-35 x Vs
(1)
where Vs is the RMS line voltage of the supply. Clearly if there are any reactors
in the supply connections or if there is a supply transformer with a finite
impedance then the voltage will be slightly less than this figure when the drive
is operating under load. Due to the presence of the capacitor the DC voltage will
rise a little above this value at low load levels — it will take up the peak values
of the mains supply sine waves which will therefore result in
Vc = y/2 x Vs = 1 414 x Vs
The maximum voltage which can be applied to the motor will depend on the
type of switches used and the features of the PWM generation system. If the
method used allows the drive to eventually operate in six step quasi-square wave
mode then the maximum voltage will be the same as that achieved in the six step
186
The pulse width modulated voltage source inverter system
drive system referred to in Chapter 4, i.e.
Maximum motor voltage = 0-8165 x Vdc volts RMS.
(2)
and this will have a fundamental value which dictates motor performance, of
Max. Vm(fundamental) = 0-778 x Vdc
(3)
If however, PWM operation is to be retained even at the maximum voltage
condition then the value of the voltage obtainable will depend on the minimum
ON and OFF times of the switches and the frequency of switching in relation
to the operating frequency. When using thyristors at the 500 to 800 hertz
switching frequency the motor voltage at 50 or 60 hertz output frequency can
usually only reach approximately 75 per cent of the above quasi-square value.
When transistors are being used the maximum voltage will be higher than this
due to the reduced minimum ON and OFF times and due to the higher
operating frequency.
It is now necessary to study the motor to decide on the currents flowing in the
motor and in the inverter. Using the simplified motor equivalent circuit as
described in Chapter 1, the motor current will be made up of two components,
the magnetising current and the in-phase current which produces torque.
The magnetising component as before will be equal to
Imag = -.75xlsatxlog e [l-^^]
(4)
using the variables as designated on Fig. 1.13 and with F being the actual
frequency and Fr the rated frequency.
The torque component will be approximately given by the equation
It = Vm/(R1 + R27S1)
(5)
with all values in this equation being phase values.
The total phase current in the motor will then be given by the equation
Im = ^(Imag) 2 + (It)2 amp fundamental
(6)
The motor power factor equals approximately
COS $ = It/Im
(7)
and the power to the motor is equal to
Pm = 3 x It x Vm
(8)
The power out of the motor will be equal to the power in multiplied by the
efficiency and the motor torque will be given from this and the speed of the
motor,
Speed = S = 120 x F/P x (1 - SI) RPM
(9)
and
Torque = (Pm x Efm x 60)/(2 x n x S) Newton metres (10)
The pulse width modulated voltage source inverter system
187
Where F = Actual frequency, P = No. of poles, SI = Slip and Efm = Motor
efficiency.
Having found the motor current we can now see how this is shared out in the
inverter. From Fig. 5.15 we can see that switching of the inverter causes part of
the current to flow through the switches and part through the diodes, the half
sine waves of current being chopped up into pulses which pass alternately
through diodes and switches.
The peak value of these currents will always be the peak value of the line
current from the inverter to the motor.
When a low output voltage is being produced the modulation depth will be
very small and the current pulses passing through the switches will be approximately half the total half sine wave output current, the other halfflowingin the
diodes. As the output voltage increases then the modulation depth increases and
the amount of the half sine wave which flows in the switch increases.
motor power factor
05
10
switch
RMS
currents
0-5
modulation depth
peak of ref sine wave
=
peak of triangular wave
Fig. 5.17 Switch and diode PWM current ratings
The split up of the current between diode and switch is also affected by the
power factor of the current. As the power factor reduces so a larger proportion
of the current flows in the diodes and less in the switches.
The graphs of Fig. 5.17 show the way in which the inverter switch and diode
188
The pulse width modulated voltage source inverter system
currents vary in a typical PWM drive when the speed and voltage change under
various modulation depths and output power factors.
The DC link currents on the inverter side of the DC capacitor will be
pulsating at twice the inverter switching frequency as explained earlier and the
curves of Fig. 5.16 can be used to estimate the RMS and mean values.
On the supply side of the capacitor however the more accurate way of
assessing the current is from the power in the system. The power being passed
across the DC link is equal to the DC voltage multiplied by the mean DC link
current and this will be directly related to the motor input power by efficiency
of the inverter. Therefore from equation (8)
^ ^ ,. ,
Motor input power
DC link power = —
_
Inverter efficiency
Vdc x Idc = (3 x It x Vm)/Inverter efficiency
Therefore the mean DC link current is given by
Idc = (3 x It x Vm)/(Vdc x Inverter efficiency)
(11)
Therefore the DC link current is in fact proportional to the power being passed
through the drive because the DC link voltage is normally constant.
If the DC link contains a reasonable size DC reactor so that the high
frequency current pulses are contained in the capacitor and inverter then the
current flowing from the supply convertor will be relatively smooth and the
supply side AC current will be quasi-square wave shape.
Hence the RMS supply current will be given by
Is = V2/V3 x Idc
(12)
If there is no DC link reactor or any other reactors in the supply rectifier some
of the high frequency current pulses may flow in the AC mains connections thus
increasing the RMS current flowing into the drive system.
5.3.4 Examples of calculations
1) Inverter switching frequencies
Question 1
A high frequency pulse width modulated transistor inverter for a spinning
machine drive has to produce an output frequency of 170 hertz to drive the
3 phase motor at 10,000 RPM. If a non synchronised constant frequency PWM
system is to be employed, what switching frequency must be employed to ensure
no less than 9 pulses per half cycle occur in the output line voltage and how
many pulses would there be per half cycle when the output frequency was
34 hertz.
Answers
With a constant switching frequency the minimum number of pulses per half
cycle will occur at the maximum output frequency i.e. at 170 hertz.
The pulse width modulated voltage source inverter system
189
Reference to Fig. 5.13 shows that the line voltage output of a 3 phase PWM
inverter will have twice as many pulses per half cycle as in the phase voltage
and the pulsations in the phase voltage correspond to the inverter switching
frequency.
Therefore the frequency of the line voltage pulsations equals
170 x 9 x 2 = 3060
and the inverter switching frequency will be
3060
1C.A.
—— = 1530
hertz.
The number of pulsations per half cycle of the output line voltage at 34 hertz
output frequency will be equal to
1530
34
= 45
Question 2
A synchronised gear-changing PWM system is to be used with a 3 phase inverter
capable of operating at a switching frequency of up to 500 hertz to achieve a
maximum number of pulses per half cycle of line voltage of 21 at the minimum
speed, and 7 at the top speed. What are the minimum and maximum output
frequencies if the switching frequency is going to be contained within the band
300 hertz to 500 hertz and what is the minimum number of gear changes between
these speeds if the number of pulses per half cycle must be an odd number.
Answers
Maximum output frequency will occur at 7 pulses per half cycle and a 500 hertz
switching frequency.
A 500 hertz switching frequency will give 1,000 pulses per second in the line
voltage and we require 14 pulses per cycle of output frequency.
Therefore the maximum output frequency equals
1000/14 = 71-4 hertz.
The minimum frequency will occur at 21 pulses per half cycle and a 300 hertz
switching frequency.
On the same basis the minimum output frequency will be
600/42 = 14-3 hertz.
With 7 pulses per half cycle the top gear can be used down to a frequency of:
71-4 x 300/500 = 42-84.
At around 43 to 44 hertz it is necessary to change gear to jump up to 500 hertz
switching frequency and 11 pulses per half cycle in the most appropriate choice.
190
The pulse width modulated voltage source inverter system
Now with the commutation frequency being reduced to 300 hertz the output
frequency can go down to:
300/11 = 27-3 hertz.
At this point it is necessary to change gear again and this time a jump up to
17 pulses per half cycle is appropriate to keep within the 500 hertz maximum
switching frequency.
Under this condition a reduction of 300 hertz corresponds to a lower
frequency of:
300/17 = 17-6 hertz.
so that one more gear change to 21 pulses per half cycle is needed.
Therefore the minimum number of Gear Changes for this range is three.
Fig. 5.18 shows this in graphical form.
500 r
20
30
40
50
output frequency - Hz
60
70
80
Fig. 5.18 Typical gear changes
2) Drive calculations
Question
A PWM type inverter drive for a 3 phase, 4 pole induction motor rated for
25 HP, 460 volts, 60 hertz supplies an output current of 20 amps RMS line
current at 0-80 power factor at 230 volts, 30 hertz. If the inverter is 85 per cent
efficient under this condition find the DC link mean current when the inverter
is supplied from a 500 volt, 3 phase supply.
The pulse width modulated voltage source inverter system
191
Answer
The power supplied to the motor at this operating condition is given by
Motor power input = Vm x Im x 3 x Power factor,
where Vm and Im are phase values. Therefore
Pm = 230 x 20 x 7 3 x -8
= 6374 watts.
Therefore the power into the inverter is equal to this value divided by the
inverter efficiency
i.e.
DC power = 6374/0-85 = 7499 watts.
The DC voltage will be equal to the rectified value of the mains supply from
equation (1)
i.e.
Vdc =
1-35 x Vs
=
1-35 x 500
= 675 volts.
Therefore the DC link current is given by
Idc = 7499/675 =
111 amps mean.
Question
If under this condition, the motor efficiency is 82 per cent and its slip speed is
12 RPM, what torque is the motor providing to the load.
Answer
The motor output power equals its input power multiplied by its efficiency.
Power out = 6374 x -82
= 5227 watts.
The synchronous speed with a 30 hertz supply to the motor will be given by
equation (9)
S =
120 x 30/4 = 900 RPM.
Therefore the motor speed equals
900 - Slip speed = 888 RPM.
From equation (10)
Torque = (5227 x 60)/(2 x n x S)
= 5612 Newton metres.
192
The pulse width modulated voltage source inverter system
5.4 Practical circuit design considerations
This pulse width modulated inverter system is a voltage source circuit with a
relatively large DC link capacitor and as such it behaves in many ways similar
to the six step system described in Chapter 4.
The supply side rectifier has to be protected against mains borne transient
overvoltages but due to the presence of the DC link capacitor these transients
do not usually reach the inverter. Some overvoltage suppressors or resistor
capacitor circuits may be fitted but they may be relatively minor because it is
easy to obtain high voltage diodes relatively cheaply.
The inverter does not require high voltage safety margins on the semiconductors or voltage suppressors because of external transients but it does have to
be protected against those which it generates within it due to the fast switching
employed in it. The very high rates of change of current which are a feature of
PWM inverters can cause very large voltages to occur even in stray inductances
and hence detailed design and construction of the inverter has to be very
carefully considered. Compact low inductance designs are usually used and
semiconductors often have complex snubber circuits and di/dt limiting reactors
to protect them. In addition all the components of the inverter have to be
specially selected to cope with the high frequency currents i.e. the snubber
circuits, the interconnecting cables and the DC link capacitor.
The motor current waveforms are relatively sinusoidal with only a very small
harmonic content so that the conductor losses etc. in the motor are very similar
to those on normal sinusoidal operation. The voltage, however, does usually
contain a substantial content of the inverter frequency due to the pulsating
nature of the waveform and this will produce additional losses in the iron. In
addition the motor is exposed to very fast voltage pulses and this may need to
be taken into account when the motor is selected.
In some of the PWM designs in use there is no DC link reactor included and
this may lead to a portion of the high frequency currentsflowingin the supply
rectifier and into the mains supply connections. This can result in electromagnetic
and radio interference which can affect other systems in the locality or connected
to the same mains network.
Those designs which include a DC link reactor or AC line inductance are in
general better from this point of view as the high frequency currents are then
contained within the inverter assembly and radiated interference is minimised by
the enclosure.
5A.I Overcurrent protection
The fault conditions in this PWM circuit are in general very similar to the six
step circuit (Chapter 4) in that the large DC link capacitor can be a source of
high circuit currents if maloperation of the inverter switching occurs. One
difference is that the inverter switches are inherently capable of being turned off
much quicker than may be the case in the six step circuit. The other important
The pulse width modulated voltage source inverter system
193
difference is that, in its basic form, the supply side converter is unable to be
switched off at all as a back up against inverter faults and reliance has to be
placed on the supply circuit breaker, contactor or fuses.
Hence in this circuit much more reliance has to be placed on the inverter
switches themselves as the means of cutting off fault currents before further
damage is caused. It is essential that fault currents and conditions are detected
very quickly and that the inverter switches are able to turn off the fault current
before it reaches a value above which the switches themselves are able to cope.
As with other voltage source circuits the worst case overcurrent fault condition is that associated with incorrect operation of one or more of the switches.
If the two switches on one phase of the inverter are ever allowed to conduct
together they will short the DC link causing the DC link capacitor and the
supply converter to feed into the short circuit. At the same time the current in
the motor will immediately start to reduce due to the sudden loss of voltage
produced by the short circuit.
The current in the fault circuit will rise at a rapid rate due to the capacitor
energy and to the low inductance of the inverter circuitry and the principle of
protection is to cause the switches in the fault path to be switched off as quickly
as possible after the fault is detected and before the fault has an opportunity to
rise above the maximum switching level of the switches.
In many transistor and GTO systems, the individual switch arms of the
inverter are themselves fitted with high speed overcurrent measurement and
protection so that their switch off can be initiated as quickly as possible. In
addition it is likely that on small equipments a fuse will be inserted in the DC
link after the capacitor to ensure damage is not serious if the switches themselves
are unable to cope.
5.4.2 Regeneration
The inverter in this system is capable of accepting energy from the motor as well
as providing power to drive it. If ever the frequency of the inverter is reduced
below that dictated by the motor speed then the motor will regenerate to slow
down and power will be fed into the DC link causing the capacitor charge to
increase and causing the DC link voltage to rise. This capacitor charge is caused
by a reverse flow of current in the DC link and this is clearly blocked by the
input convertor and hence the DC link voltage would rise rapidly in such
regenerative circumstances.
If it is required to slow the motor down quickly by absorbing this regenerated
power it is usual in PWM inverters to add switcher resistors on the DC link or
on the AC output lines. These would be switched into and out of the circuit via
a measure of the DC link voltage. Alternatively, a reverse connected thyristor
convertor could be connected to the DC link to allow the power to be fed to the
AC input mains supply. Clearly this method is only sensible if the quantity of
energy to be absorbed is substantial or if accurate control of slow down is
required.
194
The pulse width modulated voltage source inverter system
Most standard inverters are not fitted with absorption facilities and they
usually include control methods of avoiding the regeneration of energy back
into the DC link in order to prevent the resulting rise in voltage which could
damage the components of the inverter. These methods can include the switching off of the inverter if a high DC link voltage is detected, or a feed-back into
the control which automatically keeps the inverter frequency up if the DC link
voltage rises.
One specific advantage of the PWM drive system from the regeneration point
of view is that as the DC link voltage is constant a number of inverters and
motors could be connected to the same DC link and in this case there could be
an interchange of energy between the motors, e.g. if one is being slowed down
the energy can often be absorbed by one of the other drives.
5.4.3 Factors affecting the specifications of the main components of the circuit
There is no doubt that the dominant feature of this circuit is the high frequency
operation of the inverter and the high frequency pulsed nature of the inverter
switch currents, the feedback diode currents and the current in the DC link
capacitor.
The DC link capacitor has to be of such a size that the pulsed currents can be
drawn from the capacitor without much variation in the DC link voltage. As
shown in Fig. 5.16 the capacitor has to be able to cope with a high frequency
current with an RMS value equal to approx. 70 per cent of the RMS value of
the output line current from the inverter.
The inverter is exposed to the full DC voltage level at all times and all the
components in the inverter have to withstand continuous high frequency switchings at this DC voltage level whatever the effective output frequency. In fact as
can be seen from the foregoing the output frequency is relatively secondary to
the operation of the inverter; it is basically a high frequency inverter with
currents which vary at the low output frequency rates.
The inverter switches can be forced commutated thyristors, transistors or gate
turn off thyristors but with new designs the preference now is for the two latter
switches as they can operate at higher frequencies.
5.4.4 Typical practical circuit diagram
Fig. 5.19 shows a typical circuit for a PWM voltage source drive. This includes
one particular item which has not yet been mentioned, a switch/resistor slow
charge circuit for the main DC link capacitor.
Where a diode input rectifier is used there will normally be a sudden rise of
the DC voltage when the mains supply is switched on. This will cause a very
large inrush current into the DC link capacitor and this may damage the input
circuits and components. To avoid this it is usual to include a resistor in the DC
link prior to the capacitor to restrict the inrush current. This resistor is then
shorted out with a contacter or a thyristor switch once the capacitor is charged.
diode
rectifier
Fig. 5.19 PWM drive power circuit
supply
switch
contactor
switched
charging
resistor
DC voltage
measurement
regeneration
energy discharge
circuit
motor
voltage
inverter switches
complete with voltage
and current protection
components „
to
3"
3
5?
I
196
The pulse width modulated voltage source inverter system
The DC link capacitor has to have safety discharge resistors due to the long
time for which it will otherwise retain its charge after switch off.
Regeneration is dealt with by the optional fitting of a switched resistor across
the DC link, the firing of the switch being initiated by the detection of a high
DC voltage. The switch has to be self commutated in order that it can be
switched off once the regeneration has stopped.
5.5 Overall control methods
In the majority of PWM drive systems the only two controls directly available
are both associated directly with the firing of the inverter switches.
The level of motor voltage is controlled by varying the widths of the high
frequency pulses.
The motor frequency is decided by the points in time when the effective
inverter voltage reverses — again dictated by the PWM pattern firing system.
In general there are usually arranged to be two independent inputs into the
PWM generation system so that one signal can be altered without directly
affecting the other.
As with most voltage source systems the motor is basically left to itself to
respond to these two parameters. To achieve the most satisfactory performance
it is necessary to ensure that the motor voltage and frequency are directly related
so that the flux in the air gap is reasonably constant and usually at the rated
motor value. Some degree of variation of the V/f ratio may be carried out to
achieve optimum flux and therefore torque conditions.
In all such systems the control methods employed to achieve the best motor
performance are usually based on measurements of the current in the system
only. This may be the motor current itself or the current in the DC link. The
motor current will include the magnetising current required by the motor as well
as the current needed to produce the output torque. The DC link current will,
in general, be a measure of the power being drawn by the load because the DC
link voltage is fairly constant in normal use.
Fig. 5.20 gives a typical example of the control system employed for a PWM
inverter system.
The heart of the system is the PWM generation system which may be LSI or
microprocessor based (see Section 5.2). In all cases such systems require input
signals of frequency, voltage and direction and these inputs are provided by the
remainder of the electronic control scheme.
In the system of Fig. 5.20 the overall principle of control is to decide the
frequency to be supplied to the motor and then to arrange that this dictates the
level of voltage which should be applied according to a predetermined relationship which will ensure constant flux in the motor particularly at low speeds.
The decision on frequency is the result of a complex arrangement taking
account of:
The pulse width modulated voltage source inverter system
197
1) The speed required by the operator - the speed reference value. (1)
in Fig. 5.20.
2) In (2) a signal proportional to load torque is added to the speed
reference to compensate for the slip of the motor.
3) The frequency is normally only allowed to change smoothly and an
electronic ramp is included to do this (3). The rate of acceleration and
deceleration are presettable by the user.
rectifier
inverter
slip
compensation
Fig. 5.20 Control system for PWM drive
There are usually also additional inputs into the frequency decision circuits
to cater for limiting conditions.
If the rate of deceleration allowed is too fast and regeneration occurs the DC
link voltage will rise. In this system this is taken account of by holding the
frequency ramp if too high a DC link voltage is detected. This is done by item
(6) in Fig. 5.20.
In some equipments a torque limit is also included to prevent the drive from
being overloaded. This is similar to a current limit in DC systems. If an excessive
torque is detected, in this case, via the current measurement, the frequency is
reduced until the torque is brought within the drive rating. This is done in
item (7).
From the above it will be seen that what is really required to be measured for
the most satisfactory performance of such schemes is the motor torque. Because
of the difficulties associated with measuring motor slip or shaft torque directly,
most inverter drives of this type include some means of calculating a reasonable
measure of torque from the electrical measurements made. In Fig. 5.20 the
current is measured in the motor connections and hence it includes a magnetising component. One way of obtaining the torque component from this is to find
198
The pulse width modulated voltage source inverter system
its in-phase component from an electrical comparison between the applied
voltages and the current and this is the aim of box (8) in the figure. In this case
signals indicating the zero cross-overs of the voltage waveforms produced by the
PWM system are used to assess the torque component of current and the output
of box (8) is this value which is subsequently used for slip compensation and
torque limiting or tripping.
If a current measurement based on DC link current was used it would be
necessary to change this from an indication of drive power to one of drive
torque. As power is speed times torque the usual way of arriving at an approximately correct value is to divide the DC current measurement by the frequency
signal. This is shown in box (10).
In cases where more accurate speed control is desired the drive can be fitted
with a tacho-generator to give a direct measurement of speed. In such cases slip
compensation is not required and torque can be calculated from comparing
speed with frequency to obtain slip speed. However current measurement is
usually also included for protection purposes.
It should be noted here that the measurements of current for protection
purposes are better made either in the inverter arms directly or in the connections from the inverter to the motor, because at low speeds the mean value of
the DC link current will be quite small even at full torque.
5.6 Performance and application
In general this drive system can provide very high quality performance over a
very wide speed range. With the larger number of voltage pulses per half cycle,
particularly at low speed, the current waveforms can be very near to a true sine
wave and very smooth performance at or near zero speed is obtainable. Many
present day systems also have the facility for the voltage waveform to be
changed from PWM to quasi-square wave in a properly organised way (so that
current surges do not occur) and as a result very high motor frequencies can be
produced to achieve the highest motor speeds required.
These improved performance capabilities are however achieved by employing
very high quality and highly specified semiconductor switches (and maybe
diodes) and by using relatively complicated electronic systems particularly the
PWM pattern generator itself. There is also the very complex and variable
voltage waveforms produced by the inverter which may make the operation of
the system difficult to understand. If a non-synchronised PWM generator is used
then every half cycle of voltage is likely to have a different pulsed waveform than
all the others, a fact which does not help total understanding.
Because of the high specification of the inverter switches the operating voltage
of these systems has up to now been limited to the range up to 500 volts AC,
but with the increasing use of gate turn off thyristors for PWM systems operating voltage capabilities are increasing.
The pulse width modulated voltage source inverter system
199
Being a voltage source system for induction motors this drive is not much
affected by the precise parameters of the motor which is connected to it and it
is possible to supply a number of motors from the same drive as long as they
are all required to operate at the same frequency. In such cases load sharing is
not seen to be a problem due to the inherent slip of the induction motor and the
ability of the inverter to provide the currents which the individual motors may
demand.
Most PWM pattern generators allow for the reversal of the output voltage
waveforms so that electronic reversal of the motor can be used if needed. This
is achieved simply by reversing the direction of modulation of the inverter
switching at the most satisfactory point in the cycle.
The inverters used in this system are usually fully capable of accepting power
from the motor and feeding it back into the DC link but this facility may not
be used and it may even be prevented to avoid overvoltages on the DC link. If
no special arrangements are made to absorb or feedback regenerated power
than the energy will be dumped into the DC link capacitor causing its voltage
to rise quickly. When regular motor braking is required with a PWM drive
system, then either switched resistor will be included to dissipate the energy or
an additional feedback thyristor converter will be included to pass the power
back to the AC mains network.
5.6.1 Torque I speed characteristics
The capabilities of this drive in this respect are dominated by the high frequency
inverter switching and the capabilities of the inverter switches. As explained in
Chapter 2 all of these switches are limited in the amount of current that they can
switch and this directly decides the performance of this PWM drive system. As
the switches always have to be able to cope with the peak value of the motor
current then this value has to be limited to prevent inverter maloperation.
However, the inverter switches will usually be able to operate at this peak
current level at any operating output frequency and therefore motor speed, this
is even true with the motor at standstill.
The overall result is that the drive has a limited maximum current rating over
the full speed range and in general this means that the maximum motor torque
is dictated directly by this current value. With the correct control over motor
flux the peak torque achievable is independent of speed. As mentioned previously
it is the motor losses and cooling which will decide the level of torque which can
be sustained for significant periods of time.
5.6.2 Efficiency
This system is relatively good as far as motor losses are concerned, the motor
current is much nearer to sinusoidal than most of the other DC link systems
described in this book and hence the conductor losses are very near to those
occurring under sinusoidal conditions. The voltage waveform applied to the
motor does contain a substantial harmonic content and this does increase the
200
The pulse width modulated voltage source inverter system
iron and stray losses in the motor by an amount which will depend on the
frequency of inverter switching. Clearly if the drive is one which eventually
moves into quasi-square wave operation at high speeds then the motor losses
will then be similar to those in the six step drive described in Chapter 4.
The drive losses are dominated by the inverter where the high frequency of
operation results directly in an increase in power loss. Reference back to
Chapter 2 indicates that losses are produced in the switches every time they
switch the current ON and when inverters are operated at higher frequencies, as
is the case with this drive system, then the amount of the switching loss increases
so that it may be the dominating component of the total losses. This is in fact
another one of the reasons why PWM systems are arranged to eventually
operate in the six step mode — to reduce the inverter switching losses at high
speed.
5.6.3 Supply power factor
One of the important and significant benefits of PWM drive systems is the direct
result of having a supply side diode rectifier to give a constant DC link voltage.
The result is that the power factor of the input current to the drive is always high
and it does not vary with the speed of the drive. Drives of this type will have an
input power factor of around 0.95 per unit.
In addition the magnitude of the input current is related to the power being
drawn by the drive, motor and load, rather than to the torque as is the case in
most of the other systems described. Therefore at low speeds the input current
is low enough though the motor may be generating high torques. At reduced
speeds the supply current is invariably less than the motor current.
5.6.4 Motor and supply harmonics
The motor harmonics depend on the type of PWM generation system employed
but the following points are relevant:
a) Any significant harmonics in the motor current are related to the
inverter switching frequency, if the inverter frequency if high then the
actual value of the harmonics will be reduced due to the influence of
the motor inductance. When operating on a PWM basis the harmonics
in motor current can usually be neglected.
b) If the PWM system allows quasi-square wave operation at high
speeds then the motor harmonics under this condition will be similar
to those which would result from a six step system.
c) The motor voltage always contains a substantial content of harmonics
which at low speeds may be considerably in excess of its fundamental
content. However these harmonics are at relatively high frequencies
and their effects are not usually too significant to the motor. They will
cause additional iron and stray losses in the motor and these will have
to be taken into account in deciding the rating to be allocated to the
motor.
The pulse width modulated voltage source inverter system
201
d) The choice of inverter frequency, particularly with gear change PWM
systems is made taking the effect of motor harmonics into account.
With a well designed PWM drive system the level of torque pulsations
in the motor will be very low compared to the other DC link drive
systems described herein. The low harmonic content in the motor
current at all operating speeds dictates this superior performance
where a very smooth torque is generated. Any small pulsation
produced by an inverter frequency current component will normally
not be able to excite any mechanical resonance in the load.
From the input supply side point of view the degree of harmonics in the input
current depends directly on the DC link or supply convertor inductance included
in the design. If a substantial value of DC link reactor or AC line reactor is
included then the DC link current prior to the capacitor will be relatively
smooth and the resulting supply current waveforms will be of quasi-square form
containing 5th, 7th, 1 lth, 13th harmonics etc. If the supply convertor inductance
is negligible then a proportional amount of the high inverter frequency currents
may flow into the mains network.
5.6.5 Speed control accuracy and transient performance
The superior performance capabilities inherent in the PWM drive system means
that it can be used in applications which require relatively high quality performance such as for servo-drives and for robotic actuators etc. Its ability to
provide very good motor performance at low speeds and even at standstill is a
particular merit in this respect.
Clearly when used with an induction motor the accuracy of speed control
achievable will depend on the method of speed measurement employed and it
is usual to fit digital or analogue tacho-generators to achieve the high accuracy
needed.
The transient performance of the drive is in general dictated by the speed of
switching of the inverter and as this is high with PWM system the performance
can be very good. The result is that the overall transient performance achievable
depends on the peak torque capability of the system and on the detailed
parameters of the motor.
Chapter 6
The six step current source
inverter drive
6.1 Introduction
This drive is the current source equivalent of the drive discussed in Chapter 4.
It is a DC link type system with the power being first converted to DC and then
inverted with a square wave six step inverter to produce variable frequency AC
to an induction motor. But in this case the DC link has a relatively large reactor
in it and no capacitor, the result is that the DC link current is relatively smooth
and the circuit current cannot be changed very quickly.
This system has been in use for some time now for relatively simple applications
such as fans and pumps as an alternative to the six step voltage source design.
However it does have some definite advantages in its ability to regenerate motor
power back into the mains supply easily without additional power components,
and the fact that it can be protected against overcurrent more easily then the
voltage source design.
In this system the inverter switches operate to alter the path the current takes
through the circuit and the motor, directing it to those motor windings which
will cause the appropriate level and direction of torque to be produced.
Whereas in the voltage source design the convertor produces a voltage to the
motor and the current drawn by the motor then takes up whatever value is
needed to, in this circuit the current is applied to the motor and the circuit
voltages take up the value and waveform they need to ensure the correct
operating conditions.
This system initially came about when the availability of high quality switches
was limited. This circuit, in general, uses lower performance switches and
reduced rates of rise of current and voltage in the circuit. Some supporters of
this circuit indicate that it can use converter grade thyristors having relatively
long turn-off times and this can be the case if the values of the capacitors and
reactors used are appropriate. Although there is no technical reason why
transistor or gate turn off thyristor switches should not be used with this circuit
it has up to now almost universally been implemented with thyristors.
The operation of a current source inverter circuit is initially described in
Section 3.3.2 and it will help to refer back to this section first.
The six step current source inverter drive
203
6.2 Principles of operation
The elements of this system can best be explained with reference to Fig. 6.2
which shows the mains commutated supply side thyristor phase controlled
convertor which rectifies the mains to produce variable voltage DC for supply
to the motor inverter. The dominant feature of this circuit is the relatively large
DC link reactor which ensures that the DC link current is reasonably smooth
at all times and that the current in the circuit is unable to change quickly. The
Fig. 6.1 This is a 300KW current source inverter for supplying an induction motor driven fan
in an ammonia plant. This design uses the circuit described in section 6.3.4 but with
DC link reactor coils in both the positive and negative connections. The iron cored
reactor is shown on the left and the thyristor assemblies are mounted top right. The
commutating capacitors and voltage suppression circuits are in the rear of the cubicle
{Holee Limited)
204
The six step current source inverter drive
result is that the circuit operation is dominated by the currentflowingin the DC
link.
The inverter consists of a bridge of six switches each of which is capable of
switching the circuit current ON and OFF itself. They may be power transistors,
forced commutated thyristors complete with their commutation components, or
they could be gate turn off thyristors with their gate drive systems. However this
circuit has up to now been mainly implemented using forced commutated
thyristors. The aim of the inverter switches in this circuit is to direct the current
which isflowingin the circuit, into the most appropriate motor windings so as
to achieve the required level and direction of motor torque. Each of the inverter
switches normally carries the full DC link current for one third of each cycle of
motor operation. The result is a quasi-square waveform of currentflowinginto
each motor connection from the inverter.
The frequency of this motor current is dictated by the rate of switching of the
inverter switches and this is usually decided by a voltage controlled variable
frequency oscillator which forms part of the electronic control circuits.
In this circuit, the DC link current alwaysflowsin the one direction whatever
the conditions of operation of the motor and the current alwaysflowsin this one
direction through the inverter so that there is no necessity for reverse conducting
diodes. However, as will be seen later, this does not prevent the correct relationships between motor current and voltage being achieved.
DC link
reactor
inverter
switches
Vs
supply side
thyristor
converter
—
motor side inverter
Fig. 6.2 The six step current source inverter drive
The supply side convertor controls the level of current flowing in the circuit
and the voltage which it needs to produce to do this depends on the motor speed
and loading. The supply side convertor is usually able to provide a negative as
well as a positive voltage to the DC link to cater for the possibility of regenerating power back into the mains supply. This circuit is capable of removing
electrical power from the motor as well as driving it. Which way the power is
flowing at any specific time depends on the phase relationship between the
The six step current source inverter drive
205
motor current and voltage and in this circuit this is just reflected into the level
and polarity of the DC link voltage. Whenever the motor is being driven by the
convertor drive the DC link voltage will be positive as shown in Fig. 6.2 and
when the motor is being braked by the drive the DC link voltage reverses, the
current continuing to flow in the same direction.
The direction of rotation of the motor can be selected at will, electronically,
by just changing the sequence of operation of the inverter switches. Although
this is normally done when the motor is at rest, if it is changed during operation
the result will normally only be reversal of the motor torque causing the motor
to slow down quickly.
As with most induction motor systems, this drive operates normally without
any direct feedbacks from the motor rotor, relying only on the electrical
measurements which can be made on the stator to decide the most suitable
operating conditions. The drive produces an appropriate combination of frequency, current and DC link voltage for the motor and in general the motor is
left to look after itself. The aim of the electronic controls in the drive is to
achieve the most satisfactory combination of these parameters so that the motor
can operate in its most satisfactory and effective way.
In this system it is the value of the DC link voltage which dictates the phase
angle of operation of the motor and hence it dictates how much of the circuit
current goes towards producing motor magnetisation and how much produces
torque. As with all the other systems it is the correct choice of the voltage to
frequency ratio which decides the optimum magnetising conditions in the
motor.
The presence of a large DC link reactor also has an additional benefit in that
it makes this system very robust and enables it to be relatively easily protected
against the consequence of faulty operation. The reactor means that the DC
current cannot change very quickly and this equally applies to fault currents, the
result is that it is easier to retain control over the currentsflowingin the circuit
and it is usually possible to prevent the currents ever rising above the switching
capability of the inverter switches.
The two adverse features of this system are the torque pulsations produced
in the motor and the peak voltages which can occur across the motor windings
due to the inverter switching action used.
The motor winding currents are quasi-square wave in shape and this directly
results in the stator MMF waveform stepping around the stator periphery
rather than smoothly rotating. As explained in Chapter 1 the induced voltages
in the motor windings still retain their basically sinusoidal shape even though
the currents contain significant harmonics. The result of the stepped MMF
waveform, therefore, is that the motor torque is not generated smoothly but it
contains a substantial ripple component related to the operating frequency of
the motor. With this drive the torque generated oscillates about the mean torque
level at a basic frequency of six times the motor frequency. In general these
pulsations do not seriously affect the application of this drive but they do have
206
The six step current source inverter drive
to be taken into account. If the frequency of pulsation happens to correspond
to a mechanical resonant frequency in the load system then the pulsation can
cause much larger variation in the mechanical stress levels in shafts and gears,
etc. Torsionally resiliant couplings may be used to prevent such conditions
occurring. These pulsations also clearly affect the low speed performance which
this drive is capable of producing.
Another disadvantage is peak motor winding voltages. During the switching
of the current from one phase to another the phase currents change relatively
rapidly, rising in one phase and reducing in another. This rate of change of
current induces a corresponding voltage in the leakage reactance of the motor
windings and it is necessary to limit the rate of change of current to a value
which the motor windings can accept. In general peak voltages of up to twice
normal may be allowed to be produced during switching and the effect of this
on the motor winding insulation has to be assessed.
6.3 Detailed analysis of the system
Although this system can use any type of switches for the motor inverter it has
up to now used forced commutated thyristors and in the majority of cases the
inverter circuit which is shown in Fig. 6.12 has been employed. As this circuit
is particularly important to this six step current source drive it is dealt with
separately in Section 6.3.4. However the basic principles of this drive are
applicable to a variety of possible inverter circuits using thyristors, transistors
and gate turn off thyristors. Initially, therefore, this drive will be considered in
its general form with ideal switches in the inverter bridge and this analysis
applies to all inverter circuits including the conventional circuit described in
Section 6.3.4 except for the special points raised here.
This current source drive system is, as expected, dominated in its operation
by the circuit current. The DC link reactor ensures that the current remains
relatively constant during the transfer of the current from one inverter switch to
the next. This same currentflowsin the supply side cables, the supply convertor,
the DC link reactor, the motor inverter, the motor windings and back again via
the negative side of the DC link. The switches in both the supply convertor and
the motor inverter just share this current on a time basis and they ensure it
passes into the correct motor windings at any specific time to produce the
required magnetisation and torque conditions.
In this system it is the voltages which are allowed to vary and which have to
be worked out from the specific conditions of operation of the circuit.
Also, in this system, the DC link voltage and the motor voltages are not
simply related like they are in the six step voltage source drive. The phase
relationship between the motor current and voltage depends on the magnetisation
and torque conditions being demanded and this directly affects the phasing of
the inverter switchings with respect to the motor voltage. As with all other such
The six step current source inverter drive
207
circuits the DC link voltage has to correspond with the motor voltage while the
appropriate inverter switch is closed (i.e. ON) and hence the power factor of the
motor diredly affects the DC link voltage. The control is normally arranged to
ensure a constant motor flux and therefore the motor induced voltage will be
directly proportional to the motor frequency under normal operating conditions. The DC link voltage will then be a direct guide to the real power flowing
in the circuit.
The detailed study of this drive therefore has to start with the DC link current.
6.3.1 Circuit waveforms
Current waveforms
Let us first of all assume, therefore, that a smooth and constant value of DC link
current isflowingand let us deduce how this current flows throughout the rest
of the drive and motor system.
The DC link current flows through the three inverter switches on the same
side of the link in turn, with each switch carrying the full current for a third of
the time. The same occurs on the negative side of the inverter bridge but the
points of switching the current from one switch to the next occur midway
between the switching points on the positive side (assuming steady state
conditions of operation).
Fig. 6.3 shows the sequence of inverter switch firings and the way in which the
DC link current is distributed to the three windings of the motor stator in this
case, assuming the motor stator is star connected. The switch numbers refer
to those shown in Fig. 6.2. The positive link DC current is switched into
switches 1, 3 and 5 in turn and the negative DC link current (which is obviously
the same value as that in the positive side) is sequentially switched into switches
2, 4 and 6. Phase A current is therefore the sum of the currents flowing in
switches 1 and 4 as shown.
Except during the short periods of transfer of the current from one switch to
the next there is only one switch on each side of the bridge which is carrying
current at any one time.
The frequency of switching of the inverter switches will directly decide the
frequency applied to the motor. Six inverter switchings are required to produce
one cycle of motor frequency. The order of switchings does not need to be as
shown in the figure. If the sequence had been 1, 5, 3 and 2, 6, 4 then the motor
would rotate in the opposite direction.
The chosen sequence of switching as above occurs at all times during the
operation of this drive irrespective of the value of the current flowing or the
condition of the motor and load.
If the motor happened to be delta connected then the currentsflowingin the
windings would differ from those shown in Fig. 6.3 which shows the currents
flowing in the cables connected to the motor. In the delta connected case this
current would split between the two windings connected to the same motor
208
The six step current source inverter drive
terminal. Because the current does not change much during the ON period, the
current splits according to the winding resistance only and as there are two
windings in series in one path and only one in the other then the current splits
on a two thirds/one third basis. Fig. 6.4 shows the winding currents in a delta
firing p oints of invi»rter s»witch<?s
;
!5
1
2
:
:I
i
i«
*I
!5
1
DC*ci rrent
1
3
5
1
3
5
2
4
6
2
4
DC-cu rent
6
li ieA
c jrrent
j
B
cur ent
(ine
curr ?nt
Fig. 6.3 Motor line currents
connected motor as related to the line currents being fed into the motor. The
peak of the phase current is equal to two thirds of the DC link current and zero
periods do not occur. Although these phase current waveforms look more
sinusoidal than the line currents they do in fact contain the same proportion of
harmonics.
The six step current source inverter drive
209
These currents flowing in the stator windings cause an air gap MMF
waveform which steps around the stator periphery with each cycle of operation
consisting of six step movements in the MMF.
However, the basic energy transfer from the stator to rotor is dictated by the
fundamental value of the stator currents and by the smoothly rotating fundamental MMF vector which results from these: the harmonics will be treated
separately later.
line A
lineE
eC
phas'A-i
phaseB-C
phas*C-A
Fig. 6.4 Phase currents in delta motor
The rotating MMF induces corresponding currents into the rotor of the
motor and the effective difference between the rotor and stator currents dictates
the magnetising flux in the motor (see Chapter 1).
210
The six step current source inverter drive
In the general case the current being fed into the motor provides both the
necessary magnetising current needs of the motor and the torque needs of the
load. If the torque is low then the majority of the current will be producing air
gap flux and if the load torque is high this will reduce the proportion of the
current which is generating flux, so reducing the flux. In practice the level of
current is altered when the load torque changes so that the flux can be maintained constant. If, for the present, we assume that the motor is running, and
at a constant rated level offluxwe can proceed on to establish the motor voltage
waveforms.
Motor voltage waveforms
As explained in Chapter 1 the magnetising circuit of an induction motor is
basically inductive by nature and tends to ignore the harmonic currents which
may beflowingin the motor stator. The induced voltages in the stator and rotor
windings therefore tend to be fairly sinusoidal even though the stator winding
currents may have quasi-square waveshapes. The motor terminal voltages,
however, also contain additional features caused by the high rates of change of
current which occur every time the inverter switches are commutated. While the
current is transferring from one switch to the next it changes very rapidly in the
motor winding and this causes voltage 'spikes' to occur due to the leakage
resistance of the stator windings. The value of these 'spikes' will depend on the
method of commutation used and the rate at which the current changes,
however most inverter systems try to minimise the commutation time in order
to reduce switching losses and hence, in many practical cases, the voltage peaks
are relatively high compared to the induced sinusoidal voltage.
Fig. 6.5 shows typical voltage waveforms which occur with a Star connected
stator winding. The phase voltage of the motor has four commutation 'spikes'
corresponding to the sudden changes in the winding current and the line voltage
which is the difference between the two phase voltages has six 'spikes' equally
spaced around the sine wave with two being larger than the remainder. Examination of the current waveforms of Fig. 6.5 in a similar way will reveal that the
line voltage with a delta connected motor will be the same as that with a star
connected one. The position of these 'spikes' on the sine wave will depend on
the phase relationship between current and voltage and hence on the particular
loading condition existing.
The magnitude of the sine wave of induced voltage will depend on the motor
flux condition and the frequency and the size of the 'spikes' is related only to the
level of the currentflowingin the circuit so that under low speed conditions the
'spikes' are the dominant feature of the waveform whereas at high speeds the
sine wave is much more significant.
DC link and inverter switch voltage waveforms
The voltage existing on the motor side of the DC link reactor will be decided by
the terminal voltages of the motor and the periods of time when the inverter
The six step current source inverter drive
line
211
urrent
iecurr it
C11 ne cur r ?nt
F I iase v tage
\
B p \ ase v tage
\
A-B
e vott ge
7
\
\
Fig. 6.5 Motor voltage waveforms
switches are closed. When an inverter switch is ON then the DC link voltage
must correspond with the appropriate motor terminal voltage and hence the
total DC link voltage consists of three 120 degree sections of the three motor
waveforms in each cycle. The precise shape of the DC voltage wave depends
212
The six step current source inverter drive
on the points of inverter switching in relation to the motor voltage waveforms.
Fig. 6.6 shows the way in which sections of the motor terminal voltage
waveforms are transferred by the inverter switches to the DC link in the inverter
side of the DC link reactor. The wave contains a six pulse ripple related to motor
frequency and, depending on the type of switches used and the method of
commutation, the voltage 'spikes' also appear in the DC voltage.
The voltage occurring on the other side of the DC link reactor will be dictated
by the supply convertor, it will have the same mean value as that coming back
from the motor (under steady state conditions) but it will contain six pulse
harmonics related to the supply frequency. The voltage across the link reactor
will therefore be a complex mixture of six pulse harmonics of supply and motor
frequencies. In practical circumstances where the reactor has a finite value this
complex voltage gives rise to complex ripples on top of the DC link current.
UQ link'vbltagfc/bn the'motor.iide of.-reactor
high dv/dt
/VcHtag^'dcross \
switch
A-
A
Fig. 6.6 Typical inverter voltages
Fig. 6.6 also shows the voltage which occurs across one of the switches.
Although this waveform changes considerably as the phase angle between
voltage and current changes, this shows that the switches have to be capable of
accepting the full line voltage which occurs across the terminal voltage of the
motor, 'spikes' as well. The transient conditions which occur across the switches
during commutation will also be affected by the characteristics of the switches
The six step current source inverter drive
213
and the method of commutation being used. A more careful study of the
commutation conditions is therefore needed before fully specifying the requirements of the switches from a voltage point of view. The waveform also shows
that there will usually be a high dv/dt occurring at the end of the conduction
period and it may be necessary to include components in the inverter circuit to
reduce this to a level acceptable for the switches.
63.2 The motor vector diagram
Although the motor currents are generally quasi-square in shape the performance of the motor is really dictated by the fundamental sine wave values of these
currents. The harmonics contained in the waveform do not produce any steady
undirectional torque which can do useful work, they just cause ripples in the
mean torque level which can for the present be left for later study.
Similarly the voltage 'spikes' occurring on the motor voltages do not contribute to generation of torque and they can be ignored when considering the
normal operating and performance conditions of the motor.
In the last section I indicated that with this drive the particular operating
condition of the motor dictates the phase relationship between the motor
currents and voltages and the level of the DC link voltage also changes with this
phase relationship. The best way to understand this more fully is to consider a
motor vector diagram which relates these vectors over a range of operating
conditions. Fig. 6.7 shows such a diagram for this current source drive drawn
using the induced voltage vector as a reference because this is the most conventional way that such induction motor vector diagrams are drawn. This shows a
low load current IL at a low power factor cos 4>L and a high load current IH at
a higher power factor cos cj)H with the current locus following a circular shape
between these two points and continuing into the regenerative mode where the
power factor angle increases above 90 degrees. This is in fact the diagram which
would exist for constant flux conditions and other parallel circular current loci
are shown for different constant flux values.
Although this constant flux vector diagram is correct even with this current
dominated system because the control is always arranged to produce these
conditions, it is not the best way to fully understand the system.
In practice a current is fed into the motor and the voltage conditions then
follow according to the load on the motor and hence a vector diagram with the
current as the reference vector can be more enlightening. Fig. 6.8 shows one such
diagram where afixedcurrent at afixedfrequency is applied to the motor. The
circular locus shows the way in which the induced voltage varies as the load on
the motor is changed.
At zero load the full value of the current is used to magnetise the motor and
the maximum induced voltage vector VL (OA) is produced leading the current
by 90 electrical degrees. As the motoring load is increased a larger proportion
of the current is needed to produce torque so that less is available for magnetisation, thus reducing the length of the voltage vector and the angle between
214
The six step current source inverter drive
current and voltage; OB is a typical load condition. If a regenerative load
is applied then the angle between current and voltage increases to the typical
point OC.
induced voltage
Fig. 6.7 Motor vector diagram
This vector diagram does help in realising an important fact associated with
the transient performance of this drive. The torque produced is given by the
inphase component of current multiplied by the magnitude of the voltage. As
the motoring torque is increased the voltage vector follows the locus AB,
reducing in value as the angle between current and voltage decreases. At point
D the maximum torque is produced and further progress down the locus DO
would see the torque reducing again due to the collapse of the motor flux. In
practice torque can only be applied to this drive at a rate decided by the speed
The six step current source inverter drive
215
at which the motor current can be increased. If a sudden torque is applied too
quickly the motor flux can collapse and drive to the load can be lost.
In practical drives therefore the control is arranged to alter the value of
current so that the necessary motor flux is maintained at all times as the load
changes and hence the vector diagram of Fig. 6.8 is correct and appropriate for
steady state conditions.
fixed
current
vector
I mag
Fig. 6.8 The vector diagram of a current source system with current as the reference vector
6.3.3 Circuit relationships and equations
With reference to Fig. 6.2 the two independent controls to this system are the
firing angle of the supply convertor, Alpha, and the frequency of the motor
convertor. All other parameters in this system are dependent on these two
quantities.
The aim of the drive is to provide a current into the motor such that it will
be correctly magnetised and that it will produce the required level of load
torque. The phase angle of the motor vectors will be directly dependent on these
conditions and the value of the DC voltage will be directly affected.
Let us start by assuming that the motor is running at a steady speed on load
(with low slip value), with an input convertor firing angle of Alpha and a DC
link current of Idc. The currentflowingin the DC link will directly indicate the
value of the currentflowingin the motor and the mains supply system. As they
will both be quasi-square in shape then they will both be equal to:
Idc x yj2/^/3
= Idc x -8164 amps RMS.
(1)
216
The six step current source inverter drive
As the work done in the motor is the result of the fundamental value of
the motor current, the fundamental line current to the motor will be given
approximately from:
Idc x -8164 x 1/1-05
= Idc x -778 amps fundamental
(2)
The mean value of the DC voltage will be the result of the firing delay angle of
the supply convertor, Alpha, i.e.
Vdc = 1-35 x Vs x COS (Alpha) approximately
(3)
To be more accurate one should allow for the voltage drop in any supply
reactance and in the convertor itself. If we assume a three per cent drop in
voltage due to these effects at rated load current then
Vdc = 1-35 x Vs x JCOS (Alpha) - 0-03 x i l l
(4)
where Isr is the rated value of the supply line current.
We now know that the power crossing the DC link must be equal to the
product of the mean values of DC current and voltage, i.e.
Pdc = Idc x Vdc
= 1-35 x Idc x Vs x JCOS (Alpha) - -03 x i l l
(5)
I
Isr J
Now let us return to the motor side. As above, a fundamental line current of
approximately -778 times the DC current is being fed into the motor. This
current has to provide the two components of current required for correct motor
operation, namely, the magnetising current and the torque component of current.
In this current fed system it is the voltage which decides how the motor current
splits into these two components. If the motor voltage increases then a larger
magnetising current will be required and this will affect the proportion of the
motor current which will be available to generate torque.
Chapter 1 contains a detailed analysis of the motor, if required, but here I
propose to use a simplified analysis for the purpose of understanding. Fig. 6.9
shows the simplified single phase equivalent circuit and vector diagram assuming that the motor stator resistance is negligible and that the leakage inductances
and iron losses are insignificant. With these assumptions then the input current
Im splits into a magnetising component Imag and a torque component It and
the voltage vector is always at right angles to the magnetising current. From
this, the voltage and magnetising current are related as always by a magnetising
saturation curve and by an equation of the type:
Em = Vsat x ^
x {1 - 2-71(~1'33xImag/Isat)}
(6)
The six step current source inverter drive
217
where Isat and Vsat are shown on Fig. 1.13 and where F is the actual Frequency
and Fr is the rated frequency. The magnetising current is related to the motor
input current by:
Imag = Im x SIN (f>
(7)
where <>
/ is the motor power factor angle.
Imag
magnetising
impedance
Fig. 6.9 Simplified motor conditions
Imag
The other component of the current is given by:
/
It = Im x COS <>
(8)
and also by
It = Em/(R2'/Sl)
(9)
The power into the motor will be the sum of the power in the three phases and
218
The six step current source inverter drive
is given by:
Pm = 3 x It x Vm
= 3 x Vm x Im x COS </>
(10)
The power to drive the load will be given by multiplying the motor input power
by the motor efficiency and the motor torque can be obtained from this and the
speed of the motor. The motor speed can be obtained by the relationship:
S = 120 x F/P x (1 - SI) RPM
(11)
and hence the motor torque in Newton metres will be given by:
Torque = (Pm x Efm x 60)/(2 x n x S)
(12)
Where P = number of poles on the motor
SI = Actual slip
Efm = Motor efficiency
F = Actual frequency.
All that is required now to complete the relationship for the drive as a whole
is to relate the motor conditions to the inverter, DC link and supply. The
currents have already been sorted out in equations (1) and (2), the only remaining item is the relationship between the voltages.
The most accurate method of relating the voltages is to use the powerflowin
the circuit. The power into the motor is the same as the power out of the inverter
and the input power to the inverter will be given by its output power divided by
the inverter efficiency.
Therefore the DC link power is given from (10) as:
Pdc = (3 x Vm x Im x COS 0)/Efinv
(13)
where Efinv = Per unit efficiency of the inverter. The DC link power is also
given by equation (5) so that
Idc x Vdc = (3 x Vm x Im x COS <£)/Efinv
(14)
and using equation (2)
Vdc x Efinv = 3 x Vm x -778 x COS </>
i.e. the motor phase voltage is given by
Vm = (-4284 x Vdc x Efinv)/COS </>
(15)
and the line voltage to the motor
= (0-742 x Vdc x Efinv)/COS 0
(16)
Therefore the DC voltage and the motor voltage are related by the power factor
angle of the motor.
These relationships can be checked against the results taken from a much
more rigorous analysis of this drive, taking account of motor winding resistances,
The six step current source inverter drive
219
leakage reactances and iron losses and the particular loss characteristics of a
typical current source inverter.
Fig. 6.10 shows the effect of variation in the motor frequency while the torque
is kept constant: the DC voltage and therefore the supply power factor reduce
with frequency, whereas the current remains constant because of constant
torque. Fig. 6.11 shows the effect of torque variation under constant frequency
conditions. The circuit current at low torques is just the magnetising current
requirement of the motor and it increases with torque demand. The DC voltage
also alters with load even though the motor frequency voltage and speed are
constant and this is due to the change in motor and inverter power factor.
110 600h
D C current
100 500
total drive
efficiency
80 400
at rated torque
560 300
I
i
40 200
DC voltage
20 100
10
20
30
frequency-Hz
40
50
Fig. 6.10 Variation in frequency
6.3.4 Standard current source inverter circuit
The most frequently used circuit employed with this system is that shown in
Fig. 6.12 and it is necessary to consider this in detail if a complete understanding
of this system is going to be obtained.
The inverter uses six thyristors and six diodes with a delta connected set of
220
The six step current source inverter drive
commutation capacitors on each side of the inverter bridge. The two sides of the
bridge commutate independently and the diodes are to allow the commutating
capacitors to retain their charge during the periods between commutations.
The switching of the current between the thyristors is done by allowing the
current to divert through the capacitors temporarily and the speed of current
change is limited by the leakage reactance of the motor shown diagrammatically
in Fig. 6.12.
120
D.C current
100
500
400
motor current
total
efficiency
u
?60
a
300
01
%
0
200
a
at 30 hertz
frequency
o9
20
100
50
percentage of rated torque
Fig. 6.11
100
Variation in load torque
To show how the switching takes place let usfirstassume that thyristors 1 and
2 are conducting the circuit current which will beflowingthrough the DC link
reactor, thyristor 1, diode 1, motor phase A, phase C, D2, thyristor 2 and back
to the DC link. Under this condition the capacitors will be charged as shown in
Fig. 6.12(a).
When thyristor 3 isfiredthe current immediately transfers into thyristor 3 due
to the charge on C3, the current now flowing through the DC link reactor,
thyristor 3, C3, C5 and Cl, Dl, etc. The current in thyristor 1 immediately drops
to zero and thyristor 1 becomes reversed biased from C3. The rate of switch-
The six step current source inverter drive
221
over of the current to thyristor 3 is very rapid and di/dt reactors may be included
to limit this rate of rise of current (see Fig. 6.12(b)). The capacitors then carry
the load current and gradually change their charge. When Vc3 has reached the
value of Va-b then D3 starts to conduct and D3 and Dl conduct together as the
motor current is transferred from Dl to D3 limited by the leakage reactance of
the motor (see Fig. 6.12(c)). The capacitor charge will continue at a reducing
rate until the current in Dl reaches zero and D3 reaches the full DC level. The
three capacitor voltages will then be as shown in Fig. 6.12(d).
V dc
AJ
&>
complete circuit
Fig. 6.12 CSI commutation
The values of the voltages and currents during this transition are shown in
Fig. 6.13 which is shown for a specific condition of the motor voltages. From
this you will see that:
1) The transfer of current from phase A to phase B is delayed by the time
222
The six step current source inverter drive
that it takes for Cl to discharge to the voltage across lines A and B
(in this case near zero).
2) The current switches from thyristor 1 to thyristor 3 instantly, immediately thyristor 3 is fired. This is where the highest rate of rise of
current occurs.
3) The capacitor voltages are impressed onto the motor terminal voltages
due to the rate of change of the diode and winding currents and the
motor leakage inductance.
turn-off
time
^
»,
VC1
zero
VC3
V
VC3
VC5
VC5
1
IT3
1
zero
IA andID1
\
zero
l B a n d l D3
'
"
zero
——.
-I-"""
V
V
A
——-
-I
I
thyristor 3
fired
6-12(a)
vA
6-12(b)
Fig. 6.13 Commutation currents and voltages
6-12(c)
6-12(d)
The six step current source inverter drive
223
4) All three commutating capacitors on the same side of the bridge are
involved in the switching transition.
The actual voltage conditions which occur in the circuit during the commutation depend on the current in the circuit at the time and the precise values of
the induced voltages in the motor windings.
The capacitors will rise to a voltage approximately decided by the equation:
Capacitor peak volts = E x SIN
+ - x Idc x
where E is the peak value of the motor induced voltage
<f> is the motor power factor angle
L is the commutating inductance of the motor
C is the size of the commutation capacitors
and Idc is the DC link current flowing.
The capacitor voltage will oscillate between this value in either direction and
zero as shown in Fig. 6.14 during each cycle of operation.
voltage across thyristor 1
Fig. 6.14 Circuit voltages
|
224
The six Step current SQUfce inverter drive
The voltage 'spike' which occurs on top of the motor generated sine wave will
also be equal to this value and you will see from the equation that its value can
only be reduced by employing a larger commutating capacitor or by reducing
the motor leakage reactance.
Examination of the diagrams of Fig. 6.12 will show that as one thyristor on
each side of the bridge circuit is always conducting, then the capacitor voltages
will also dictate the voltage across the thyristors; also the diodes isolate the
motor induced voltage from the thyristors. The voltage appearing across the
thyristor 1 during the cycle is shown in Fig. 6.14(b).
The turn-off time allowed for the thyristor is shown in this diagram and on
Fig. 6.13 and as this is the period for the capacitor charge to change then it will
vary with the value of currentflowingand with the point on the motor voltage
wave when the commutation takes place.
In general the values of commutating capacitors are chosen to give turn-off
times for the thyristors which are in excess of 100 microseconds and to give
voltage 'spikes' which are within the reasonable voltage ratings of the diodes
and motors. It is usual to try and keep the maximum voltage 'spike' to less than
the peak of the motor voltage sine wave i.e. restricting the maximum voltage on
the motor to twice the sine wave peak.
6.3.5 Examples of calculations
1) Calculation of rated conditions for a drive
A current source thyristor inverter is to be used to vary the speed of a blower
fan on a boiler in an industrial plant. The fan requires a shaft power of 50 KW
at its rated speed and the 3 phase star connected induction motor driving it will
then be supplied at 600 volts line and will operate at an efficiency of 91 per cent
and a power factor of 0-88 per unit. The inverter drive will be supplied from a
660 volt, 3 phase, 50 hertz supply and it will operate at an efficiency of 90 per
cent when supplying the above motor at its rated load. Assume a quarter of the
drive power losses are in the supply side converter.
Question
Calculate the rated values for:
a) The RMS line current into the motor
b) The DC link current
c) The DC link voltage
d) The supply current from the mains
e) The approximate power factor of the supply current.
Answers
The power into the motor is equal to the power output divided by the efficiency
i.e.
Motor power in = 50/0-91 = 54-95 KW
The six step current source inverter drive
225
The KVA into the motor will be equal to the power in divided by the power
factor
KVA in = 54-95/0-88 = 62-44 KVA
The fundamental line current into the motor will therefore be equal to
62-44 x 1000/(600 x ^3)
= 6008 amps
As the motor input current will be quasi-square in shape then the RMS current
will be approximately 5 per cent above this
i.e.
Im line = 63-09 amps RMS
(a)
From equation (1) then the DC link current is given by
= Im/-8164
= 63-09/-8164
Idc = 77-31 amps DC
(b)
The drive power output to the motor from above
= 54-95 KW
The drive power input will therefore be
= 54-95/-90 = 61-06KW
Therefore the total drive power losses are
= 61-06 - 54-95 = 601 KW
Therefore the supply side convertor power loss
= 1-50KW
and the motor inverter power loss
= 4-51 KW
Therefore the power crossing the DC link
= 54-95 -h 4-51 = 59-46 KW
Therefore the DC link volts
= 59-46 x 1000/77-31 volts
= 769-1 volts DC
(c)
The supply current from the mains will be the same as the RMS line current into
the motor
Is = 6309 amps RMS
(d)
226
The six step current source inverter drive
From equation (4)
COS (Alpha) - 03 = 7691/1-35 x 660
= 0-863
COS (Alpha) = 0-893 per unit.
(e)
2) The effects of incorrect setting up
Question
If the drive was set up incorrectly to operate at a DC link voltage of 725 volts,
approximately what would be the new values of motor current, power factor
and voltage, if the motor magnetisation curve is assumed to be linear.
Answer
If we assume that the DC link power remains the same as in the above calculation then we can find the new DC link current
Idc = 59-46 x 1000/725 = 8201 amps DC
Therefore the motor line current
= 66-96 amps RMS.
The motor fundamental current therefore
= 63-77 amps fundamental.
Assuming that the motor is also operating at the same power then from
equation (10)
54-95 x 1000 = 3 x Im x Vm x COS <t>
where Vm = phase voltage of the motor.
Therefore
Vm x COS 4> = (54-95 x 1000)/(3 x 63-77)
Vm x COS <t> = 287-23.
We now have to find a method which will give the individual voltage and
power factor values from the above.
The result of this change in the DC link voltage has been to increase the motor
current. Clearly the new motor current has to satisfy both the magnetisation and
torque needs of the motor. If the motor voltage reduces then the magnetisation
component of the current can reduce but the torque component of current must
increase.
We therefore need to first of all establish the relationship for the magnetising
current.
In the correct initial setting condition the fundamental motor current was
6008 amps at a power factor of 0-88 p.u. The magnetising current was therefore
The six step current source inverter drive
227
equal to approximately
Im x SIN (ft, i.e. 60-08 x SIN (ACS -88)
= 28-54 amps
and this magnetising current led to a phase voltage on the motor of 346 volts. The
new magnetising current will therefore follow a linear characteristic and will be
given by
Imag = Vm/346 x 28-54
and again this will be equal to the new Imag x SIN (ft
i.e.
63-77 x SIN 0 = (Vm x 28-54)/346
But
Vm x COS (ft = 287-23 from above
Therefore
63-77 x SIN 0 x COS (ft
SIN (ft x COS (ft
(287-23 x 28-54)/346
•3715
Therefore
SIN (20) = -743
2(ft = 48 degrees
(ft = 24 degrees
COS (ft = -9135
Therefore the motor phase voltage is given by
287-23/-9135 = 314-4 volts
Motor line voltage = 545 volts approximately
Therefore summarising, the old and new parameters are
DC volts
DC current
Motor line current
Motor line voltage
Motor power factor
OLD
NEW
769
725 volts DC
820 amps mean.
670 amps RMS.
545 volts.
•9135 p.u.
77-3
631
600
0-88
3) Conditions at reduced speed
Question
If the drive as detailed in Example (1) is to be used at one third of its rated motor
frequency to supply a load requiring 10KW under constant motor flux conditions what will be the values of the following parameters if the motor efficiency
228
The six step current source inverter drive
under this condition is 76 per cent and the inverter efficiency 60 per cent
Motor power factor
Line current in the motor
DC link current
DC link voltage
Answers
At one third motor frequency and constant flux then the motor induced voltage
will be one third of its rated value, approximately 200 volts line. The motor
input power will be equal to
Pm = 10000/76 = 13158 watts.
From equation (10)
Im x COS0 = 13158/(3 x 200/1-732) = 380 amps.
The motor magnetising current requirement is independent of frequency and
so the same value will be required as with the rated condition. The original
magnetising current will be equal to
Im x SIN <j> = 6008 x SIN (ACS -88)
= 28-54 amps,
so now:
Im x SIN <f> = 28-54
Im x COS <f> = 3800
Therefore TAN $ = 0-751
cj) = 36-9 degrees
Therefore COS <j> = 0-8 p.u.
Im = 47-5 amps fundamental
Im = 49-9 amps RMS.
From equation (1)
Idc = 611 amps DC.
The power in the DC link equals the motor input power divided by the inverter
efficiency
Pdc = 13158/0-6 = 21,930
Therefore
Vdc = 21930/61 1 = 359 volts.
The six step current source inverter drive
229
6.4 Practical circuit design considerations
This section is included to assist in understanding the practical designs which are
in use. All drives need to include additional facilities and features to make the
drive reliable and economic. They need the necessary auxiliary power supplies
for the electronic circuits, items to protect the unit and its components against
faulty operation, items to extract the heat losses, etc. In addition different
manufacturers have different approaches to these requirements and this section
may help to explain some of these variations. It is the aim, however, to deal
mainly with those items which are directly associated with this particular current
source six step drive system.
6.4.1 Over current protection
In common with most current source systems the presence of a reasonable sized
DC link reactor in the circuit makes the system relatively easy to protect against
overcurrents. The circuit current normally stays relatively steady and the
maximum rate of change of current is dictated by the circuit voltage and the
reactor inductance.
The protection of the circuit against mis-commutation of the inverter and
overloading of the motor, etc. can all be carried out by using the supply
convertor which is usually a six pulse thyristor convertor and which can invert
the circuit current to zero in a cycle or two and certainly before any damage is
done to the circuit switches. The supply convertor is usually current controlled
with current limiting facilities and very fast phase back in case of detected
overcurrents.
The only case where these facilities are ineffective is during regeneration when
the supply convertor is already operating in its inversion mode and the circuit
current is being forced by the motor and the motor inverter. A sudden disturbance
in the mains supply can cause an inversion failure and the DC link current
passes straight through the supply convertor. To protect against this it is
necessary to apply current limit control to the motor inverter to cause it to phase
back to reduce the current to zero. This is usually done on a drive which is
intended for regular inversion duty.
For these reasons non-regenerative systems may well measure current for
protection purposes either on the DC side or on the mains AC input. However
regenerative systems are more likely to measure currents in the DC link or on
the motor connections.
If high speed switching devices such as transistors or GTO thyristors are being
used in this circuit it is not necessary to fit the same highly sensitive overcurrent
switching protection as is required in voltage source inverters because the circuit
current is limited and cannot rise quickly. In practice it is better to temporarily
remove the firing signals from the switches when a fault is detected or to switch
all switches into the ON state. Damage is usually done to such switches when
an attempt is made to turn them off at too high a current. In this system it is
230
The six step current source inverter drive
usually sufficient to use the supply converter as the means of turning the circuit
current off, backed up by a supply switch or circuit breaker.
Another important reason for keeping a tight control over the circuit current
is that the circuit peak voltages (see next section) are dependent on the level of
current and these voltages can quickly damage the semiconductor switches if
currents in excess of the design level are allowed to exist in the circuit.
6.4.2 Overvoltage protection
The main problem with the overall design of this six step current source system
is in controlling the voltages which occur in the circuit due to the motor inverter
commutation. As shown in Fig. 6.5 voltage 'spikes' are caused mainly by the
current changing in the leakage reactance of the motor.
Clearly this voltage will depend on the rate of change of the current and if
transistors or gate turn off thyristors are being used the preference would be for
relatively fast switching which would produce very high voltages.
Most circuits of this type therefore include some means of limiting the
magnitude of these 'spikes'.
-W-
-tt-
diode
rectifier
clamp
drive
inverter
motor
Fig. 6.15
Suppression of voltage 'spikes'
With the conventional circuit discussed in Section 6.3.4 the size of commutating capacitors and the preference to use convertor grade thyristors usually
means that the transfer of current is relatively slow and the voltages generated
during the 'spikes' is usually no more than the peak circuit voltages leading to
the maximum circuit voltages during the 'spikes' being up to twice the normal
sine wave peak. This may be acceptable in some designs and no further measures
may be included.
When suppression of these 'spikes' is necessary the main method used is to
The six step current source inverter drive
231
add voltage clamping circuits to the inverter/motor which will slow down the
rate of change of current and hence limit the size of the voltage 'spikes'.
The simplest system is shown in Fig. 6.15 in which a capacitor is connected
to the motor via a diode rectifier, and a resistor is connected across the capacitor
to discharge the capacitor between commutations. This works by diverting some
of the current during commutations into the capacitor thus slowing down the
rate of change of current in the motor and hence reducing the peak voltages.
However in this form there is a significant power loss in the resistor which
reduces the efficiency of the drive. A more satisfactory arrangement is produced
if the resistor current is switched on by a voltage controlled semiconductor
switch so that the resistor current onlyflowsduring the voltage 'spike' and even
then only when the 'spike' would otherwise exceed the acceptable level.
Even more complex arrangements have been proposed particularly when
transistor or GTO thyristor switches are being employed. They are all based on
this clamp approach but the resistor is replaced by a system to feed the energy
back into the DC link to increase the system efficiency.
6.4.3 Circuit variations
There is a surprising degree of agreement between designs of this current source
drive and very few important variations in the circuit exist. Most of those
units already in operation use the thyristor/diode inverter bridge detailed in
Section 6.3.4 and the six pulse thyristor converter on the supply side is universal.
In some instances the DC link reactor may be split into two with half the
inductance on each DC connection. This is done to ensure protection against all
earth faults but it is not all that common in practice.
Well designed units will include snubber circuits across all the semiconductors
and small di/dt reactors may be included in the circuit, particularly in series with
the thyristors of the circuit described in Section 6.3.4.
It is possible to use a half controlled bridge for the supply side converter but
this will then prevent regeneration back into the supply occurring, therefore
some means of preventing regeneration will have to be included. This approach
also makes protection more difficult as the current in the DC link cannot now
be reduced to zero quickly. Very few current source drives include fuses or
circuit breakers in the main loop because a sudden chop off of current could
cause high voltages to occur in the circuit inductances. Reliance is usually placed
on static protection to reduce current with a supply side switch as a back up.
6.4.4 Factors affecting the specification of main components
The supply side convertor is a very conventional circuit as used in a wide variety
of DC drives and convertors. No special measures are needed for this drive. The
presence of a significant DC link reactor means that the current is continuous
over a wide range of circuit operation.
The DC link reactor is a dominating item in the circuit and its size will be
chosen from consideration of the following points:
232
The six step current source inverter drive
a) It will limit the rate of rise of load current and hence the transient
performance of the drive. The application of the maximum DC
voltage to its inductance will give the maximum rate of rise of circuit
current.
b) The reactor is included to produce continuous current flow in the
circuit in spite of the large amount of ripple in the voltage impressed
on its terminals.
c) The rate of rise of fault current is also dictated by this reactor and it
may be chosen so that inversion failure or miscommutation faults can
be dealt with by gate control over the semiconductors in the circuit.
d) The reactor may be air cored or iron cored but in this latter case
saturation of the core on overload may reduce its effectiveness in
limiting fault currents.
The components in the motor inverter circuit are very much decided by the
particular inverter switches used. With the circuit discussed in Section 6.3.4 the
commutating capacitor is one of the most important items. A large value of this
capacitor will lead to lower circuit voltages and longer turn off times for the
thyristors but this can be expensive and the optimum size is aimed at. A size
which will allow thyristors with readily available turn off times to be obtained
and which does not lead to circuit voltages much higher than twice the normal
peak sine wave is normally a suitable economic compromise.
Because of the tight current control employed it is not normally necessary to
allow large margins in current rating on any item in the circuit.
The motor needs to be considered with some caution due to the two important
factors of voltage 'spikes' and harmonics. The transient voltage 'spikes' can be
quite high particularly when fast switches are employed and the rate of rise of
voltage can be important. Such 'spikes' can overstress the windings of the
motor, particularly those without significant insulation safety margins, because
the stresses may not be equally shared through the stator winding coils. Most
motors will accept some increase in peak voltage but if the motor has already
been in service for some time it would be preferable to see that the drive includes
a voltage clamping circuit. The currents in the motor windings are of quasisquare wave shape containing a total harmoninc content of around 25 per cent.
The harmonics cause additional winding and iron losses and allowance has to
be made by derating the motor by approximately 10 per cent from its sinusoidal
duty.
6.5 Overall control methods
There are only two independently controllable variables in this drive system, the
phase angle of the supply side convertor and the frequency of the motor inverter.
These two factors have to be controlled together in order to provide the
optimum motor conditions to achieye maximum torque.
The six step current source inverter drive
233
The supply side convertor controls the DC voltage and it can be used to
decide the current flowing in the circuit.
The frequency of the motor inverter controls the speed of rotation of the
stator MMF wave and hence directly affects the speed of the rotor of the motor
as long as a reasonable level of flux is generated.
The motor flux is the result of the current being supplied to the motor, the
frequency of the inverter and the torque load that is being put onto the motor.
The flux then dictates the motor induced voltage and hence the terminal
voltage.
If optimum conditions are to be obtained it is essential to ensure that the
voltage induced into the motor windings is maintained proportional to the
frequency of the inverter and the only way in which this can be done is to use
the supply side convertor as the means of controlling this. It is not able to do
this job simply because the DC voltage is related to the induced voltage by the
power factor of the motor and this can change over quite a wide range. However
the supply convertor is able to cause the appropriate change in the current
fed into the motor to effect the necessary control over the motor flux
conditions.
Fig. 6.16 shows the fundamentals of the control of most drives of this type.
The supply convertor (1) is controlled via a firing circuit (2) by a fast closed
loop current control which is included for protection purposes. The measurement of current can be anywhere in the circuit as the same current passes
through the mains input connections, the supply side convertor, the DC link, the
motor inverter and the motor. The reference to the current loop is usually a
voltage control amplifier system (6) based on a motor voltage measurement. The
reference for the voltage loop is the frequency reference (which dictates the
inverter frequency) and maybe a boost signal (10) derived from the same source
to ensure constant motor flux at low speeds.
The inverter is controlled directly from the speed or frequency reference via
an oscillator (11).
You will notice that no slip compensation arrangements are made in this
system. This is because there is no convenient measurement which is related
directly to motor torque. The circuit current is not suitable because of the wide
variation of motor power factor with load. To obtain such an indication it
would be necessary to measure DC link voltage, multiply it by the current and
then divide it by frequency — a complex and relatively expensive procedure.
Protection against overloading normally operates into the frequency control
system because this will result in the drive speed being reduced if overloading
occurred. Perhaps the approach of including the current limit in the supply
convertor current loop might be thought to be satisfactory but it isn't
because it would just cause a reduction in DC link volts and motor power
factor and loss of flux in the motor leading to its falling out of control
completely.
D.C. reactor
ramp
15
frequency 7
current
limit
frequency
setting
D.C.
current
Fig. 6.16 Control of a six step current source drive
voltage
amplifier
current
amplifier
firing
circuit
pulse
amplifiers
5
shunt
k k k «
supply convertor
V/f
function
voltage
controlled
oscillator
inverter switch
drive circuits
* * I I
inverter
10
13
direction
11
12
motor
voltage
motor
CD
CO
S8
The six step current source inverter drive
235
6.6 Performance and application
The six step current source induction motor drive is in general a robust, reliable
and well protected drive for general purposes rather than sophisticated use. Its
ability to regenerate the motor power easily into the mains system to produce
controlled braking is a significant benefit which has led to its widespread use on
machinery for moving large quantities of earth or ore. Such systems need to be
well protected because one of the most frequent fault conditions is associated
with mains supply loss or disturbance during regeneration and drives have to be
capable of being returned to service quickly after such an event.
However in its simplest form the drive does not possess much overload
capacity due to the likelihood of producing very high peak voltages in the circuit
under such conditions. It is necessary to take account of the full duty in deciding
the size of the motor and drive for any specific application. Nevertheless the
consequence of overloading is not normally too severe because the tight current
limit system normally included just causes speed reduction until the load
demanded is within the rating.
The majority of the drives of this type in service use the capacitor commutated
arrangement described in Section 6.3.4 and it will be noted that the motor
leakage inductance is an important feature of the operation of this circuit. The
practical result of this is that significant variation of this factor is not normally
possible and this drive is in general considered to be a single motor drive where
the motor and drive have to be matched together to achieve optimum results.
In this respect it is not asflexibleas the voltage source systems which can readily
be considered as multi-motor drives where the number of motors connected can
be changed without affecting the drive itself except in its loading.
The drive is usually used for power frequency applications in the range 5 to
100 hertz.
6.6.1 TorqueIspeed characteristics
This drive is normally operated over a limited speed range of say 20 per cent to
100 per cent of nominal frequency and lower speed operation is usually limited
to starting due to the relatively high level of torque pulsations (see Section 6.6.4).
The torque capability is usually limited by the level of current to which the drive
has been designed; most inverter designs are only allowed to carry a current up
to a specific level so as to avoid the possibility of inversion failure of the
switches. This value will usually be just a little above the rated current for the
system. The size of the drive and motor has to be selected to be appropriate for
the peak load torque duty required.
If the control arrangements provided on the drive are capable of ensuring that
the rated flux can be maintained over the speed range then the drive is usually
capable of producing a consistent torque over this range. The drive can run
successfully at any value of torque below this maximum limit.
Even under light load conditions the circuit current is rarely allowed to drop
236
The six step current source inverter drive
below the normal level of motor magnetisation current and this may be a factor
in the choice of the motor inverter switch and commutation system.
Because it is necessary to maintain control over the DC link voltage at all
times it is not normal for these drives to be allowed to operate in the constant
motor voltage, reduced flux, high speed region. If operation in this area is
essential then the voltage source drive is more appropriate.
6.6.2 Efficiency
The capacitor commutated thyristor system normally employed with this drive
produces a high efficiency drive because the commutation energy is retained in
the commutation capacitors for use at the next commutation and no significant
energy loss occurs. There will be an increase in the motor power loss due to the
harmonic content in the motor current but this is not usually large. The situation
is however only the case if the peak voltages generated by commutation are
within the capabilities of the circuit components including the motor. If it is
necessary to suppress the voltage 'spikes' as referred to in Section 6.4.2 the
additional losses in the clamp can be very large even amounting to as high as
10 per cent or more of the drive power rating. This loss can, as mentioned, be
avoided if a more complex clamp system is included which causes this energy to
be returned to the DC link.
6.6.3 Supply power factor
As with all drives having a phase controlled thyristor input convertor, the input
power factor is directly dependent on the DC link voltage. In this drive the DC
link voltage varies with the speed and with the load so that wide variations of
supply power factor are possible.
The relationship between DC volts and motor speed and load is shown in
Fig. 6.17. Under rated torque conditions the motor power factor will remain
reasonably high and the DC voltage will increase approximately linearly with
speed and frequency. When reduced torque is demanded the effect is for both
the motor current and the motor power factor to reduce and the result is a
reduction in the DC voltage so that at very low load torque the DC voltage
reduces to low value while the current remains at the motor magnetising current
level. In practice the flow of the magnetising current in the inverter and motor
causes some power losses and the DC voltage adjusts itself to supply the total
required power including the load and the motor inverter losses. The need to
supply these losses dictates the lower limit of the DC voltage as shown in
Fig. 6.11.
6.6.4 Torque pulsations
Unfortunately the square wave currents fed into the motor by this drive result
in the torque generated not being smooth but containing a significant amount
of sixth harmonic ripple.
As the current wave is basically quasi-square the MMF of the stator winding
steps around the air gap rather than rotating smoothly (which would be the case
The six step current source inverter drive
237
with sinusoidal currents). The rotor tends to rotate at a relatively smooth speed
due to the presence of mechanical inertia in the shaft system and the stepped
MMF reacting with the rotor produces an oscillation in the torque applied to
the rotor.
100
high constant torque
^ 50
50
motor speed 7o
100
Fig. 6.17 Supply power factor
Another way of looking at it is to consider the torque being generated from
the interaction of the air gap flux and the stator MMF. Due to the inherent
inductance of the magnetic circuit it is found that in spite of the square current
waveforms thefluxin the air gap is sinusoidal at fundamental motor frequency.
The torque pulsations are then the result of the square stator currents reacting
with the sinusoidal flux.
Fig. 6.18 shows plots of the torque typically generated by an induction motor
fed from this drive system, with its large sixth harmonic content.
As the torque ripple is the result of the current waveform it is reasonable to
understand that the magnitude of the ripple will increase with the value of the
current. The value of the mean torque also increases with the current (assuming
low slip operation) and so the relationship between mean torque and torque
ripple tends to remain as in Fig. 6.18 at reasonable values of mean torque. At
low values of load however the torque ripple will still be related to the level of
magnetising current being fed into the motor even though the mean torque will
238
The six step current source inverter drive
be very low and hence the pulsations then become much larger than the mean
torque. Peak/peak torques of 30-40 per cent of the rated torque are not unusual
under no-load circumstances.
(a)
at high load torque
time
zero
(b)
at no load torque
zero
Fig. 6.18 CSI torque pulsations
In general these torque pulsations are not harmful to the system as the load
inertia tends to smooth out the result. The pulsations however occur at six times
the operating frequency and at a low frequency the pulsation frequency reduces
to the point where the motor steps round rather than rotating smoothly. It is usual
therefore to limit the normal operating speed range of this system to be from 10
per cent up to rated speed and to fix aflexiblecouple to prevent the pulsations
setting up resonant oscillations or vibrations at the higher operating speeds.
Various methods have been proposed to reduce the level of these pulsations,
see Bibliography, and there is no doubt that with the increasing use of gate turn
off thyristors and transistors in current source circuits such methods will become
more widespread.
Chapter 7
The six step synchro-convertor
system for synchronous motors
7.1 Introduction
This current source drive is naturally commutated and specifically for use with
synchronous motors. Because it uses convertor grade thyristors it is possible to
produce the larger power ratings economically and systems operating at up to
11 kV for high power applications.
This drive has been variously called the synchro-drive, the load commutated
inverter, the brushless DC motor, etc. and it is used under various trade names
such as SYNCDRIVE (G.E.C.), LCI (G.E.), SYNCHROSIL (BRUSH).
The principle used in this drive wasfirstused as a means of electrically starting
large pumped-storage motors. The convertor was applied to the machine stator
terminals and the frequency was slowly increased to accelerate the large generator
with its turbine. When it was up to speed the generator could be synchronised
onto the supply system busbars and the convertor disconnected. It is still used
for this and similar purposes but it now has a much wider role as a fully
controlled variable speed drive suitable for many industrial applications.
Because of the ability to produce large drives of this type at frequencies well
in excess of the 50 or 60 hertz from mains networks this drive is now extensively
used as an electrical replacement for steam or gas turbines running at speeds of
up to 10,000 RPM.
As the system is fully regenerative in its standard form it is often used as a
fully variable load system capable of recovering the power from a variable speed
machine and feeding it back into the electrical power network. The same
principle is also employed in electrical transmission networks to tie two different
AC power systems together, with a DC link to allow flow in either direction.
There are many such DC transmission links now in service around the world
running at very high powers and voltages and they are all very closely related
to this drive system.
240
The six step synchro-convertor system for synchronous motors
Fig. 7.1 This cubicle contains all the power electronic equipment for a 550 KW synchroconvertor drive for a 415 volt brushless synchronous motor. The two identical thyristor
bridges are shown centre right alongside each other and the AC controller for the field
exciter is immediately below them. In this case the electronic control cards are
mounted on the inside of the door, which is shown open in this photograph. (G.E.C.
Industrial Controls, Ltd.)
The six step synchro-convertor system for synchronous motors
241
7.2 Principles of operation
The synchro-convertor drive is again a DC link system in which the fixed
frequency mains power is first rectified into DC before being converted into the
chosen frequency by another convertor. The particular feature of this drive is
that it uses a synchronous motor as a means of commutating the inverter
thyristors. The synchronous motor is always capable of generating sinusoidal
voltage waveforms on its terminals and these are used to allow the motor
convertor to commutate naturally in synchronism with these sine waves.
supply side
thyristor
convertor
k
supply
DC link
• reactor
motor side
thyristor
convertor
—TVYVYW-
v
synchronous
motor
dc
i i > i t t
n
excitation
firing circuits
Fig. 7.2 The six step synchro-convertor drive system
Fig. 7.2 shows the basic circuit diagram of this scheme. The supply side
convertor operates at mains frequency using the mains sine waves for commutation and it is a conventional bridge which would be used for a DC drive. The
motor side convertor operates at the frequency corresponding to the motor
speed and the firing and commutation of the convertor is directly controlled by
the motor voltage waveforms. This convertor is therefore locked in synchronism
with the machine at all times. The inductance in the DC link is there to smooth
the DC link current and to allow the two convertors to operate independently.
The way in which the motor convertor switching is locked to the motor
rotation means that it acts in a very similar way to the mechanical commutator
of a DC machine. The thyristors switch the current into the appropriate motor
windings in order to generate unidirectional torque. Clearly as only six switchings occur per cycle and in this time the motor rotor will have moved the
distance spanned by two poles, it is equivalent to a very coarse commutator but,
nevertheless, the analogy is useful and relevant. In fact, this drive system has
other similarities to a separately excited DC motor drive. If the armature voltage
to a DC motor is changed, its speed will change in the same way, i.e. increasing
the voltage will cause the speed to increase and vice versa. The same is true of
this system. If the DC link voltage is increased then the motor speed will also
increase in proportion. The response to changes in excitation is also similar, in
242
The six step synchro-convertor system for synchronous motors
that, a reduction in field current will cause the speed to rise and a strengthening
of the field will cause the speed to drop.
It is not therefore surprising that some people refer to this drive as the
brushless DC motor drive.
The principles of operation explained up to now are those which occur while
the motor is running and while the motor is generating voltages on to its
terminals. At low speeds, however, the level of the voltage generated will be too
low to ensure correct switching of the convertor thyristors, and an alternative
method is needed. The simplest method and that normally employed by drive
manufacturers is to use the switching capability of the supply convertor (which
is available at all times) to assist in the commutation of the motor convertor. At
low speeds the current in the motor convertor is changed from one thyristor to
another by first switching the current off using the supply convertor, then
altering thefiringpattern of the motor convertor, and then switching the current
back on. These two operating modes, those at high speed and low speed will now
be described in turn in more detail.
7.2.1 Starting and low speed operation
Starting the motor from rest is achieved by switching current into the motor
windings so that interaction between this current and the motor flux will cause
the correct direction of torque to be developed so that the motor turns in the
required direction. Initially, two of the motor convertor thyristors are selected
for switching on, e.g. thyristors 1 and 6. If now the supply convertor is fired to
produce a low voltage, current can flow in the DC link through thyristor 1 into
the A and B motor windings and return via thyristor 6. It can be shown that this
will generate a torque in the motor if the field is already applied and that the
value of torque developed will depend on the position of the rotor (and hence
thefieldflux).Fig. 7.3 shows that the torque/position relationship is sinusoidal
and that it can be either direction depending on the rotor position.
The aim when starting this drive is to select the position which will give the
maximum torque, i.e. the position X-Y on the diagram. This is done by fitting
a rotor position sensor to the motor and using it to decide which thyristors to
fire. Let us assume that the rotor is in the 60 electrical degree position when the
current is applied, a high torque will be developed and this will move the rotor
forward. When the rotor reaches the 120 degree angle the torque would start to
drop. In fact there are six potential selections of pairs of motor convertor
thyristors and each selection will result in a different torque/position sine wave
as shown in Fig. 7.3. Once the rotor has reached point Y therefore, the aim is
to switch the current on so that it nowflowsinto thyristors 1 and 2 in order that
the maximum level of torque can continue to be developed. This is achieved by
removing the firing pulses from thyristors 1 and 6, phasing back the supply
convertor to reduce the current to zero, firing up thyristors 1 and 2 and finally
turning the current on again. This sequence is shown on Fig. 7.4 where the rates
of reduction and rise of currents are dictated by the circuit voltage levels and the
value of the DC link reactor. The period of delay before switching the current
The six step synchro-convenor system for synchronous motors
243
back on is dependent on the reliability of the method used to check that zero
current has been reached. When the rotor has moved another 60 electrical
degrees it will be time to switch the current over again to thyristors 3 and 2
following the same sequence. The initiation of the sequence of switching is
provided by the rotor position encoder which enables the optimum point of
switching to be used.
motor convertor thyristors conducting
/0\
SO,
\ rotor position
torque .with
\ thyristors 5 and 6/V
conducting
/
\
\
Fig. 7.3 Low speed torque
y
production
remove firing frc
from
ind
thyristors land6
fire thyristors
1 and 2
i
^ ^
i
supply convertor
voltage increased
DC link
current
period to confirm
current is zero
Fig. 7.4 Low speed commutation
Fig. 7.5 shows the currents and where they flow during the start up sequence
of this drive and Fig. 7.3 shows that if the optimum points of switchover are
used then a high level of torque will be developed. This same figure also shows
244
The six step synchro-convenor
system for synchronous motors
that a reverse torque and therefore reverse rotation can be produced just by
altering the selection of thyristors fired, e.g. if thyristors 1 and 6 were fired
during the rotor position 240 degrees to 300 degrees then full reverse torque
would be available.
switching points
DC. link current
1
,
I*
/
v\
}
\,
\
motor thyris ors fired
1
1
3
3
6
2
2
A
line A
urrent
\
/
5
1
1
3
3
6
6
2
2
A
\
line B
\
5
\
/
/
I
urrent
/
/
lineC
\
/
\
M
irrent
\_J
Fig. 7.5 Motor currents at low speed
It is also possible to deduce from Fig. 7.3 that the accuracy of the switching
points is not very critical. As long as the switchings take place within 60 electrical degrees of the ideal point the torque will still be in the correct direction.
This leads onto the point that it is possible to start up this drive without a shaft
position sensor as long as the starting requirements are not too critical. When
operated in this way it is necessary to know the initial position of the rotor and
from then on the rate of switching is gradually increased to match the motor
inertia and the starting current employed.
High torque can be developed during starting using a position encoder but as
there is a loss of torque during the switching sequence, when the current is
turned off, the torque will reduce as the speed (and therefore the rate of
switching) increases. Although in many systems it is possible to operate at up
to 20 Hz or more in this starting mode, it is usual to restrict this mode to only
The six step synchro-convertor system for synchronous motors
245
a few per cent of the full speed and then change over to the normal running
mode.
7.2.2 Normal running conditions
When the motor is generating sufficient voltage on its terminals to allow satisfactory natural commutation of the motor convertor thyristors the drive is
operated in its normal running mode.
The supply side convertor is controlled so that the appropriate level of current
is circulated in the system and the motor convertor thyristors direct the current
into the correct motor windings to obtain motoring torque. The supply convertor
is therefore operated as a rectifier and the motor convertor in the inversion mode
passing power from the DC link to the motor. Reference back to Chapter 3 may
be worthwhile at this point as Section 3.2 deals with natural commutation of
bridge convertors under rectification and inversion conditions.
In this case the magnitude of the motor terminal voltages varies with the
speed of the motor (assuming constant excitation) and it is always necessary to
operate the inverter near to its maximum voltage inversion condition. The firing
points of the motor convertor thyristors have to be selected so that the current
can be transferred safely from the outgoing to the incoming thyristor under the
influence of the motor voltage sine waves. This is shown in Fig. 7.6 which shows
the waveforms occurring during the process of switching from thyristors 4 to 6.
The firing of thyristor 6 will, because Vb is more positive than Va, cause the
current in thyristor 4 to reduce and that in thyristor 6 to increase.
The rate of change of these currents depends on the value of the effective
inductance of the motor and the magnitude of the voltages while the current is
changing. In addition the period when both thyristors are conducting (the
overlap angle) will depend on the level of current flowing in the circuit. The
overlap period must be completed before point X is reached otherwise the
voltage which is causing the current transfer will reverse and the transfer will not
be completed, resulting in an inversion failure and a high fault current in the
circuit. Hence thyristor 6 must be fired a sufficient time before X to ensure that
the current has come to zero in thyristor 4 and to ensure that this thyristor has
fully recovered its voltage blocking ability before point X has been reached.
This specific point is very important to many of the limiting rating conditions
of this drive because:
1) The maximum current which can be switched, and therefore the
maximum power which can be passed to the motor, is directly
dependent on the period of overlap, and therefore on the motor's
effective inductance.
2) The motor will always pperate at a leading power factor because the
current in the motor windings must start well prior to point X and the
value of the power factor will depend on the point of firing of the
motor convertor thyristors.
3) The most effective use of the motor can be made by using terminal
246
The six step synchro-convertor system for synchronous motors
voltage measurements to decide the point of firing of the thyristors.
If, alternatively, a shaft position encoder was used to decide the firing
points then allowance would have to be made for the phase shift
occurring in the motor terminal voltage as the load circuit changes
(see Section 7.3.2).
4) The value of the DC voltage, under any specific operating condition
is directly dependent on the motor terminal voltage and the firing
point of the motor convertor thyristors.
\
current in thynstor A and line A
firing of thyiristor6
current in thynstor 6 and line B
^ over lap angle u.
Fig. 7.6 Normal running commutation
Under this normal running mode of operation therefore it is usual to tie the
motor convertor firing to terminal voltage measurements and the optimum
operating conditions will occur with the motor operating at its highest power
factor and the DC voltage being approximately proportional to motor speed.
The six step synchro-convertor system for synchronous motors
247
7.2.3 Reversing and regeneration
The direction of rotation of the motor is decided only by the particular sequence
of the motor convertor thyristors and this can usually be decided electronically.
It is not therefore necessary to reverse the motor cables to change the direction
of the motor and if it is beneficial the motor can be easily arranged to go in either
direction.
The direction of powerflowcan also be reversed if required so that the motor
load energy is fed back through the convertor to the mains supply system.
Inspection of Fig. 7.2 will shows that the two convertors are in fact identical
and there is no reason why their roles should not be reversed with the motor
convertor operating as a rectifier and the supply side convertor operating in its
inversion mode. This can in fact readily be done without any additional power
equipment just be altering the firing points of the thyristors so that the DC
voltage reverses. Fig. 7.7 shows these two conditions of motoring and regenerating showing that the DC current continues toflowin the same direction and that
the reversal of voltage is achieved by changing the firing points of both motor
and supply convertors.
firing
angle
D.C current
150
0 to90°
motoring
power
flow
motor
zero0
90° to 160°
£
D.C.current
regenerating
Fig. 7.7 Motoring and regeneration
This facility can be used temporarily for slowing down the motor quickly and
this is particularly useful if a high inertia load is connected, or it can be used
continuously so that energy can be removed from the electrical machine (now
248
The six step synchro-convertor system for synchronous motors
operating as a generator) and whatever machine is connected mechanically
to it.
7.2.4 Motor excitation
Up to now I have assumed that the flux in the motor is kept constant and this
is in fact the main aim of the excitation system that is used with this drive.
Reference back to Chapter 1 will show that the rotatingfieldsynchronous motor
is a non-compensated machine in which the stator current alters the value of the
flux and therefore it is essential that continuous control over the excitation is
employed if optimum performance is to be achieved. Hence in all drives of this
type an automatically controlled excitation system is employed.
With a slip ring motor this will take the form of a thyristor convertor to
provide the correct level of field current to ensure that the flux remains
approximately the same under all speed and load conditions.
The majority of drives use brushless excitation to avoid the maintenance
necessary when sliprings and brushes are used. In this case the requirement to
provide high excitation even at standstill means that the methods conventionally
used with fixed speed synchronous machines are not suitable. To achieve this
requirement it is necessary to use a rotating exciter machine of the induction
generator or rotary transformer type directly mounted on the shaft of the main
motor. The 3 phase AC stator field is then arranged to rotate in the opposite
direction to the main motor in order to ensure that excitation is available over
the whole of the speed range.
three-phase supply
thyristor
controller
rotating assembly
rotary
transformer
exciter
diode
rectifier
to synchro convertor
motor
field
winding
Fig. 7.8 Brush/ess excitation
Fig. 7.8 shows the electrical circuit of a typical brushless excitation system
with the power for excitation being obtained from the rotor of a rotary transformer exciter and being rectified by rotating diodes to provide the necessary
DC current for the main motorfield.Control of the level of excitation is carried
out by a thyristor AC voltage controller connected to the exciter stator winding.
In such an arrangement it should be appreciated that the output of the exciter
and therefore the amount of field current will depend not only on the voltage
applied to the exciter stator but also on the speed at which the motor is running.
The six step synchro-convertor system for synchronous motors
249
Also the frequency in the rotor circuit will increase as the motor speed increases.
The magnitude of the increases in frequency and exciter power output will also
depend on the number of poles on the exciter and the range of speed variation.
It should also be understood that in such a brushless excitation system the
excitation power needed by the main motor field winding will originate from
two sources, namely, the three phase supply on the exciter stator and from the
motor shaft via the exciter. At standstill all the power comes from the electrical
supply and as the motor speed increases more of the excitation power is
provided from the shaft.
The amount of torque which will be available in the main motor will depend
directly on the air gap flux level and if a high torque is required at low speeds
then the excitation system will have to be designed so that it can provide the
necessary field current when the exciter is only able to provide its minimum
output i.e. at low speeds. At higher motor speeds the exciter stator voltage will
then in general be much reduced even though full field current is still available
in the main motor.
As will be seen later in Section 7.3.2 the amount of field current required in
the main motor will vary considerably with the level of stator load current and
hence continuous control over the motorfieldcurrent is needed to allow for this
also.
In most drive systems of this type the field control system will be organised
so that the air gap flux is maintained relatively constant even though changes
in speed and load may be occurring all the time.
7.3 Detailed analysis of the system
In this section it is my intention to study the operation of this drive while in its
normal running mode only. The starting mode is normally only a transient
condition and it has been looked at sufficiently already.
This drive is a current source system in which the DC link reactor prevents
rapid changes in circuit current and hence the starting point in studying it in
more detail is the relatively smooth current flowing in the DC link.
It is also a system in which the operation of the motor convertor is intimately
associated with the characteristics of the motor and one in which the control of
the field has to be taken into account along with the stator conditions.
I have therefore split this section into four sub-sections dealing with the stator
conditions first, then the motor and field conditions. The other two sections try
to consider the system in total and try to take account of all the interactions
which occur.
73.1 Convertor and motor waveforms
When a synchronous motor is running and is provided with a field current it
produces good sinusoidal induced voltage waveforms the magnitude of which
250
The six step synchro-convertor system for synchronous motors
are proportional to the air gap flux level and the speed of the motor, and the
frequency is proportional to the speed. The terminal voltage of the motor will
differ slightly from the induced voltage as a result of the stator winding current
passing through the windings resistance and leakage reactance.
The supply side convertor is assumed to be connected to a fixed frequency
sinusoidal three phase system operating at constant voltage. Reference can be
made to Section 3.2.1 regarding the supply convertor as it is simply a naturally
commutated bridge-connected convertor giving a variable voltage output to a
high inductance DC link circuit. The DC voltage being produced by this
convertor will contain an amount of harmonics, mainly at six times the mains
frequency, depending on the level of DC mean voltage being produced see
Fig. 3.11. A wide range of phase angle control is used in the supply convertor
to enable the variable DC link voltage to be produced from afixedmains supply.
The motor convertor, normally operating in the inversion mode, will be fired
synchronously with the motor's rotation and its range of firing angle will be
relatively small so as to keep the power factor of the motor as high as possible.
The motor voltage as seen by the DC link will therefore contain a small amount
of harmonic at six times the motor frequency and its mean value will be
approximately proportional to the motor voltage and hence proportional to the
motor speed and air gap flux. For this section let us assume that the air gap flux
is maintained constant by the field control system, in which case the motor
voltage and the DC voltage will both rise and fall with speed.
The current flowing in the DC link is still relatively smooth and continuous
even in spite of the harmonics in the DC voltages at either side of the link reactor
and the supply convertor chops the DC current into sections which flow in the
AC supply and the motor convertor chops the same current up into similar
sections but at a different frequency to flow in the motor windings.
Fig. 7.9 shows the motor stator and motor convertor waveforms which occur
over the whole of the speed range. The motor currents are quasi-square wave
in shape and the only factor which varies on these is the time it takes for the
current to rise to the DC value. In practice the top of the motor current waves
will not be so smooth as it is not practicable to include too high a value of DC
reactor. The top of these waves will reflect the ripple in the DC link current.
The motor phase and terminal voltage waveforms will contain the conventional notches in them due to the periods of commutation and in practice there
will be a certain amount of'ringing' occurring at these points due to the presence
of snubber circuits, etc.
The DC voltage shown in Fig. 7.9 is that which appears on the motor side of
the DC link reactor. As you can see it is relatively smooth with only a small
amount of sixth harmonic. On the supply side the mean value of the voltage will
be the same but it will contain a substantial amount of sixth harmonic as
detailed in Fig. 3.11. The voltage across the reactor is the sum of the supply side
harmonics and the motor side harmonics.
The current in the DC link will then be the result of the complex DC link
The six step synchro-convenor system for synchronous motors
251
reactor voltage and its inductance. This voltage will be the sum of the harmonics
from both the supply and motor convertors, which will be occurring at different
frequencies, and the reactor impedance to each of these frequencies may be
different, particularly when an iron cored reactor is in use. In practice, therefore
the DC current will contain a small amount of harmonic ripple whose size will
depend on the DC reactor. The magnitude and frequencies of this ripple will
vary significantly as the conditions of the drive are changed.
1
ZL
f s \
I
line A current
vz
line B current
line C current
A
2
DC voltage
Fig. 7.9 Synchro-comertor waveforms
7.3.2 Armature reaction
In this system the motor and convertor have to be considered together as they
are interdependent. One of the important factors in this respect is the effect of
the stator current on the air gap flux which is referred to as armature reaction.
252
The six step synchro-convertor system for synchronous motors
In Chapter 1 it is clearly explained that the currentsflowingin a 3 phase stator
winding will produce a continuously rotating MMF waveform and in the case
of a synchronous motor this pattern will be rotating at the same speed as the
rotatingfield.ThefieldMMF waveform and the stator MMF waveforms therefore combine to produce the resultant air gap flux. The magnitude of the stator
MMF will be proportional to stator current and its position in space in relation
to the rotor MMF will be dependent on the phase of the stator current. Hence
the resultant MMF and flux will vary as the stator current and its phase angle
changes.
Reference back to Chapter 1 shows that the stator winding MMF will directly
oppose the rotor field MMF if the stator current is 90 degrees leading the
voltage, i.e. zero power factor. It also shows that the resultant flux can best be
found by an MMF vector diagram as in Figs. 1.20 and 1.22.
In practice the magnitude of the armature reaction MMF in synchronous
motors of this type may be as large or larger than the resultant MMF required
to generate the rated value of air gap flux and hence the field MMF may have
to be considerably larger at full stator current than is needed at no load.
100
300
500
100
1
o
§
1 50
100 kw, 400 volt
synchronous motor
50
motor stator current %>
Fig. 7.10 Exciter performance
100
The six step synchro-convertor system for synchronous motors
253
In this drive the power factor angle in Fig. 1.20(b) is decided by the needs of
thyristor commutation and its value will be at least 20 electrical degrees and
often above 40 degrees at peak load.
Armature reaction clearly has a major effect on the field system of this drive
and as the optimum conditions can usually be obtained by aiming at a constant
level of air gap flux it is clear that thefieldwinding has to be capable of producing much higher values of MMF due to this effect. Fig. 7.10 puts this in
graphical form for a typical brushless motor showing the level of exciter capability
required to achieve constant air gap flux.
From this you can see that it is the low speed, high stator current condition
which requires the maximum exciter output. With the brushless system the
exciter becomes more effective at the higher speeds due to the increase in exciter
rotor volts and frequency. This means that the exciter stator voltage required
reduces as the speed increases.
7.3.3 The motor vector diagram
Although the motor current is quasi-square in shape and contains substantial
levels of harmonics these do not contribute to the development of torque in the
motor. The harmonics effectively produce rotating fields which move at many
times the speed of the rotor and hence the development of real unidirectional
torque is not possible. In fact these harmonics cause harmonic pulsations in the
torque without affecting the mean torque, they also cause some additional
power losses.
The point I am coming to is that the power operating conditions of the motor
can be calculated and understood using the fundamental values of the stator
current and a vector diagram can help in appreciating the relationship which
exists between the motor and the motor convertor.
IX
angle of
firing, beta
IR
y
/delta
power
factor
angle
Fig. 7.11 Motor vector diagram
airgap flux 0 g
applied
field MMF
beta-u/2
254
The six step synchro-convertor system for synchronous motors
Fig. 7.11 is the result of drawing two related diagrams together; the flux
diagram as in the previous section and the conventional current/voltage vector
diagram.
The diagram has many similarities to that drawn in Fig. 1.23 but now the
motor always operates in a leading power factor condition.
The induced voltage in the stator winding Em is produced by the air gap flux
(pg rotating at the speed of the motor. This flux is the result of the combined
effect of the MMF produced by the field winding and that generated by the
stator current. Because the induced voltage is the result of the rate of change of
flux it is conventionally shown at right angles to the flux. On the same basis the
armature reaction MMF is in phase with the stator current and hence the field
MMF required can be drawn and the diagram shows that it is usually larger than
the resultant MMF and that the rotor must adjust itself by the angle Gamma.
The stator current is shown at a leading power factor as required to ensure
satisfactory convertor commutation. The actual thyristor firing point is shown
at a leading angle of Beta to the induced voltage (from which it is normally
derived), the angular difference u/2 being the result of the commutation time
required for the current to transfer from one thyristor arm to another (u being
the overlap angle shown in Fig. 7.6).
The fundamental value of the terminal voltage will be slightly different from
the induced voltage due to the overlap notches caused by the current and this
can be taken into account on the vector diagram by the ImRs voltage drop due
to the stator resistance and the ImXs drop due to the stator leakage reactance.
The resulting angular difference Delta leads to the final power factor angle
shown.
In most drives of this type thefiringangle varies between 20 and 40 electrical
degrees depending on the load current if the air gap flux is kept sensibly
constant. Therefore the main effects of load changes are in the length of the field
MMF vector and the value of the rotor angle Gamma.
7.3.4 Relationships and equations
The vector diagram and the circuit of Fig. 7.2 should be referred to in order to
establish the relationships below.
Because the current is reasonably smooth in the DC link and the input and
motor currents are both quasi-square in shape then the supply and motor
currents
= Idc x ^= amps RMS
(1)
or
x/2
1
= Idc x ^-= x
amps fundamental
(2)
73
1-05
The mean value of the DC voltage is the result of the firing phase delay
(Alpha) applied to the supply convertor.
i.e.
Vdc = 1-35 x Vs x COS (Alpha) approximately
(3)
The six step synchro-convertor system for synchronous motors
255
To be more accurate one should allow for the voltage drop in any supply
reactance and in the convertor itself. If we assume a three per cent drop in
voltage at rated load current due to these effects then:
Vdc =
1-35 x Vs(COS (Alpha) -
03 x Is/Isr)
(4)
where Isr = the related value of the supply line current.
We now move to the motor side and to the vector diagram. The fundamental
motor current, as related to Idc in Equation (2), is Im in the vector diagram
— the phase current in the motor (assuming a star connected motor stator
winding).
Due to the operation of the motor convertor the fundamental value of the
motor phase voltage will be given by:
Vm = Vdc/(l-35 x COS (Beta - u/2 - Delta) x ^3)
(5)
In practice the values of Vm and Em are usually very close and Delta is
reasonably small but the values of Em and Delta can be found from a knowledge
of the motor stator leakage reactance and resistance values. The motor's leakage
reactance is also needed to estimate the value of u/2 which is also reasonably
small.
For all general calculations u/2 and Delta can be neglected and Em can be
assumed to be equal to Vm. Then
Em = V m ^ Vdc/(V3 x 1-35 x COS (Beta))
(6)
As this value of Em has to be generated by the flux 0g, rotating at motor speed
S then:
Sr x ^
x ^ RPM
(7)
Emr
<£g
Where the parameters with an r suffix are the values which occur under rated
operating conditions, i.e. full speed and load.
The power at the DC link we will assume to be Pdc where:
S
=
Pdc = Idc x Vdc
(8)
and if we neglect the power losses in the circuit (which are usually less than eight
per cent) then this power must equal the motor power which must equal:
Pm = 3 x Im x Vm x COS (Beta - u/2 - Delta)
(9)
The motor torque will be given by:
Motor power (KW) x 1000 x 60 ^T
F
Tm =
,DVDU;—r
Newton metres
Speed (RPM) x 2 x n
Pm x 9 5 5 0 ^
Tm =
Newton metres
(10)
256
The six step synchro-convertor system for synchronous motors
The relationship associated with torque can be obtained from equation (9).
Torque is proportional to power divided by speed.
i.e.
Tm oc Pm/S
i.e. from (9)
Tm oc
Im x Vm x Motor power factor
Motor voltage is however the result of the air gap flux and the motor speed:
Vm oc Flux x S
Therefore motor torque is proportional to:
Motor current (Im) x flux x Motor power factor
(11)
DC volts at rated speed
DCV
DCV
40%
20%
50
percent rated torque
100
Fig. 7.12 Variation of load torque
The graphs of Figs. 7.12 and 7.13 have been drawn to show these relationships
in practice. They were taken on a 200 KW, 1500 RPM, brushless synchroconvertor drive system operating with a motor voltage of 400 volts line, the
drive being fed from a 415 volt, 3 phase, 50 hertz supply.
The six step synchro-convertor system for synchronous motors
257
DC current at 100°/o
torque
100°/»T
motor volts
50
percent rated speed
100
Fig. 7.13 Variation of speed
7.3.5 Examples of calculations
1) Calculation of related currents and voltages
Question
A 100 KW output drive is to be designed to operate at 1,000 RPM with a motor
line terminal voltage of 400 volts using a four pole motor. The mains supply is
at 415 volts 3 phase 50 Hz. The motor efficiency at rated conditions is to be 95
per cent and the convertor efficiency 98-5 per cent with the convertor losses
equally distributed between the supply convertor, motor convertor and DC link
reactor. If the motor convertor is operated at a fii ng angle so as to make the
motor power factor 0-9 per unit find the approximate values of the following:
a) Rated motor line current.
b) Rated DC link current.
c) DC link voltage on the supply side of the DC link reactor.
d) The supply current.
e) The Power Factor (COS(Alpha)) of the supply current.
258
The six step synchro-convertor system for synchronous motors
Answers
Rated motor line current
Motor power output = 100 kW
Motor power input = -—-.
efficiency
= -— = 105-3 KW
-95
From equation (9)
Im =
Motor
Powerx Factor
/3 x Line
Voltage
Power Factor
105-3 x 1000
= 168-9 amps
73 x 400 x -9
This is the fundamental value of the current as it is only this which produces
power output. The RMS line current will be approximately 5 per cent above the
fundamental value.
Im (RMS) = 177-3 amps
Rated DC link current
From Equation (1)
Idc = ^ p x Im (RMS)
= 217-2 amps mean
DC link voltage
Power losses in the motor convertor and DC link reactor total 1 per cent,
therefore:
Power on the supply side of the DC link reactor equals:
^
= 106-4 KW
From equation (8)
Vdc = DC link power
DC link current
106-4 x 1000
217-2
= 490 volts DC mean
The six step synchro-convertor system for synchronous motors
259
Supply current
This is the same value as the motor current and is therefore:
Is (RMS) =
177-3 amps
Supply power factor
From Equation (3)
COS (Alpha)
Supply power
=
=
-875 per unit
=
DC link power
supply convertor efficiency
1064
•995
Supply KVA
49
°
1-35 x 415
Vdc
1-35 x Vs
=
177 3
'
=
106-9 KW
',:::
" ^
Supply power factor = -——• =
=
127.4 KVA
-839 per cent
2) Variable speed operation
Question
With the above drive what will be the approximate values of the same parameters when the drive is running at a speed of 600 RPM and driving a load
torque of 500 Newton Metres. Efficiencies and motor power factor can be
assumed to remain unaltered and motor air gap flux can be assumed to be
controlled at a constant level.
Answers
Motor current
From equation (10)
Motor output power =
Torque x Speed _,, T
^—
KW
9550
500 x 600
KW
9550
= 31-4 KW
Motor input power
31-4
= ——- = 33-1 KW
260
The six step synchro-conyertor system for synchronous motors
Motor induced voltage will be proportional to speed therefore:
600
Vm (line) = 400 x - ~
1UUU
= 240 volts.
From (9)
Im (line) = —=
= 88-5 amps fundamental
y/3 x 240 x -9
Im(RMS) = 92-9 amps
DC link current
From (1)
Idc = ^ = x 92-9 = 113-8 amps mean
DC link voltage
Power on the supply side of the DC link reactor equals:
H I = 33-4 KW
From (8)
Vdc
=
33-4 x 1000
^M o ,
TTTS
= 293-8 volts
113*o
Supply current
Is (RMS) = 92-9 amps.
Supply power factor
From (3)
COS (Alpha)
293*8
= j^^—rrr
=
524 p.u.
33-4
Supply power = —— = 33-6 KW
Supply
KVA
=
9 2 9 X 415 X
Supply power factor =
1000
66-77
^
= 66-77
= -503 p.u.
The six step synchro-convertor system for synchronous motors
261
3) Field conditions
Question
If, in the above motor drive, the armature reaction MMF at rated current is
equal to the resultant MMF to produce rated air gap flux. What values of field
MMF is required under rated conditions if the angle Delta on Fig. 7.11 is four
degrees?
Answer
From Fig. 7.11
Beta - u/2 = ACS (0-9) + Delta
= 25-84 + 4 = 29-84
If the rated resultant MMF = 100, then the field MMF is found by solving
the MMF triangle:
,
TAN (Gamma) =
TAXWr
100 x COS 29-84
1 0 0 + 1 0 0 x SIN 2-84
= -579
Gamma = 301
100 + 100 x SIN 29-84
Field MMF =
COS (Gamma)
= 100 x 1-731 = 173 units
7.4 Practical circuit design considerations
The fact that this system uses naturally commutated thyristors universally,
directly dictates the practical aspects of the system. The techniques of cooling
and protection are well documented elsewhere and I will only deal with the more
specific aspects here.
The presence of a relatively large DC link reactor is an important factor. It
ensures the independent operation of the two convertors and smooths the DC
current. It limits the rate of rise of currents and therefore assists in protection.
The supply convertor is connected to a mains supply system which will be
subject to faults and surges at times and these need to be taken into account. The
motor convertor conditions are more predictable, being caused by motor
circumstances which can be under direct control.
Although this circuit is a current source one (having some similarities with the
circuit of Chapter 6) it does not produce high motor voltage 'spikes' during
switching. The rate of commutation of the current in the motor convertor is
dictated directly by the induced voltages in the motor and the motor's leakage
262
The six step synchro-convertor system for synchronous motors
reactance and as a result voltage peaks are directly related to the motor sine
waves.
Because the system is naturally commutated in both sides it is possible to use
combinations of thyristors in each arm of the circuit. Series or parallel operation
of thyristors is entirely practical as long as appropriate measures are taken to
ensure that the voltage or current is shared reasonably equally between them.
7.4.1 Over current protection
The DC link reactor reduces the rate of rise of current in the circuit to predictable levels. Even under the worst case condition when both convertors are
incorrectly operated at their maximum possible positive voltage levels then the
average rate of rise of current in the DC link is given by the equation:
di
-max
=
2 x Vdc max
Lr
Where Lr = inductance of the reactor, and from a fault point of view it is the
air cored value which is most appropriate to be used. If an iron cored inductor
is used it will have a higher inductance than this at normal circuit current levels.
In practice if the maximum voltage was suddenly applied to the reactor the
fault current would rise asymmetrically at a slightly higher rate than the average.
However the normal approach to protection of this drive is to switch the current
off before it has time to reach its full short circuit level and this average rate of
rise approach is quite acceptable.
motor
fault
limiting
reactors
Fig. 7.14 Methods of overcurrent protection
Many of operational faults which can occur in this circuit result in a fault
current in the DC link as shown in Fig. 7.14. The current in the loop is normally
the result of unbalance between the voltages either side of the link reactor. If this
balance is disturbed then higher currents circulate. While the drive is motoring,
faults in the supply side convertor will only produce a modest value of
The six step synchro-convertor system for synchronous motors
263
unbalanced voltage and hence fault current, as it is already producing a positive
voltage proportional to speed. The motor convertor, however, is normally
inverting and any fault in this would result in its voltage reversing, hence
producing a much larger fault current.
When the drive is regenerating, the roles of the two convertors are reversed
and, hence, supply side faults can be more serious. The most common fault in
these conditions is a short loss of mains supply causing an inversion failure on
the supply convertor.
The maximum fault conditions can only apply at high speeds when the motor
voltage is high.
A number of methods are adopted by different manufacturers to cope with
these conditions and some of them are shown in Fig. 7.14.
All control systems will include normal current limit features to ensure that
whenever possible the current is never allowed to exceed normal levels and the
measures below are only called into play if this feature is unable to maintain its
proper control.
Fuses may be included in the convertors to open the fault current loop
whenever a high current occurs and a DC circuit breaker can be included in the
loop for the same purpose. These methods are not universal however for this
drive, partly because they may not be necessary and partly because interrupting
the current sharply in a current source, reactor dominated circuit is not always
the best thing to do; high voltages can be induced into the circuit and the
currents are not easily broken by fuses or arcs.
With proper design the circuit can usually be protected statically. Because
there are two convertors in the fault loop, controlling the one which is not faulty
can usually cause the current to be brought down. If only motoring is involved
then the worst case faults are caused by the motor convertor and current
measurement and fast phase back on the supply convertor can be very effective.
If regeneration is regularly required then a similar arrangement may be applied
to the motor convertor.
The above deals with all faults where the current flows in the DC reactor.
There are some cases where this does not occur. If a short circuit occurs in the
i supply convertor or on its output terminals then the fault can rise extremely
quickly (unlimited by the DC reactor) and fast phase back may not be quick
enough. Reactors in the input supply lines, convertor fuses and the AC circuit
1 breaker may be used to prevent this rare fault causing any damage. If the same
< occurs on the motoring side then fast suppression of the field will eventually
i remove the fault; the motor will however limit the level of fault current which
< :an flow.
; 7.4.2 Factors affecting the specification of the main circuit components
The choice of the DC link reactor in this circuit depends on a number of
i important factors:
a) The size of the reactor directly affects the level of ripple current in
r
264
The six step synchro-convertor system for synchronous motors
Fig. 7.15 The picture shows a 1000 KW, 3,300 volt synchro-convertor drive suitable for
use in a wide range of variable speed synchronous motor applications. The convertor
bridges are contained in the left hand cubicle and the electronics in the central
cubicle. The right hand cubicle contains the auxiliary and annunciator circuits
used to customise the equipment for the particular application. (G.E.C. Industrial
Controls Ltd.)
The six step synchro-convertor system for synchronous motors
265
the DC link and the degree of impedance of operation of the two
convenors. A large high inductance reactor would be preferable from
this point of view.
b) It is necessary to include sufficient inductance in the circuit to enable
good overcurrent protection.
c) The reactor will limit the rate of rise of current which can be achieved
during normal operation of the drive and this may restrict the transient
performance obtainable.
d) Its size and cost.
As usual therefore the final choice is a compromise between having a large size
to give good smoothing and protection and a smaller unit taking up less space
and cost and allowing fast current changes to achieve good performance.
The reactor can be iron or air cored depending on the importance of the above
factors. An iron cored reactor will have a higher inductance at rated currents
and below but the iron is usually allowed to saturate under fault conditions and
the inductance gradually reduces to the air cored value.
The motor for this drive is more similar to a synchronous alternator or
generator than to a fixed speed synchronous motor. All fixed speed motors
contain significant design alterations to allow them to be self started. They are
in fact a cross between a synchronous machine and an induction machine. An
alternator is normally run up to speed via the prime mover and therefore does
not need such modification. With this drive the motor and convertor are always
in synchronism and the torque is always developed due to the synchronous
action at all speeds.
Because the motor reactance affects the commutation of the motor convertor
its value is important to the performance achievable for the drive; in general
lower reactance values are preferable.
The motor windings always carry a square wave current containing significant
harmonics and although these only increase the RMS value of the current by less
than 5 per cent they can cause other stray effects in the iron circuit which have
to be allowed for. These harmonics make it essential for the iron circuit to be
made of laminated steel sheet. If a solid iron rotor is used or considered,
allowance has to be made for the high eddy currents which mayflowdue to these
harmonics and the losses and temperatures that these will produce. In the case
of high speed motors where laminations cannot be used for mechanical reasons
special measures may be needed to reduce the level of harmonics in the system.
The cooling arrangements made for the motor may affect the torque/speed
capability of the system in the same way as with all other variable speed drives
under consideration. If high torques and currents are needed at low speeds then
additional cooling arrangements may need to be made because the rotor fan
itself may be insufficient.
7.4.3 Circuit variations
This drive is often used for high powers and the variations which are made to
the drive are usually the result of it being a high power application.
266
The six step synchro-convertor system for synchronous motors
Although the supply harmonics are in similar proportions to these produced
by the other drives their importance increases with the size of the drive and it
may be necessary to use the same techniques which are used on large DC drives.
The supply side convertor can be split up into two or more bridge circuits
connected in series or parallel: if the voltage to these are phase displaced then
some of the largest harmonics can be removed. If it is split into two then a twelve
pulse circuit having negligible fifth and seventh harmonics will be produced.
This is usually done by including supply transformers with multiple phase
displaced secondary windings.
The harmonics in the motor convertor not only cause extra motor heating but
they also cause the torque to pulsate rather than being completely smooth (see
Section 7.6.5). The same techniques as used to reduce supply side harmonics can
be used on the motor side by splitting the motor convertor into a number of
bridges and feeding the motor via a multi-winding transformer. A better way is
to split the motor winding into two phase displaced sections and to feed these
from separate motor convertor bridges connected in series or parallel into the
DC link. This again removes the fifth and seventh harmonics and considerably
improves the torque pulsations and, as it happens, the effects of harmonics in
the iron circuit of the motor. The individual motor windings however still carry
six pulse quasi-square wave currents.
7.5 Overall control methods
In this drive there are three independently controllable parameters in addition
to the load itself. Phase control of the supply convertor will alter the DC link
voltage, phase control of the motor convertor temporarily affects the balance of
the voltages on the DC link but mainly alters the motor power factor and the
effectiveness of the current in producing motor torque. Control of the field
current alters the air gap flux and the induced voltage of the motor.
The speed, the frequency and the circuit currents cannot be controlled directly
except via the above three controllable variables. The frequency and speed will
adjust themselves automatically to obtain a balance in the voltages on the DC
link and alterations in any of the three controllable variables will result in a
change in the frequency and speed. The interactions are even more complex
because any change in circuit current, and hence motor current, directly affects
the air gapfluxand this then affects the speed and frequency, and this may affect
the current, and so on and so on.
It is therefore essential for all three controllable variables to be continuously
adjusted to obtain the optimum response to any change in load, or speed
requirement, or in supply voltage. Although this is in practice the case it is usual
for the three variables to look after individual functions and they should be
discussed separately.
They will be explained in relation to Fig. 7.16, which shows a typical overall
control scheme for this drive.
feedback
Fig. 7.16 Synchro-convertor control scheme
ACCB
synchronous
motor
shaft
position
encoder
No
1
3
o
I
CO
I
3
I
CO
><•
CO
s
268
The s/'x step synchro-convertor system for synchronous motors
7.5.1 Supply convenor control
Phase shift control over the supply converter directly results in a change in the
DC link volts but it is usual to alter this by adding a high speed current control
loop so that the circuit current is closely controlled by this convertor ((1) on
Fig. 7.16).
If the motor air gap flux is controlled to be constant and the motor convertor
is maintained at a relatively constant firing angle then the DC link voltage will
be approximately proportional to speed. This leads directly to the conclusion
that the supply convertor is the most suitable vehicle for the speed control of the
drive (2).
What this means in practice is that the supply convertor control is very similar
to that of a convertor for a separately excited DC motor drive. It is often
possible to use a standard DC motor control arrangement satisfactorily with
this drive.
In the circuit shown in Fig. 7.16 the overall speed control is carried out via
a frequency measurement (6) derived from the encoder during starting and from
the motor volts at high speed.
It is necessary also to add suitable arrangements for operation at low speed
where the supply convertor is turned on and off six times per cycle of output.
This is shown as Box (3) on the diagram. The encoder is used to initiate
commutation and this controller inhibits the motor convertor and tells the
supply convertor to turn the current off. When the current is proved to be zero
and the motor convertor firing pattern is changed then the pulses and current
are released again.
When the voltage detected on the motor is sufficient the mode switch (4) is
changed over and the starting mode (3) inhibited.
7.5.2 Motor convertor control
Alteration in the firing point of the thyristors of the motor convertor has two
direct effects:
(a) It changes the ratio between the motor voltage and the DC link
voltage.
(b) It alters the effectiveness of the current to produce torque. Equation
(11) in Section 7.3.4 shows that torque varies as the cosine of the
firing angle, Beta.
The most effective way to use the motor convertor is therefore to operate it
at as low a firing angle as possible. In this way the currentflowingwill produce
the maximum possible torque that it can. This is the way in which most drives
of this type are operated. Thefiringangle is usually derived from a measurement
of the terminal or induced voltage from the machine and its value is set to be
the minimum consistant with safe commutation.
The safe value of Beta does vary with the level of current flowing: at higher
currents the angle of overlap increases and the system used to control Beta may
increase its value as the current rises.
The six step synchro-convertor system for synchronous motors
269
7.5.3 Excitation control
Normally the aim of the excitation control is to insure that the air gap flux
remains at its designated level under all conditions of operation, so that
maximum motor torque can be produced. It therefore has to compensate for the
effects of armature reaction and, if a brushless excitation system is used, for the
variation in power output and voltage from the exciter. As these will alter
significantly whenever any change to the drive condition occurs it is preferable
to find a way of monitoring and hence controlling the air gapfluxdirectly using
the field controller.
A flux measuring coil can be built into the motor to obtain such a direct
measurement and there is no doubt this would be a very good way of insuring
proper control over this variable. However this is often inconvenient and costly
and a more practical method is available.
If the air gap flux is maintained constant then the motor induced voltage will
be directly proportional to speed and frequency. Therefore if measurements are
made of the voltage and speed or frequency a measure of air gap flux can be
obtained by diving the two. Most drives are operated on this principle using a
volts/frequency measurement as the feedback for the excitation control system
(see (5) in Fig. 7.16).
In some circumstances, for example, to allow the drive to run faster under
reduced torque, it may be sensible to reduce the air gapfluxto prevent the motor
voltage from rising too high and facilities may be included for this purpose.
7.6 Performance and application
This drive is a particularly rugged and well protected drive which is capable of
performing very satisfactorily in a wide range of practical circumstances. It has
good torque capability over a wide range of speed and its ability to regenerate
motor power back into the supply for controlled braking can be very useful.
Because it is naturally commutated it is relatively simple in its basic operation
and the fact that it has many similarities to a DC motor drive system makes it
attractive and understandable.
It is a very efficient drive system because it does not employ any forced
commutation circuits and because the synchronous motor in general has less
losses than its induction motor equivalent.
There are however some disadvantages to this drive as well:
It is a current source drive with a relatively large DC link reactor which means
that the motor currents are quasi-square waveshape and the result is pulsations
in the motor torque which can cause mechanical resonance in the load system.
The DC reactor also limits the rate of rise of circuit current and this affects
the dynamic performance which this drive can achieve.
The armature reaction effects in the motor limit the peak level of torque which
can be achieved. However, this drive is still as good as most of the other AC
270
The six step synchro-convertor system for synchronous motors
systems in this respect. In any case, higher peak torques can always be designed
into systems by using a larger drive.
As the drive and motor are so closely related together during operation this
drive is, in general, for use with one motor only.
speed
Fig. 7.17 Torque/speed capabilities
7.6.1 Torque/speed characteristics
In the running mode of operation the current which can be satisfactorily
commutated by the motor convertor depends on the level of voltage being
generated by the motor. This has a direct effect on the torque which this drive
can produce particularly at low speeds. This can be seen in Fig. 7.17 which is
drawn assuming constant air gap flux in the motor. This shows clearly why two
modes of operation are necessary for starting and running. However, let us stay
with the running mode for the present. The level of the torque at the higher
speeds depends on the degree of armature reaction in the motor and the ability
of the field system to maintain constant flux. A typical motor current/torque
curve is shown in Fig. 7.18. Up to point A the field system is fully able to
maintain air gap flux but from there on thefluxis reduced by the stator current
The six step synchro-convertor system for synchronous motors
271
and, hence, the torque reaches a maximum and then reduces. Motors will
normally be used in the linear portion of this curve and they can be designed to
produce whatever peak torque/rated torque ratio required by making the motor
larger.
stator current
Fig. 7.18 Torque at high currents
In the starting mode when an encoder is used to ensure optimum conditions,
more torque per amp of current is available than in the running mode, but as
explained earlier the torque will reduce as the speed and frequency rises due to
the periods of zero current. This is shown in Fig. 7.17 showing a linear reduction
in torque with speed; the angle of this line depends on the width of the zero
current periods.
The shaded area shows the range over which changeover to the running mode
is carried out.
With these characteristics the drive is able to be used for a wide range of
constant torque applications as well as for variable torque, fan, pump and
compressor loads.
7.6.2 Efficiency
This drive is one of the most efficient systems discussed in this book because of
its natural commutation method of operation and its use of a low loss motor.
Also the No Load losses are relatively low so that the efficiency stays high over
a wide speed range. Fig. 7.19 shows the efficiency and losses occurring in a
complete drive system under different load conditions, this is typical for this
drive.
272
The six step synchro-convertor system for synchronous motors
7.6.3 Speed control accuracy
The synchronous motor is inherently afixedspeed motor because it always runs
in synchronism with the applied frequency. Changes in load and torque only
cause an angular change in the rotor position with respect to the rotating stator
field. Because of this most drives of this type have a speed control system based
not on a tacho-generator but on a frequency measurement from the stator. With
this simple and reliable system it is possible to achieve a very high order of speed
control accuracy particularly over the top half of the speed range.
efficiency curves
100 - constant torque
ciency, pen
ft
r 50
1
y^^^
-
^^-torque (x speed2
^t^00^"^
1 torque constant
'
—
^
- ^ ^ ^ ^
power loss curves
torque oc speed2
0
100,
~
/
/
-
o
a
- 50
^
50
percent rated speed
Fig. 7.19 Typical efficiency and power loss curves
10
This method, however, is not so good at low speeds due to the fact that
usually the frequency being measured is relatively low and the stator voltage
zero crossing method used, only allows six points to be monitored per cycle. If
high accuracy is required at low speeds the drive can be fitted with an analogue
or digital tacho-generator which will give a fast and steady measurement even
at low speeds.
7.6.4 Stability and transient performance
This drive is relatively easy to control because the convertor frequency is locked
to the motor rotation and the flux conditions in the motor are controlled
separately with the field controller. The result is that stable performance is not
difficult to achieve.
If a supply voltage or frequency disturbance occurs with this drive there is
normally no noticeable effect on the motor. As long as the circuit current is
maintained the motor side of the system will remain unchanged and the fast
current control on the supply convertor, and the DC link reactor both prevent
current change occurring.
The six step synchro-convertor system for synchronous motors
273
A load change requiring an adjustment in torque and current cannot be
achieved instantly with this drive because of the presence of the DC reactor and
so some speed disturbance can occur in this instance. The size of the DC link
reactor is therefore critical to the transient performance achieveable and faster
performance requires a low inductance reactor.
Because of the effects of the stator current on the field flux of the motor as
explained in the previous sections one may expect that changes in the stator
current will also result in a change in air gap flux which would temporarily affect
the motor torque. This may occur with slip ring motors where the current in the
field winding is prevented from changing rapidly but in most brushless motors
there is an instantaneous reaction to the potential air gap flux change which
suddenly alters the field current to prevent that change. As long as the flux
control system is able to respond within the time constant of the induced
transient effect the motor flux will be maintained during the change in stator
current.
The DC link reactor is therefore the dominant feature in the transient
response of this drive.
/uu "
motor current lines
/
600 - 100
1
^ ^ A
907.^
500
I
" 80%
90%
/
, 80°/•
^-/
"""^A
7o # ^^^y^
400
- 60%
— ^^V /\ \ '
/
N
v
/
^70%
V
x
X
/
/
50%
. 40%
40%^^/^
o
^
/ \
^
100
100
200
unco
\
300
200 - 30%/^
motor speed
lines
60
\
%
\
J
rl--i—-1
1.
i
.
1.1
300
-—•'
400 500
600
reactive KVAR
30%
^ ^ 20%
^-107
700 800
Fig. 7.20 Synchro-t ?onvertor input power factor
7.6.5 Supply power factor
From this point of view this drive is similar to a DC motor drive in that the
power factor is proportional to speed. In fact it is proportional to the DC link
voltage and it is caused by the phase control of the supply side convertor. The
power factor is therefore affected also by the values of the supply and the motor
voltages chosen for the drive. If the motor voltage is significantly lower than the
274
The six step synchro-convenor system for synchronous motors
supply voltage then the DC voltage will never reach its full potential and this will
mean a low input power factor. The most practical arrangement for this drive
is to use the same value of motor voltage as the supply voltage (even if the
frequencies differ) and Fig. 7.20 shows the displacement factor performance
which would then result. From this figure it is possible to find the supply vector
and displacement factor corresponding to any operating condition. The circular
lines are the lines of circuit current, and therefore motor torque, and the radial
lines indicate the speeds. The vector OB is that for 50 per cent speed and 70 per
cent current, and therefore 70 per cent torque. OA is the vector for rated
operating conditions.
7.6.6 Torque pulsations
The square wave currents fed to the motor cause the stator MMF to step around
the air gap rather than rotate smoothly and the result is that the torque
generated in the motor is not steady but it contains a ripple component related
to the stepping speed. In a normal 3 phase motor with a six pulse motor
convertor the stepping frequency is six times the motor frequency. The magnitude of this ripple is directly dependent on the level of current flowing and it is
altered by the power factor of the current.
Chapter 8
The current source inverter for
the capacitor self-excited
induction motor
8.1 Introduction
This system is a combination of the two previous current source systems
explained in Chapters 6 and 7. It is a system which enables induction motors to
be controlled by a naturally commutated convertor, hence making it possible for
large power and high voltage drives to be produced. It can be considered as a
synchro-convertor for use with induction motors or as a load-commutated inverter
for induction motors.
It is a relatively new drive system which has only been produced in practical
form since 1984. However, significant interest has been shown in it since then
and it has now been used for a large number of high power drives. The system
has not yet attracted a simple generic description but has been referred to as the
high power induction motor drive.
The interest in this system is caused mainly by the fact that it can use
convertor grade thyristors throughout and these can be readily used in series
operation to achieve high voltages and powers. An additional benefit of the
scheme is that the motor currents and voltages under normal running conditions
contain only a small harmonic content and hence the drive can be used with
existingfixedspeed motors without derating to extend its capability by varying
its speed.
8.2 Principles of operation
During Chapter 7 it was regularly mentioned that that system could only be
used with synchronous motors because they are capable of generating voltages
which can be used to assist the convertor commutation. If an induction motor
is used with the synchro-convertor drive it is not possible to obtain magnetisation of the motor and hence it is impossible to obtain the necessary generated
voltages. Another way of looking as it is to appreciate that an induction motor
requires a lagging current to magnetise the core whereas a naturally commutated
motor convertor can only operate with a leading power factor current.
276
Current source inverter for capacitor self-excited induction motor
The main principle of this drive is to connect a large capacitor in parallel with
the motor so that together they require a leading power factor current. The
capacitor is therefore able to ensure that the motor remains magnetised and that
it can produce generated voltages which can assist in the switching of a naturally
commutated inverter.
Fig. 8.1 This shows a 1,200 KW, current source capacitor self-excited drive equipment for
varying the speed of an existing 3,300 volt, AC induction motor. The central cubicle
contains rectifiers and inverter thyristor bridge circuits. The left hand cubicle contains
voltage suppression and commutating circuit components and the electronic controls
and auxiliaries are contained in the right hand cubicle. (G.E.C. Industrial Controls, Ltd.)
Such a system will only operate in this way if the motor is already running and
if the motor/capacitor combination is operating at a sufficient frequency to
ensure the necessary resonant action between the capacitor and the motor
inductance. It is therefore necessary to operate the drive in a different way to
enable it to be started up and brought up to a speed which will ensure that
natural commutation operation can continue. This is usually done by providing
the inverter with a means of forced commutation which can be used at the lower
speeds and a method of changing over to natural commutation whenever
possible.
The basic power circuit diagram covering the principles of this scheme is
Current source inverter for capacitor self-excited induction motor
277
shown in Fig. 8.2. The motor convertor is a simple thyristor bridge suitable for
natural commutation. The motor capacitor is of large KVA — comparable with
the motor and the commutation circuit is shown across the DC link. The supply
convertor is a conventional mains commutated bridge and the DC link reactor
keeps the DC link current smooth and continuous and isolates the two convertors
from each other.
supply side
convertor
D.C.
reactor
motor
capacitor
motor side
convertor
•Idei
V
V
m
Acap
Vdci
induction
motor
Fig. 8.2 The current source convertor for the capacitor self-excited induction motor
The commutation circuit can be any of a variety of designs (see Section 8.4.2)
but in all cases the principle is to use an arrangement which will allow the
current flowing in the motor convertor to be temporarily diverted into the
commutation circuit every time a change in firing pattern of the motor convertor
is required.
In order to ensure continuous current in the DC link reactor the commutation
circuit has to be arranged so that it can carry the DC link current whenever the
motor convertor thyristors are being changed over. During this period any
currentflowingin the motor will continue toflowwith the capacitors providing
a path for this current.
In this system therefore the current is always maintained in the DC link
reactor and there is no period when the supply convertor is used to switch the
circuit current on and off.
At the higher end of the speed range the increase in frequency causes the
capacitors to become more dominant and the power factor of the motor/capacitor
combination to become more leading. At some point the current demanded by
this load will become sufficiently leading to allow natural commutation to occur
in the motor convertor. At this point the forced commutation circuit on the DC
link can be switched off and the drive will be running in a very similar way to
the synchro-convertor explained in Chapter 7.
Again the firing of the motor convertor has to be tied very closely to the
generated voltages produced in the motor but because the motor is now an
induction motor its speed will not be exactly related to the stator frequency. The
278
Current source inverter for capacitor self-excited induction motor
motor speed will vary with load and will be related to the stator field rotational
speed by the slip.
One important difference between this system and the other current source
drives is that the motor current is not directly related to the convertor and DC
link currents. It is the result of both the convertor current and the capacitor
current. The result is that the motor current is not quasi-square wave in shape
but it tends to be sinusoidal containing less of the harmonics from the convertor.
The currents in the capacitor will depend on the voltage occurring on the
motor and on the frequency of operation. In addition the motor voltage will
depend on the air gap flux level and therefore on the value of the magnetising
MMF which is dictated by the motor current. Hence we have a relatively
complicated interaction between the motor, capacitor and convertor currents.
The objective of the control system is to ensure stable operating conditions for
this motor system and the setting of the level of motor flux and magnetisation
is crucial to this.
8.2.1 High speed running
Natural commutation in the motor convertor can only be ensured if the convertor
current is at an appropriate leading power factor compared to the voltage sine
waves generated by the motor/capacitor combination. The motor on the other
hand can only be correctly magnetised if its current is lagging its generated
voltage. The capacitor is there to make up the difference between these two
essential requirements.
no-load motor current
Fig. 8.3 Vector diagram at high speed
The vector diagram in Fig. 8.3 shows how this is achieved. The curve AB in
the current locus for an induction motor operated at its correct magnetising
condition, OA represents the current vector at no load and OB that at high
Current source inverter for capacitor self-excited induction motor
279
torque. If the converter is to be naturally commutated by the motor voltage
the convertor current must lead Vm by an angle as shown, OC representing
convertor current. Under a high torque condition OC represents the convertor
current, OB the motor current and therefore BC must represent the capacitor
current.
The capacitor and motor currents cannot however be controlled directly and
the aim of the drive is to control the length of the convertor current vector and
its angle Beta to ensure that the correct speed and frequency is produced and
to ensure that the correct magnetising conditions in the motor are maintained.
In order to ensure natural commutation Beta has to exceed a minimum level
related to the period of current transfer and the time for the thyristors to recover
their blocking capability. Let the critical Beta at this speed be Beta 1 as shown.
The length of the capacitor vector is dependent on both the frequency and
voltage occurring on the motor and if we assume these remain constant then the
length BC must remain constant. As the load torque requirement changes
therefore the convertor current vector has to follow the locus CD if the motor
current is to be maintained on AB. The no load convertor vector is therefore
represented by OD with AD being the capacitor current under this condition.
As the speed is reduced so the frequency and motor voltage will reduce and
hence the length of the capacitor vector reduces on a square law basis. At high
torque the convertor vector could move to OP while still in natural commutation
but this point indicates the limit. The length BP therefore indicates the limit of
the natural commutation method of operation. At light load however the
convertor vector can move to OQ and hence AQ gives the capacitor vector and
hence the speed which can be tolerated. The area of high speed running with
natural commutation is therefore limited by the size of the capacitor and
Fig. 8.4 has been drawn for a typical case where the capacitor current under
rated conditions will be similar to the motor current. This shows that if operation
below these critical speeds is required then other means of commutation are
needed.
Although the area of natural commutation could be extended downwards in
speed byfittinga larger capacitor it is clear that the forced commutation method
chosen has to be capable of operation over a wide speed range from zero to at
least 50 per cent speed. Over this range the method used with the synchroconvertor of turning off the current using the supply convertor is not suitable
and an alternative approach is usually used.
8.2.2 Lower speed running
When a forced commutation method is applied to the motor convertor it
removes any limitations regarding the firing angle of the thyristors and the
convertor vector in Fig. 8.3 can be allowed to take up any Beta angle required
to maintain the correct balance. Negative as well as positive Beta angles are then
acceptable.
A variety of commutation methods can be used to switch the motor convertor
280
Current source inverter for capacitor self-excited induction motor
thyristors but they are all by-pass type systems which temporarily divert the DC
link reactor current out of the motor convertor to allow switching to take place.
Some of the alternatives are discussed in Section 8.4.2.
100-
natural commutation area
50
forced commutation area
J_
50
100
percent torque
Fig. 8.4 Range of commutation methods
The current in the capacitor will be very low at low speeds because it is
affected by both frequency and voltage and hence the capacitor can be neglected
as far as starting conditions are concerned. Initial starting is therefore very
similar to that of the current source inverter described in Chapter 6. A circuit
current is set up and the motor convertor is clocked round at the appropriate
slip frequency. The motor will start to turn and after a short time a motor
generated voltage will be detectable so that an estimate offluxcan be made using
volts divided by frequency, to ensure that sufficient current is being applied to
maintain magnetisation.
Under this condition the motor currents are almost equal to the convertor
currents and hence they are quasi-square in shape, similar to the other current
source systems.
As the speed increases the voltage and frequency will increase so that the
capacitor current now starts to become significant. The operation can then best
be appreciated by again looking at the motor convertor vector diagram. Fig. 8.5
has been drawn to show the lower speed running conditions. AS in Fig. 8.3 AB
represents the motor current locus but now A is now on the base line because
Current source inverter for capacitor self-excited induction motor
281
motor losses will be very small at low speed. The curves XY represent the
corresponding loci of the motor current plus capacitor current vectors at different frequencies and speeds of operation. A typical point of operation in the
forced commutation range is indicated by the triangle of currents OMC where
OM equals motor current, OC the convertor current and MC the capacitor
current. Another lower speed condition is shown as triangle O M C The
limiting condition is represented by the triangle OBC where OB represents the
rated motor current and OC" is the convertor current at the limiting angle
Beta 1 above which natural commutation would be used.
»\
Fig. 8.5 Forced commutation vector diagram
Therefore in the low speed operating mode the convertor current vector has
to be able to be placed anywhere in the OABC' area. At low speeds it will be
required to follow the locus AB and as speed increases the locus will follow one
of the other parallel curves XY moving to the left as speed increases.
The conditions I have just described in fact refer to a drive having a constant
torque with speed capability because the vertical direction on this figure is in
phase current which is proportional to torque (if constant motorfluxis assumed).
If full torque is not required at low speeds then the area required to be covered
by the convertor vector is reduced; OAP is the area appropriate to a typical fan
drive where torque is proportional to speed squared.
8.3 Detailed analysis of the system
As with the other current source systems the circuit currentflowsthroughout the
circuit, the supply convertor producing a smooth DC current which is chopped
282
Current source inverter for capacitor self-excited induction motor
up to produce quasi-square type current waveforms into the load which, in this
case, consists of the induction motor and the capacitor in parallel. During the
forced commutation of the motor convertor the current is by-passed around the
convertor and the motor current continues to circulate in the capacitor. During
natural commutation the conditions are almost identical to the synchro-drive
with all the DC link current flowing in the load circuit.
In this system, however, the motor current is not so easy to decide upon due
to the presence of the capacitor. All one can say initially with certainty is that
the convertor current plus the motor current plus the capacitor current equal
zero at all instants.
The voltage on the motor is decided by the amount offluxin the air gap and
hence by the magnetising component of the current and this voltage has a direct
bearing on the capacitor currents. As with all convertor driven induction motors
the motor induced voltage wave is always sinusoidal in spite of any harmonics
flowing in the motor.
high torque
high speed
increasing
torque
low torque
high speed
potential range
of beta angle
• 90
to
-90
Fig. 8.6 The full range of operation
It is the convertor's job to maintain the motor conditions in their optimum
state as far as possible at all times and from the previous vector diagrams it will
be clear that control over the Beta firing angle of the motor convertor is very
important to this control. In general the Beta angle is used to control the level
Current source inverter for capacitor self-excited induction motor
capacitor
283
motor
Z2
zc
Zl
zm
Z2'
Capacitor
impedance
Stator
impedance
Magnetising
impedance
Rotor
impedance
Rated conditions
full load 50 Hz
2-4
0-2
8-0
2-5
Low speed
high torque 5 Hz
24
01
0-8
0-25
5th Harmonic at
high speed
0-5
0-6
400
10
5th Harmonic at
low speed
4-8
015
40
1-25
All values
in OHMS
Fig. 8.7 System harmonic impedances
of magnetisation in the machine correct after allowing for the capacitor current.
Hence in this drive the Beta angle changes over quite a wide range to cope with
different operating conditions. The vector diagram, Fig. 8.6 shows a combination
of the forced and natural commutation conditions showing that, to cover the
284
Current source inverter for capacitor self-excited induction motor
full range of current and speed, the Beta angle variation will be almost 180
degrees, 90 degrees either side of the voltage vector. At low speed the capacitor
current will be small and the convertor current will be to the right of the voltage
vector and at high speed with a large capacitor current the convertor vector will
be to the left. The effective power factor of the motor convertor is therefore
continually changing in response to load and speed requirements.
The conditions in the motor at high speed are quite different from the
situation at low speed. At low speed the capacitor current is relatively low and
hence the convertor and motor currents are very similar to each other. At high
speed the capacitor current will be comparable with the motor current and
impedance of the capacitor is relatively low so that the harmonics tend to flow
in the capacitor rather than the motor. At low speeds the motor current will tend
towards the quasi-square wave shape of the convertor current whereas at high
speed the motor current is quite close to sinusoidal as the harmonics are diverted
into the capacitor. Fig. 8.7 shows the comparable relationship which occurs
between the impedances in the single phase equivalent circuit of a typical
100 KW induction motor with a capacitor across its terminals at high and low
speed and under fifth harmonic conditions. This clearly shows that at high speed
whereas the fundamental current will divide reasonably equally between the
motor and capacitor, the fifth harmonic current will mainly flow in the
capacitor. At low speed however the majority of convertor current will in fact
flow in the motor.
83.1 Circuit waveforms
The waveforms occurring in the circuit depend on the speed at which the drive
is assumed to be running.
Low speed conditions
At low speeds the motor capacitor can be neglected and it can be assumed that
all the convertor current flows in the motor. Also the commutation time,
whatever commutation circuit is used, will be a relatively small period in the
cycle and it can initially be ignored in assessing the circuit waveforms. During
commutation the presence of the motor capacitor allows the currents to change
relatively rapidly and hence the motor convertor current waveforms tend to be
very square. What happens is that the initial sharp rise of current flows in the
capacitor as the motor current builds up against the motor leakage inductance.
The induced voltage in the induction motor is usually very close to sinusoidal
and the terminal voltage of the motor will be only very slightly distorted from
this due to the square currents flowing in the leakage reactance of the stator.
Fig. 8.8 shows the low speed waveforms associated with the motor neglecting
the transient effects which may be caused by the commutation circuit.
The convertor currents will be sharply quasi-square with current pulses
slightly less than 120 degrees long due to the short period of commutation. The
top of the current pulses will normally reflect the DC link current ripple but I
Current source inverter for capacitor self-excited induction motor
285
have assumed it to be very small for the present. The motor currents are similar
but with a rounding off of the waveforms caused by the motor inductance. The
difference current between these two currents is what flows in the capacitor
— just a series of sharp pulses occurring at the points of commutation. The
motor phase voltage waveform shows a small amount of distortion due to the
currents changing.
convertor currents
u
j
r
LJ
Fig. 8.8 Low speed motor waveforms
High speed conditions
When the drive is running at high speed with natural commutation the capacitor
becomes the dominating influence. The currents in the motor convertor are still
square in shape but now the commutation period is negligible because of the
presence of the motor capacitor. The harmonics in the convertor flow mainly
into the capacitor, the motor current being a relatively good sine wave. The
motor terminal voltage will therefore also be close to sinusoidal with very little
distortion.
Fig. 8.9 shows that because the convertor and motor currents are now phase
286
Current source inverter for capacitor self-excited induction motor
displaced by a relatively large angle, the capacitor current is large with sudden
changes of level when the commutation switchings occur. In the case shown the
peak value of the capacitor current is approximately equal to the sum of the
peak of the motor current plus the DC link current.
• motor voltage
capacitor
current
convertor
current
Fig. 8.9 High speed naturally commutated waveforms
8.3.2 The motor vector diagram
As the performance of the motor is dependent on the relationship between the
fundamental values of current and voltage it is useful to study this in a little
more detail.
Fig. 8.10 shows the single phase equivalent circuit of the motor and the
capacitor and the vector diagram to go with it.
The induced voltage El produces a value of rotor equivalent current 12 which
depends on the load which alters the value of the slip, SI. Under the normal
conditions of air gap flux the slip will be low, making R2/S1 dominant in the
rotor current, making 12 almost in phase with El. The equivalent stator current
II is the vectorial sum of 12, the iron loss current IL and the magnetising current
Imag.
The stator current causes voltage drops II x Rl and II x XI thus giving
the value of the motor terminal voltage VI.
The capacitor current will lead the terminal voltage vector VI by 90 degrees
and the vectorial sum of the motor and capacitor currents will result in the
convertor current vector Ic.
The angle between the motor terminal voltage VI and the stator current II
gives the motor power factor and the effective convertor power factor is given
by the angle between Ic and VI, i.e. cos (Beta-Delta).
Fig. 8.10 has been drawn for a high speed high torque operating condition
where the capacitor current is large resulting in a positive value of Beta. The
vector diagram under other conditions will differ from this. At a lower torque
the vertical components of the currents will reduce and at lower speeds voltages
will reduce and the capacitor current will reduce due to both the voltage and
Current source inverter for capacitor self-excited induction motor
287
frequency reduction. Fig. 8.11 has been drawn showing in (a) the same chart as
Fig. 8.10, in (b) at the same speed but at half the load torque, in (c) at the same
torque as (a.) but at half speed and in (d) at half torque and half speed. In all
cases the optimum flux conditions in the motor are assumed to be maintained.
11
u
R1
X1
TXT7
1 T V I!
L2'
X2'
12
I mag
V1
-L r
El
V1
11X1
I cap
Fig. 8.10 The vector diagram
These figures show that the currents and phase angles vary with speed and
load over quite a wide range. In many practical cases however the speed and
load are related in some way. If the load is a fan or pump the load torque will
increase with speed on a square law basis whereas a mixer or extruder may have
the same torque to work against at any speed. Hence in these cases the range
of variation is somewhat less. The graphs of Fig. 8.12 show how the magnitudes
of the convertor and capacitor currents and the Beta angle vary with speed in
practical drives. The capacitor current always varies with the square of frequency
255
Current source inverter for capacitor self-excited induction motor
and therefore speed and the convertor currents and Beta angles depend on the
characteristics of the load.
(a) high speed
high torque
(b) high speed
50% torque
V1
V1
E1
(c)
50V. speed
1007. torque
(d)
50% speed
50% torque
V1
Fig. 8.11 Vector diagram variation
8.3.3 Relationships and equations
From the convertor point of view the relationships in this system are very similar
to those of the synchro-convertor described in Chapter 7. The main difference
is the wide variation of the Beta angle in this system which results from the
presence of the motor capacitor and the necessity to control and magnetisation
of the motor.
The magnitudes of the convertor currents are similarly related in that:
The input supply line current Is is related to the DC link current by
Is = Idc x V2/V3 amps RMS
Is = 0-816 x Idc amps RMS
Current source inverter for capacitor self-excited induction motor
289
The DC link voltage is dictated by the firing angle of the supply side convertor
(Alpha) and neglecting convertor supply resistance and reactance.
Vdc = 1-35 x Vs x COS (Alpha)
(2)
However there will be a small drop in voltage due to the circuit current
flowing through the resistance and reactance of the supply system and supply
beta angle -degrees
90
60
30
0
-3 0
torque
constant
torque ex speed l
-60
-90
/
/
convertor I c
current
torque constant
/
^
\
/
^
;
t
capacitor
current leap
/
/
^y
Fig. 8.12 Variation with load
i
I
/
/
i
50
percent speed
/
/
/
/
•
/
/
/
I
/
•
/
torque a
speed 2
I
I
100
290
Current source inverter for capacitor self-excited induction motor
convertor and an allowance should be made — say 3 per cent,
Vdc =
1-35 x Vs x (COS (Alpha) - -03 x Is/Isr)
(3)
where Isr = the rated value of the supply line current.
The power being passed across the DC link from supply we will assume to be
Pdc where:
Pdc = Idc x Vdc watts.
(4)
When the system is working in the forced commutated mode, however, the
DC current and voltage may be modified in some way by the commutation
circuit. Most of the commutation circuits employed cause the DC link current
to be temporarily diverted across the DC link in which case the mean level of
current passed to the motor convertor will be less than Idc. Fig. 8.8 shows a
typical example in this respect where a 'notch' of current is by-passed across the
link six times per motor frequency cycle. In such a case Idcl (refer Fig. 8.2) will
be lower than Idc in proportion to the width of the 'notches'. Then
_A t
6 x Gamma
T«
Idcl = Idc x
——
amps mean
...
(5)
where Gamma is the commutation angle, the 'notch' width. The DC voltage
applied to the motor convertor will also be affected by the commutation circuit
but as most commutation circuits do not cause any significant power losses then
the DC link power remains the same on both sides of the commutation circuit.
i.e.
Vdc x Idc = Vdcl x Idcl
(6)
Clearly when the drive is running in its naturally commutated mode the
commutation circuit has no effect, i.e.
Vdcl = Vdc and
Idcl = Idc
The current flowing in the output lines from the motor convertor (Ic) will also
be affected by the commutation circuit when it is in operation.
Under naturally commutated conditions the convertor output current will be
quasi-square as in the synchro-convertor and then
Ic = Idc x v^/73 amps RMS
or
Ic = Idc x -816 x 1/105
= Idc x -778 amps fundamental
(7)
When the forced commutation circuit is in operation the 'notches' will alter
the relationship between the output current and the DC current. In the case of
a commutation circuit which fully by-passes the DC current for an angle
Gamma at each commutation, then, the convertor output current will be given
Current source inverter for capacitor self-excited induction motor
291
by:
Ic = Idc x /
V
—-- amps RMS
K
90
x (1 — — ) amps fundamental
(7a)
n
\
60/
Referring now to the motor side circuit and vector diagram of Fig. 8.10, the
only parameter on this diagram which we know is Ic (see equation (7)). However, assuming a steady state of operating condition the DC link voltage and the
motor voltage must be related by the firing angle of the motor convertor Beta,
i.e. neglecting commutation effects.
Vdcl = VI x y/3 x 1-35 x COS (Beta-Delta)
(8)
where VI = motor terminal voltage per phase.
The value of the fundamental value of the capacitor current leap can be
established from the motor terminal voltage and the frequency.
leap = Vl/(2 x PI x F x C)
(9)
where F is the frequency and C the value of the capacitor in farads per phase.
The values of the induced voltage and the motor terminal voltage are dependent on the air gapfluxand hence on the magnetising current Imag and this can
be found once the rotor current 12 has been established. 12 is load dependent and
is related to the induced voltage by:
12 = E1/Z2
(10)
Z2 = 7 ( 2 x jr x F x L2)2 + (R2/S1)2
(11)
where
and
TAXWA *>\
2 x 7T x F x L2 x SI
TAN(An2) =
—
The magnetising current can then be found from
(12)
Imag = leap x COS (Delta) - 12 x SIN (An2) - Ic x SIN (Beta)
(13)
Reference to Chapter 1 will show that the magnetising current and induced
voltage will be related by an equation similar to:
El = Vsat x ^- x [1 - 2-71(~l 33xImag/Isat)]
(14)
It is clearly best to use a computer to solve all of these equations to establish
the values of all parameters under steady state condition. In practice however,
292
Current source inverter for capacitor self-excited induction motor
a number of assumptions can be made to arrive at approximate results much
quicker and this clearly helps in understanding. These assumptions are:
1) If the motor is correctly magnetised the value of slip will always be
small and this leads to angle An2 being negligible.
2) The voltage drops in the stator winding are relatively small and can
be neglected in arriving at approximate results. Angle Delta is then
negligible.
The above equations then become:
Vdc = VI x V3 x 1-35 x COS (Beta)
(8a)
Imag = leap - Ic x SIN (Beta)
(13a)
and equation (14) will give VI. The speed of the motor will be given by
S =
120 x ™ x (1 - SI) RPM
(15)
The power into the motor can be found from
Pm = 3 x Ic x VI x COS (Beta-Delta)
(16)
and the motor stator current II can be found from the convertor and capacitor
currents.
8.3.4 Examples of calculations
1) Calculation for the motor capacitor
Question
A 132 KW delta connected induction motor is normally supplied from a 415
volt, 50 hertz supply and under rated flux conditions it requires a magnetising
current of 25 amps per phase and operates at an efficiency of 97 per cent. What
is the minimum value of motor capacitor per phase required to ensure that
natural commutation is achieved when this motor is fed from a convertor of this
type when operating at rated motor current and torque and 40 hertz assuming
that natural commutation requires a Beta angle of 15 degrees at this frequency?
Answer
Referring to the vector diagram Fig. 8.13(a).
Under rated motor conditions the vertical component of the motor phase
current will give the power into the motor
132 x 1000/97 = 3 x Ip x 415
therefore
Ip = (132 x 1000)/(-97 x 3 x 415)
=
109-3 amps.
Current source inverter for capacitor self-excited induction motor
293
This must also equal the inphase value of the convertor current and therefore
Ic = 109-3/COS 15 = 113.2 amps fundamental.
The capacitor current to ensure natural commutation therefore must exceed
Ic x SIN 15 + Imag.
from equation (13a), i.e.
leap > 54-3 amps.
At a frequency of 40 hertz the motor voltage will be approximately 332 volts
per phase therefore the impedance of the capacitor must equal
Zcap =
332
54-3
= 611 ohms.
= 1/(2 x PI x F x C)
therefore
C = 1/(2 x PI x 40 x 6-11) farads
= 651 microfarads per phase.
El
VI
\15°
I mag
2 5 amps.
a
I cap
El
VI
Imag
= 25
Fig. 8.13 Example vector diagrams
294
Current source inverter for capacitor self-excited induction motor
2) Convenor conditions
Question
What would be the input current and its displacement factor (COS (Alpha))
be under rated motor operating conditions for the above motor/capacitor
combination when fed from a current source inverter of this type if the supply
mains voltage was 460 volts, 3 phase, 50 hertz.
Answer
Referring to Fig. 8.13(b). As the motor is to be operating at rated torque
conditions then the vertical component of current will still be 109-3 amps.
However now the capacitor current has increased because the motor is to be
operated at 50 hertz, therefore from equation (9)
leap = 415/(2 x PI x 50 x 651 x 10~6)
= 84-9 amps.
Therefore the reactive component of Ic must equal
84-9 - 25 = 59-9 amps.
Therefore
Beta = ATN (59-9/109-3) = 28-7 degrees,
and
Ic = 124-6 amps fundamental per phase.
As the motor is Delta connected the line current out of the motor convertor will
equal
Ic (line) = 215-8 amps fundamental,
and from equation (7) this must lead to a DC link current of
Idc = Ic (line)/-778
= 277-6 amps mean
If we ignore the convertor power losses then equation (4) giving the power input
to the motor will decide the DC link voltage
Vdc = Pdc/Idc = 132,000/(-97 x 277-6) = 490-2 volts.
from equation (1)
Is = 277-6 x -816 = 226-6 amps RMS.
Current source inverter for capacitor self-excited induction motor
295
and from equation (3)
(COS (Alpha) - -03) = 490-2/(1-35 x 460)
COS (Alpha) = -819.
8.4 Practical circuit design considerations
This drive has many similarities with the other current source drives and
particularly with the synchro-convertor described in Chapter 7. This system uses
conventional thyristors for the convertor bridges but the characteristics of the
forced commutation circuit may affect the requirements of the motor convertor
thyristors. The presence of a large capacitor connected to the motor also has an
influence under fault conditions as well as during normal operation. The fact
that the motor is an induction motor is also important in that it can loose its
magnetisation if the reactive component of its current is reduced, and this can
be done by a change in phase of the motor convertor current.
8.4.1 Protection
In general overcurrent protection follows the same pattern as with the synchroconvertor with the DC link reactor limiting the rate of rise of fault currents and
allowing the convertors to control the faults statically to prevent circuit damage.
The same means of protection are also used and reference should be made to
Section 7.4 before proceeding. This drive is normally used in the motoring mode
and hence the fault conditions which can occur under regeneration are not
usually important to this drive.
The different feature of this drive is the fact that magnetisation of the motor
can be lost relatively easily. As will be seen from Section 8.5 thefluxin the motor
is maintained by the convertor current and its angle, and at the higher speeds
continuous control over thefluxis essential as the motor/capacitor combination
would otherwise be unstable. The result is that any maloperation of the convertor
system can easily result in loss of flux control and either no flux or maximum
flux. Neither of these effects, however, occur instantly and the normal methods
of current control, i.e. current limit and fast phase back can usually ensure that
the circuit current is under control. If excessive flux occurs this could cause
overheating in the motor if it were allowed to persist; at high speeds it can also
cause an excessive voltage to be produced which may affect the insulation. The
level of voltage which can occur depends on the magnetisation curve of the
motor and the ratio between saturation flux and normal flux.
This problem of potential overfluxing and over volting the motor is an
important one because it can be caused by the convertor supply being switched
off or by the convertor turning the current off. When the capacitor current is
large the immediate result of turning the convertor current off is to push the
motor flux much higher and the capacitor energy takes some time to dissipate
296
Current source inverter for capacitor self-excited induction motor
so that the condition can persist for a little while. The motor inductance and the
capacitor will resonate for a few cycles as the capacitor energy dies away and
during this period the motorfluxwill gradually reduce from its initial high level.
If this condition is likely to cause damage due to the temporary excessive voltage
then it may be necessary to disconnect the motor from the capacitor or to
introduce some additional resistance into the motor/capacitor circuit to speed
up the dissipation of the capacitor energy.
8.4.2 Commutation methods
Although it is essential to have a forced commutation facility in the motor
convertor circuit to enable operation of this system at lower speeds to take place,
the detailed operation of this system is not greatly affected by the particular
method used to affect the necessary switching. Hence I have not dealt with this
part of the circuit up to now.
Clearly there is no technical reason why the switches used in the motor
convertor could not be transistors or gate turn off thyristors which would be
capable of switching the current by themselves. If this were done this drive will
work efficiently and successfully. However in this case it would probably be easier
and cheaper to use the current source inverter system described in Chapter 6, or
it may be even better to use a pulse width modulated voltage source inverter as
described in Chapter 5.
Up to now the main advantage which has been seen to be associated with this
capacitor self excited induction motor system is the fact that it can use convertor
grade conventional thyristors which can be used in series or parallel to enable
relatively large power ratings to be produced.
The commutation circuits which are used are therefore those which result in
relatively long turn-off times of between 300 and 600 microseconds.
The methods normally used are DC link commutation circuits which are
operated six times per output cycle to switch all the motor convertor thyristors
i.e. one commutation system for all six arms. There are two types of these, those
which rely on the flow of DC link current and those which are comparatively
independent. As it would be impossible to describe all the potential circuits
which could be used, only one example of each of these will be described in order
to show the principles involved.
The DC link current commutation circuit
In this circuit the DC link current is diverted from the motor convertor into the
commutation circuit and is used to charge up the commutation capacitor ready
for the next commutation.
Fig. 8.14 shows the example we are to study, the commutation capacitor is Cc
and the DC link reactor acts as the commutation inductance. Thyristors 7, 8, 9
and 10 are similar switches to those used in the arms of the motor convertor. The
sequence of operation of the circuit is as follows:
1) If wefirstassume that the current isflowingfrom the DC link through
Current source inverter for capacitor self-excited induction motor
297
thyristor 1, into the motor windings etc. and back via thyristor 2 and
we want to switch the current from thyristor 1 into thyristor 3. The
commutating capacitor will be already precharged in the direction
shown.
Idc
DC
reactor
motor
convertor
7 1
8^7
Cc \7 7
J
9jJ
no
i
A
B
C
to motor
and
capacitor
7j J
commutation
circuit
Fig. 8.14
The DC link current commutation circuit
2) As soon as thyristors 7 and 9 are fired the current in the motor
convertor will be immediately diverted into Cc, and if thefiringof the
motor convertor thyristors had been previously removed then they
would all turn off. This would not interrupt the motor current because
of the presence of the motor capacitor.
3) The current is allowed to continue to flow into the capacitor Cc
until its voltage reverses and until the next pair of motor convertor
thyristor arms are fired. The capacitor is normally allowed to reverse
its voltage to the same value as it was originally. Once this has been
achieved and all the motor convertor thyristors have fully recovered
the appropriate thyristors are fired in the motor convertor and the
current immediately returns toflowin the motor circuit and thyristors
7 and 9 turn off naturally.
4) The commutating capacitor is now charged correctly so that when it
is necessary to switch the current from 2 to 4 this can be achieved by
firing thyristors 8 and 10.
Fig. 8.15 shows the voltages and currents occurring in the commutation
circuit during this sequence. If the DC link inductance is sufficient to maintain
the current relatively constant during this period the capacitor charging will be
linear and the commutation time will be inversely proportional to the level of
currentflowingat the time. During the commutation period the voltage across
the motor side of the DC link is the same as that across the commutating
capacitor, the three pairs of series thyristor arms i.e. 1 and 4, 5 and 2, 3 and 6
therefore all see this capacitor voltage. The proportion of it which appears
across each thyristor depends on the magnitude of the sine wave voltages from
the motor at that instant. The turn off time which the motor convertor thyristors
have therefore depends on the value of the appropriate sine wave voltage at the
298
Current source inverter for capacitor self-excited induction motor
point of commutation. As the Beta angle can vary from nearly + 90 to - 90 a
considerable variation can occur in this turn off time. The maximum range of
this can be seen from Fig. 8.15. If the sine wave voltage is equal to half of the
commutating capacitor voltage then if commutation takes place when the
voltage on phase B is at its maximum negative peak then a turn off time of one
quarter of the total commutation time (Time 1) will occur on Thyristor 1 and
if the voltage is at its maximum positive peak then the turn off time will be three
quarters of the total commutation time (Time 2). Clearly the commutation
system has to be designed to allow a satisfactory time for thyristor turn off under
all conditions and the worst case will probably be with maximum current
flowing when the total commutation time is at minimum value.
range of
motor voltage
^inewaves
firing of
switches
3and2
Fig. 8.15
Commutation waveforms
At the end of commutation the charge has been fully reversed on the commutating capacitor and the next commutation is achieved by firing thyristors 8
and 10 after which a similar sequence takes place.
The independent commutation circuit
With this circuit the divert switch operates relatively independently of the actual
value of the load current and the circuit consists of additional switches and a
commutating reactor which cause similar levels of commutating capacitor current
toflowat all points of switching. Fig. 8.16 shows a system demonstrating these
Current source inverter for capacitor self-excited induction motor
ldc
motor
convertor
A
B
to motor
and capacitor
commutation
circuit
Fig. 8.16 The independent commutation
firing of
thyristors
7and9
circuit
firing of
thyristors
3 and 2
current in
thyristors
3 and 2
current in
commutating
capacitor C c
range of
motor voltagesinewaves
Fig. 8.17 Commutation waveforms
299
300
Current source inverter for capacitor self-excited induction motor
features. The commutating capacitor Cc has its own reactor Lc and two switches
8 and 9 to assist the reversal of charge. The sequence of operation is as follows:
1) When commutation of the main bridge is required the capacitor Cc
is at its full negative charge as shown. Commutation is initiated by
firing both thyristors 7 and 9. The firing of 7 causes the main bridge
current to be immediately diverted into the commutating capacitor
and thyristor 9 causes the capacitor to be short circuited via Lc.
2) The capacitor voltage reduces and eventually reverses due to the load
current and the Lc current.
3) The Lc current follows a half sine wave eventually coming to zero and
thyristor 9 turns off.
4) The reverse voltage to turn off thyristor 1 will be provided by the
initial charge on Cc and the turn off time allowed will be given by the
time the capacitor takes to discharge to the same voltage as line A.
5) Once Cc has reversed in voltage and thyristor 1 has fully recovered
then thyristors 3 and 2 can be fired to return the current to the main
bridge and the load.
6) The commutating capacitor is recharged byfiringthyristor 8 to cause
a reverse current limited only by the commutating reactor.
Fig. 8.17 shows the currents and voltages during commutation showing that
the capacitor current is now dominated by the current through Lc and its
voltage changes more sinusoidally.
In this circuit the load current causes the capacitor voltage to increase and
hence it is usual for the current through Lc to be much larger than the load
current so that the variation in capacitor voltage due to load current is not large.
As a result the size of the capacitor increases to a number of times that required
in the previous commutating circuit.
8.4.3 Factors affecting the specification of the main components of the circuit
The DC link reactor for this system is selected on a similar basis to that required
in the other current source circuits, i.e. by assessing the amount of ripple current
which would occur and by considering the requirements of protection. In the
case of this specific system the amount of ripple voltage occurring across the
reactor will be similar to that occurring in the circuit described in Chapter 6 and
it will be somewhat more than occurs with the synchro-convertor of Chapter 7.
The reason is that the Beta angle in these two induction motor circuits is allowed
to vary over quite a wide range whereas in the synchronous motor system it is
usual to keep the Beta angle relatively low and constant.
The motor used with this drive does not have to be special in any way. The
fact that the currentsflowingin the motor contain less harmonics than the other
circuits means that the motor conditions are generally good. The motor currents
at low speeds do revert to the quasi-square shape but normally the currents are
low at low speed and derating because of this is not usually required.
The motor capacitor can be a relatively standard type of power factor correction
Current source inverter for capacitor self-excited induction motor
301
capacitor as it is connected onto the motor terminals where the voltage is
relatively sinusoidal. It does however have to carry the harmonics which are
normally present in the convertor current at all times and these need to be
allowed for. In some cases part of the capacitor may be split off from the
remainder to use it as a harmonic filter by connecting resonant reactors in series
with the capacitor section. Obviously if this is done the main part of the
capacitor may not see much of the total harmonics.
8.5 Overall control methods
In this system there are again only three independently controllable variables
and all the other parameters will be the result of the specific combination of
these three variables. These three variables are:
1) The firing angle of the supply convertor.
2) The frequency of the motor convertor.
3) The firing angle of the motor convertor.
A beta
beta
Fig. 8.18 Motor/capacitor instability
The additional complication of this system is that the magnetisation of the
motor has to be continuously controlled by the above variables at all times. The
presence of the motor capacitor means that the motor cannot be left to look
after itself because the motor/capacitor combination is inherently unstable at
high speeds. This feature can best be shown with reference to Fig. 8.18 which
shows the vector diagram of the motor, capacitor and convertor currents related
to the motor voltage vector. If we assume the drive to be running under a
condition given by the current vectors OC, OM and MC being the convertor,
motor and capacitor currents respectively, a small reduction in Beta will cause
302
Current source inverter for capacitor self-excited induction motor
OC to move to OC and OM to OM. This means that the motor magnetising
current has been increased and this will cause the flux to increase so increasing
the terminal voltage. The result is that the capacitor current will therefore
increase causing the motor current to move to OM, this causes more magnetising
current and hence more volts etc., and so the motor soon goes into saturation.
And all of this is just caused by a small reduction in Beta. If Beta was however
increased the opposite can occur and very soon the flux has disappeared
completely.
The above assumes that the frequency is unchanged, but this may not be the
case as the original change in Beta also causes more torque and so does the
increased flux so the drive may well speed up and the frequency may also be
increased. This makes the situation even worse because the capacitor current is
dependent on both the terminal voltage and the frequency.
So it is easy to see that close control over thefluxis necessary at all times and
that this flux system should dominate the control scheme.
In the naturally commutated mode of operation it is necessary, as with the
synchro-convertor, to tie the motor convertor firing closely with the back emf
generated by the motor/capacitor in order to ensure that an adequate Beta angle
is maintained. Under this mode therefore the frequency is not a completely
independent variable (it is directly related to the motor speed) and the supply
convertor Alpha and motor convertor Beta angles have to look after the health
of the complete system.
8.5.1 Supply convertor control
As with the other current source drives the supply side convertor is used to
control the current in the system and hence the load torque and maybe speed.
Although the phase angle of the supply convertor controls the DC link voltage
it is the balance between the DC voltage on the supply side and the reflected
motor voltage on the motor side of the DC link reactor which dictates the
current. Hence control of Alpha can do this and can be arranged to respond to
changes in the voltage on the motor side of the reactor so as to maintain the
required level of current.
In this system the level of current required has to take account of the needs
of motor magnetisation as well as load torque. In fact, to control magnetisation
it is necessary to alter Ic x SIN (Beta) and to control torque the control
function is Ic x COS (Beta).
8.5.2 Motor convertor control
The system of control for this convertor may need to be different for low speed
forced commutation operation compared with that for high speed natural
commutation. At high speeds the frequency of the convertor may be intimately
tied to the motor rotation via a terminal voltage measurement whereas at lower
speeds it is possible to use a freerunning oscillator to decide the convertor
frequency.
Current source inverter for capacitor self-excited induction motor
303
The control over the motor convertor in this system is particularly complex
because it influences both the torque fed to the motor and the magnetisation of
the motor #t the same time. Thefluxin the motor depends on the reactive value
of the convertor current (Ic x SIN (Beta)) and the motor torque depends on
the active value of the convertor current, Ic x COS (Beta) and the motor flux.
It is usual however in this drive system to use the Beta control as a means of
flux control with some degree of interaction between the supply convertor and
motor convertor controls to allow for the interdependence of current and Beta.
8.5.3 Motor magnetisation control
The aim of this control is to maintain the air gapfluxin the motor at its designed
and optimum level. This does not mean that it is impossible to operate the motor
with a higher or lower flux level but if a lower level is used then the torque will
be reduced and there is a risk of loosing the flux altogether. If a higher level of
flux is adopted then the terminal voltage at high speed may be excessive. The
simplest means of assessing the air gap flux is to use a voltage divided by
frequency signal and this is normally used.
The factor which directly influences the motor magnetising current and
therefore thefluxis the difference between the capacitor current and the reactive
value of the convertor current, i.e. equation (13(a)).
Imag = leap - Ic x SIN (Beta)
Therefore the aim of the motor flux control scheme is to alter Ic x SIN (Beta)
in order to compensate for changes in the capacitor current so as to maintain
the magnetising current reasonably constant. This is one of the ways to control
the motor flux using a measurement of capacitor current as a feedback for a
Beta control scheme. The method is to effectively compute Beta from the above
equation
T> +
• /leap — Imag\
Beta = arcsine — £
—Ic
/
\
This method has the benefit of being able to control the value of Beta when it
is both positive and negative.
An alternative approach is to use a voltage divided by frequency measurement to indicate the air gap flux and to control the Beta angle of the motor
convertor on a closed loop basis.
Whichever method is used it is necessary to take account of the fact that Beta
values will be either side of the vertical i.e. positive and negative and that the
gain of the loop will vary depending on the angle of Beta at the time and the
value of the convertor current.
i.e.
8.5.4 Typical overall control scheme
The control scheme proposed in Fig. 8.19 shows the basic arrangements which
could be used for a drive of this type.
304
Current source inverter for capacitor self-excited induction motor
The scheme shows two alternative control arrangements one of which is used
for low speed operation (forced commutation) and the other for the high speed
(natural commutation) mode.
The supply convertor is used to control the circuit current at all times via the
current amplifier (1) which receives its current feedback measurement from the
output of the motor convertor (2), this is because the commutation circuit may
alter the relationship between this current and the supply or DC link currents.
supply
convertor
ACCB
low speed
voltage controlled
oscillator
voltage/
frequency
regulator
Fig. 8.19 Typical control scheme
When running at the lower speeds, the system is arranged to operate in a
similar manner to the current source inverter of Chapter 6, i.e. with a variable
frequency oscillator (3) directly driving the motor convertor. The magnetisation
of the motor is then maintained correct by controlling the supply convertor via
a voltage amplifier (4) set to control the motor terminal voltage. In this case the
Beta angle will automatically adjust itself depending on the load conditions and
the motor capacitor current.
Under high speed, natural commutation conditions, the motor convertor
needs to be tied more closely to the motor and the motor convertorfiringcircuit
(5) is now arranged to be synchronised to the motor terminal voltage rather than
the free-running oscillator. The control is also changed so that the motor
convertor Beta angle is used to set the motor flux level using the v/f regulator
(6). In order to improve the control in thisfluxloop, infeeds of capacitor current
and motor convertor current are introduced in accordance with the equation
(13(a)). In addition, the voltage amplifier feeding the supply convertor current
loop is changed into a speed amplifier by bringing in the frequency feedback.
Current source inverter for capacitor self-excited induction motor
305
Hence in this high speed mode the torque and speed are controlled via the
supply convertor and the motor convertor Beta angle is used to set the flux at
all times.
8.6 Performance
The main reason for this drive being developed was because of its ability to drive
large induction motors at variable speed. The drive can be produced at high
voltages and the conditions imposed on the motors are relatively good so that
the drive has found considerable use with existingfixedspeed motors. This drive
can be fitted in the supply cables to existing large motors thus avoiding the
disruption which may be caused by removing and replacing the motor with a
different one. One of its prime performance advantages therefore is the fact that
under the natural commutation operating conditions which occur at the high
speeds the motor currents and voltages are relatively sinusoidal with only a
small harmonic content.
From other points of view the drive gives reasonable performance characteristics. It can produce high motor torques over the complete speed range and
the overall efficiency is very good particularly at high speed when the forced
commutation system is not needed.
Being a current source drive it is seen and used as a general purpose drive
rather than one capable of sophisticated performance. It can be produced
in sizes from a few hundred KW up to tens of megawatts and at voltages from
380 volts 3 phase to 13-8 KV without difficulty. Its most common uses are for
driving large pumps, fans and compressors in heavy industry.
8.6.1 Motor current waveforms
As explained in section 8.3.1 the harmonics which of necessityflowin the motor
convertor are diverted into the motor capacitor under high speed operating
conditions of this drive. The result is that the current whichflowsin the motor
windings is relatively free of harmonics and hence it is able to produce a
smoothly rotating field in the motor.
The oscillogram of Fig. 8.20 shows a typical motor voltage and current
waveform under high speed conditions and this waveform contains less than
two per cent of harmonics which is very good for a variable speed drive system.
However as will be understood from section 8.3.1, the proportion of harmonics
in the motor current will increase as the speed and frequency reduces and at low
speeds the majority of the convertor current harmonics will appear in the motor.
This situation is no worse than the conditions which exist in the other current
source systems and it is rare for the low speed torque requirement to be high
enough for these harmonics to cause difficulties.
An important characteristic of this system is that the large capacitor connected
to the motor (which is mainly inductive) will cause a resonant condition to occur
306
Current source inverter for capacitor self-excited induction motor
and somewhere over the speed range its frequency can correspond to firing
frequencies in the convertor. This causes the resonant frequency to be increased
in magnitude and it can appear in the motor current waveforms. In many cases
therefore the motor capacitor is combined with a harmonic filter to attenuate
this resonant condition.
line voltage
420V R.M.S.
line current
140 amps
R.M.S.
Fig. 8.20 Motor line current and voltage waveforms
8.6.2 Torque/speed capability
In general the torque produced by an induction motor can be up to twice the
normal torque as long as sufficient current is available from the convertor to
supply it.
The presence of the motor capacitor means that the transfer of current from
one phase to another during natural commutation can be very quick because it
is not now limited by the leakage inductance of the motor. The result is that the
level of current which can be commutated by the motor convertor can be well
above the rated value without causing commutation failures to occur. Under
natural commutation conditions therefore there is the capability to produce
torques well in excess of the rated value if this is needed.
Under forced commutation conditions, however, th maximum current which
can be carried will be limited by the capabilities of the commutation circuits. The
commutation circuits which depend on load current directly to charge the
commutation capacitors results in a main thyristor turn off time which is
inversely proportional to current and hence the more current, the less turn off
time. This limits the maximum current which can be commutated. The same
may be true of those commutation circuits which are more independent of the
normal DC link current: it depends on their detailed design. The circuit
explained in Section 8.4.2 and shown in Fig. 8.16 is likely to be able to cope with
a higher value of DC link current with only a small reduction in turn off time,
but it may be necessary to allow the commutation capacitor voltage to rise
above its normal maximum level when it occurs.
The torque capability at standstill and very low speeds, i.e. on starting, is
completely dependent on the initial charge which can be given to the commu-
Current source inverter for capacitor self-excited induction motor
307
tation capacitor. In this circuit the capacitor is initially charged from the DC
link via the link reactor and there is no difficulty in charging it to high values,
if required. Very good starting torque is therefore possible using this drive.
100
90
80
2
o
O
^
<D
70
60
I 50
>»
a
20
50
rated speed,%
100
Fig. 8.21 Typical input power factor
8.6.3 Supply power factor
As is usual in these circuits the supply power factor depends on the value of the
DC link voltage. The supply convertor angle Alpha has to be set to balance the
voltage coming back from the motor and this is approximately given by:
Vdc = 1-35 x Motor Voltage x COS (Beta)
The result is that the supply power factor is now dependent on both the value
of the motor voltage and the power factor angle of the motor convertor. Using
equation 2 and that above
1-35 x Vs x COS (Alpha) = 1-35 x Motor Voltage x COS (Beta)
i.e.
COS (Alpha) = Vm/Vs x COS (Beta)
The supply voltage Vs is usually fairly constant. The motor voltage is normally
arranged to vary in proportion to frequency. The Beta angle, as will be seen
from the many vector diagrams already drawn in this chapter, will vary widely
with load and speed conditions. The result is that a fairly complex relationship
exists between supply power factor and load condition as shown in the graphs
of Fig. 8.21. In general, the best power factor is proportional to speed, but its
value reduces with load, except at speeds which correspond to low angles of
Beta.
Chapter 9
The cycloconvertor
9.1 Introduction
The cycloconvertor is a direct frequency convertor without an intermediate DC
link which can convert power from one fixed frequency to a lower variable
frequency.
The cycloconvertor is a mains commutated system which has been known
about since the advent of grid controlled mercury arc rectifiers in the 1930's. The
system was then used extensively to produce 16f hertz power for traction
applications, from the 50 hertz mains supply. Since that time it has been used
for induction heating and for low frequency arc furnaces for slag refining. It has
also been used for motor drives since the late 1940's when it was used to supply
low frequency roller table drives in steel mills, using this time steel tank mercury
arc rectifiers. It has been used more widely since the advent of thyristor switches
and its most important application is for supplying large synchronous motors
driving low speed cement mill furnaces where units up to 8000 KW have been
built operating at up to 10 hertz. It has been and still is being used, though less
extensively, for a variety of low frequency applications from steel rolling mills
and tables, for mine hoist drives and for ship propulsion drives.
From this you will appreciate that the drive can be produced at any power
level and this is definitely one of its advantages. It invariably uses thyristors as
its switching elements and in normal applications there is no point in using more
sophisticated semiconductor devices such as transistors or gate turn off thyristors.
Over most of its range the cycloconvertor produces a reasonable sine wave
output leading to good motor performance, particularly at the lower frequencies.
Traditionally satisfactory performance has been understood to be available up
to approximately 40 per cent of the input frequency e.g. 20 hertz output from
a 50 hertz supply. Above this level the waveforms become more distorted due
to interaction between the mains and output frequencies and performance
progressively deteriorates. Much work has been done by researchers over the
years to find ways of improving this limitation of the cyclcoconvertor but there
is little evidence of their success in practical applications at reasonable power
levels.
The cycloconvertor
309
Another factor in its application is that because it is a direct frequency
convertor some of the output low frequency variations are reflected into the
mains supply system and these can cause interference with the operation of
another plant. However if the drive is fed from an appropriately powerful
supply system no adverse effects are experienced.
A.C. supply
bridge A
bridge B
A.C. supply
Fig. 9.1 A single phase cycloconvertor
310
The cycloconvertor
The cycloconvertor can be used to supply either a synchronous motor or an
induction motor and they will both operate entirely successfully. For relatively
straightforward applications like pumps and fans induction motors will normally
be used because precise control over the motor is not essential. If, however more
accurate control is required (perhaps because of a more dynamic load like a steel
mill) it may be preferable to use a synchronous motor. In this case because the
excitation of the motor can be controlled from a separate independent source,
the motor flux can be set more precisely allowing finer control over the torque
to be achieved. The synchronous motor also allows operation at a higher power
factor to be possible and this can increase the power output which can be
achieved from a specific set of transformers, convertors etc.
9.2 Principles of operation
The cycloconvertor is based on the principle of using an AC/DC thyristor
convertor circuit as described in Chapter 3 and then continuously varying the
firing points of the thyristors to achieve a low frequency AC output.
volts and current
* — positive
*
I-
volts and current
* — negative — •
I*
V-
voltage^
I-
B bridge B
rectifying
\
/
'
\
\
/
A bridge
rectifying
current
/
/
\
\
/
\
bridge B
inverting
«
t1
t
A bridge inverting
• volts
t
B bridge inverting
- current
bridge A
inverting
volts
1
Fig. 9.2 Satisfying had conditions
bridge A
rectifying
• current
bridge B
rectifying
I
t
B bridge inverting
•
The cycloconvertor
311
9.2.1 The fundamental principles
The convertor bridges of Fig. 9.1 are capable of producing any value of output
voltage betweefi maximum positive and maximum negative; together they are
capable of causing the current toflowin either polarity through the load. If the
firing points of the thyristors are altered continuously so that the voltage applied
to the load varies sinusoidally at a low frequency, then the load will operate in
the alternating current mode and the complete equipment will be seen as a direct
AC/AC frequency convertor.
Bridge A will always supply the positive current to the load and bridge B the
negative current. If the load happens to be a resistor only, then bridge A will also
provide the positive half cycle of voltage and bridge B the negative half cycle.
If, as is more usual, the load is partially inductive then there will be a period
when positive current is stillflowingafter the voltage has become negative and
vice versa as shown in Fig. 9.2. These conditions are satisfied by the convertor
bridges operating in the inversion mode and during these periods they extract
energy from the load and feed it back into the AC mains supply system.
The frequency of operation is changed by altering the rate at which the
voltage is varied and the magnitude of the voltage is altered by altering the range
of phase angle variation used in each output cycle. Fig. 9.3 is a plot of the output
firing
120
Fig. 9.3 Cycloconvertor firing angles
150
180 <f9 l e
degrees
312
The cycloconvertor
voltage characteristics of the two naturally commutated six pulse bridges A and
B of Fig. 9.1. If the phase angle of the bridges is sinusoidally varied from 80 to
100 degrees, then a sine wave of maximum value of approximately 17 per cent
of the maximum voltage Vmax will be produced. If the range is extended to
70 degrees to 110 degrees, then the sine wave peak will be 34 per cent of Vmax
and so on, so that a full range of output voltage can be obtained. In this figure
the angles greater than 90 degrees represent inversion. When traversing one sine
wave from zero, to positive voltage back to zero and then negative and back to
zero the phase angle will follow the path O to P, then back to O, it will then
continue to Q until the current reduces to zero when it switches to bridge B at
point R. It then moves up to the maximum negative voltage at S, back to O
continuing on to T until the current gets back to zero, when it switches to point
U, then to P and so on.
In practical equipments it is usual to use a firing phase shift and pulse
generation system, whereby the output voltage of the convertor is directly
proportional to the input signal voltage to this firing system. Then the output
voltage from the double bridge convertor will always directly follow the level of
the signal voltage being applied to thefiringsystem. When designed in this way
it is possible to consider the complete double bridge convertor with its firing
system as a power amplifier capable of faithfully following the input signal but
at a very much higher power level.
Therefore with such a double convertor bridge arrangement it is possible to
produce a wide range of sinusoidal AC output conditions with the size of the
voltage being dependent on the range of firing pulse phase shift employed and
the frequency being the result of the speed with which this phase shift is changed.
As will be seen in the next section, this principle works satisfactorily if the
output frequency is low compared to the mains frequency and its performance
deteriorates as the frequency is raised above an appropriate level. You will also
appreciate later that the changeover of the current from one bridge to the other
is also an area where the system deviates from the theoretical ideal.
9.2.2 3 phase systems
For motor drives it is necessary to produce a 3 phase output at the appropriate
frequency and voltage and to do this three double bridge arrangements as in
Fig. 9.1 are employed, i.e. one for each phase of the motor. In order that they
can operate independently, the three convertors have to be electrically isolated
from each other and this can be introduced on the supply side using transformers,
or on the motor side by using isolated motor windings rather than star or delta
connections. Fig. 9.4 shows the most effective way of achieving this isolation
using separate motor windings, with relatively small AC line reactors to prevent
the convertor bridges from interfering with each other.
The three convertors are then arranged to produce three output voltage
waveforms at identical frequencies and voltage levels but displaced in time by
120 electrical degrees when referred to the operating frequency. This ensures
The cycloconvertor
313
that a smoothly rotating motor stator MMF waveform is produced to give good
motor performance.
Using this arrangement it is not normally possible to use standard, off the
shelf, motors, as they will be star or delta connected. All six ends of the stator
mains supply
line
reactors
-«*r
-V-
• V
motor
Fig. 9.4 3 phase cycloconvertor
windings need to be brought out separately for independent connection to the
convertors. Also the maximum voltage which can be applied to these windings
is directly dictated by the voltage in the mains supply system.
For these reasons it may be more practical to use the arrangement shown in
Fig. 9.5 where the necessary isolation is provided by the use of mains side
transformers. These also allow the use of star and delta connected motors and
for the circuit voltage to be chosen to be appropriate for the motor.
In general therefore if a completely new system is to be designed and a special
motor can be employed then the circuit of Fig. 9.4 would be appropriate. If,
however, it was necessary to design the system suitable for a standard or an
existing motor, then supply transformers would be used as in Fig. 9.5. This latter
arrangement also allows a more optimum design to be produced because the
voltage levels can be chosen after consideration of all relevant factors.
9.2.3 Reversal and regeneration
It should now be clear that this system does provide complete control over the
314
The cycloconvertor
sinusoidal voltages applied to the motor and that the reversal of the motor
rotating field, and hence the direction of the motor, is simply a matter of either
altering the sequence of the three phases or reversing the polarities of all three
voltages being applied to the motor. Both of these features can be achieved by
making the appropriate changes to the sine wave reference signals being applied
to the firing system.
mains supply
supply
transformers
•-wr
Fig. 9.5 3 phase cycloconvertor
This drive is also fully capable of transferring power from the motor to the
mains supply system as well as from the supply to the motor, as is usual, i.e. it
is a fully regenerative system. Reference to Fig. 9.2 shows that with all inductive
type lagging loads there will be periods in every cycle when the current and
voltage are in opposite directions and during these periods the convertors are
operating in their regenerative modes, i.e. feeding power back into the supply.
And so in every cycle this drive is automatically switching from motoring to
generation during its normal operation. If the power factor of the load reduces
then the period of the output cycle when regenerationis occurring increases and
the motoring period reduces. If the load is a zero power factor fully inductive
one then equal periods of motoring and regeneration will occur. If the load
power reverses it is the same as allowing the power factor angle to increase
above 90 electrical degrees and in this case the period of regeneration becomes
greater than the motoring time.
These conditions are shown in Fig. 9.6 which clearly shows that regeneration
The cycloconvertor
315
is a fundamental aspect of this drive under all conditions of operation thus
making it a veryflexiblesystem.
This aspect also explains why the convertors used in this drive have to be fully
controlled bridges with thyristors in all arms. The convertors have to be capable
of being operated in their regenerative mode at any time and hence they have
to be selected with this in mind.
regenerating O-7p.f.
Fig. 9.6 Inversion and regeneration
9.2A Supply side conditions
The cycloconvertor is not particularly good from the input supply side point of
view and in general it is necessary to have a relatively high capacity, low
impedance and reliable power system to feed a cycloconvertor.
The main problem is the way in which the low frequency output to the motor
is reflected into the supply side currents. The motor current is continuously
changing, following the sinusoidal curve at the output frequency. This means
that the input current to that phase convertor is also varying sinusoidally at the
316
The cycloconvertor
output frequency. At the same time, because the motor voltage is also changing
sinusoidally, the power factor of the supply current into the convertor will
change in a similar way as the output current changes.
In general, the average power factor on the supply side is usually low because
it is dependent on the convertor output voltage level. This voltage is proportional
to the motor frequency (and hence speed) in order to ensure relatively constant
motor flux. Therefore the convertor input power factor will also be approximately proportional to motor frequency. There is also another reason which
lowers the power factor; the convertor has to be rated to supply the peak of the
motor voltage waveform and it is only at this point that the input power factor
is relatively high. During the remainder of the cycle the convertor power factor
will be varying from zero to this maximum level resulting in a relatively low
average value (see Fig. 9.23).
motor voltage
motor current
1 cycle at
motor frequency .
Fig. 9.7 Supply side conditions from one load phase
Fig. 9.7 shows what happens on the supply side under a specific motor
operating condition. In each convertor phase the input current and its power
factor follow the output sinusoidal pattern and it should be noticed that the
current is always a lagging current when looked at from the supply side,
whatever the motor condition.
Further study of this diagram will also show that the output current and
voltage will be out of phase due to the load motor requirements and that, as a
The cyclocon vertor
317
result, the changes in supply power factor will also be out of phase with the
changes in supply current by a similar amount. The supply conditions therefore
also depend on the motor power factor.
Without going into a lot of detail the result is that the input conditions on
each phase of the cycloconvertor are intimately bound up with the cyclic
changes of the motor current and voltage at the output frequency.
Fortunately when the three phases of a complete drive are added together
on the supply side the result is a considerable improvement over any of the
individual phases. Because the variations in supply current and power factor on
the three phases are displaced by 120 electrical degrees (with reference to the
motor frequency) the changes are all phase displaced so that when added
together the variations turn out to be occurring at six times the output frequency
and at a considerably reduced magnitude.
Fig. 9.8 shows the total 3 phase supply conditions for a typical operating
condition of the motor where the RMS input current variation can amount to
approximately five per cent of the peak motor rating at six times the motor
frequency.
sum of all three phases
load
1A
3A
2A
1B
3B
2B
1A
individual load phase currents and VA's
Fig. 9.8 Supply current and VA total for three phases
When conditions are varying so much in a complex mixture of supply and
motor frequencies it is not easy to appreciate the concept of supply power factor
in its normal sense. It is usual therefore to consider the average power factor
(averaged over the motor cycle) as being the most satisfactory parameter. As the
instantaneous value of the supply power factor varies from zero to a value
appropriate to the convertor output voltage peak, then this average value is
relatively low even under rated speed conditions (see curve A in Fig. 9.9).
Sinusoidal and trapesoidal control
Up to now we have been assuming that the output voltage to the motor is varied
in a sinusoidal way at all times and under such conditions we have established
that the average power factor is relatively low. At the higher output frequencies
318
The cycloconvertor
and motor speeds, it is not essential to retain a sinusoidal voltage waveform and
departing from it can improve the average power factor. If the DC output
voltage of the convertor is allowed to stay at the high values for a larger
proportion of the cycle the average power factor will improve. This is usually
known as trapesoidal control and under this the output voltage waveform is
made to follow B in Fig. 9.9 (these waveforms being drawn here with the same
RMS value). If this is taken into account in the choice of the voltage ratings (see
Section 9.3.4) an improved average power factor as shown on Fig. 9.10 can
result (curve B).
sinusoidal
trapesoidal
Fig. 9.9 Output voltage waveforms
If it were practical to use complete square wave control as waveform C then
an even higher average power factor is possible (see curve C on Fig. 9.10).
In practice it is better to use sinusoidal control at low speeds and gradually
progress to trapesoidal and maybe square wave control at high speeds; then an
average power factor curve D may be the result.
The curves of Fig. 9.10 have been drawn on the basis of the motor operating
at unity power factor. Operating the motor at less than unity will also reduce the
supply side power factor to a similar degree.
9.3 Detailed analysis of the system
9.3.1 Circuit waveforms
Convertor and motor voltages
The voltage waveforms that are applied to the motor windings are derived from
the six pulse convertors which are fed from the fixed frequency mains supply.
The cycJoconvertor
319
The convertor output voltage contains a harmonic component at six times the
supply frequency (which is nominallyfixed)and therefore the actual shape of the
motor voltagfc waveform will be different at all output frequencies and voltages.
d practical combined
p.f. curve
a sinusoidal
control
50
motor speed or frequency %
Fig. 9.10
100
Supply power factor and waveform
Let usfirststudy a typical motor voltage waveform as generated by such a six
pulse double bridge single phase convertor as shown in Fig. 9.1. It is assumed
that this convertor is connected to an inductive load so that the current will be
reasonably sinusoidal and will lag the fundamental voltage waveform. Fig. 9.11
shows this typical motor voltage waveform. This diagram shows a sinusoidal
required voltage waveform, which is the sort of signal which will be input to the
firing circuit, as well as the actual voltage waveform containing six times the
supply frequency ripple, and the current. It should be studied carefully if the
principles are to be fully understood and note should be made of the following
points. For convenience of understanding, overlap between the mains frequency
convertor thyristors is ignored in drawing this diagram.
a) During period TO to T2 the current is always positive and Bridge A
is in operation.
b) During period T2 to T4 the current is always negative and therefore
Bridge B is in operation.
c) Between TO and Tl, Bridge A is operating in the rectifying mode with
the mean voltage reaching zero by Tl.
d) Similar conditions exist for Bridge B during the period T2 and T3.
Fig. 9.11
Cycloconvertor voltage waveforms
inverting
required voltage waveform
mains frequency waveforms
inverting
A bridge rectifying
The cyc/oconvertor
321
e) Between Tl and T2, Bridge A is operating in its inversion mode and
between T3 and T3, Bridge B is inverting.
f) The actual voltage waveform contains a significant content of
harmonics at six times the mains supply frequency.
g) The time between thyristor switchings varies throughout the sine
wave of output and hence the periods of conduction of the thyristors
is continually changing.
h) The negative half of the voltage waveform is not a mirror image of the
positive half cycle because the output sine wave is not normally in
synchronism with the mains frequency waveform,
i) There is a sudden discontinuity in the voltage waveform when the
current crosses zero and therefore transfers from Bridge A to Bridge
B or vice versa. In practice there may be a period of zero current and
voltage depending on the design of the bridge changeover circuits (see
Section 9.4.2).
In fact the precise shape of the voltage waveforms depends on the way in
which thefiringpoints of the thyristors is decided. In thisfigureI have assumed
that a linear convertor output voltage/firing circuit input signal system as
previously discussed is being used but other strategies are used in specific
cycloconvertor designs.
In order to demonstrate the degree of variation involved in these motor
voltage waveforms I have prepared Fig. 9.12 which shows four conditions which
are appropriate to a motor drive cycloconvertor. These show the motor phase
voltage waveform which will exist at four different frequencies assuming that the
peak of the 16-7 hertz output wave just requires the maximum convertor voltage
available and that a constant voltage/frequency ratio is maintained to keep the
motor flux reasonably constant. From these diagrams it should be noticed that:
a) All these waveforms contain a significant content of the six times
mains frequency harmonic, the low frequency conditions contain the
largest number of these high frequency pulses per output cycle.
b) At low frequency where the depth of modulation is lowest the positive
and negative half cycles of the output are relatively equal whereas at
high frequency more imbalance occurs.
c) If the flow of current in the load is going to be relatively smooth and
continuous the load inductance must be sufficient to prevent the
current reducing to zero at each of the high frequency pulses. This is
particularly so at low frequency where the voltage wave crosses zero
many times in the cycle.
d) At the higher frequency the time between commutations is much
shorter as the voltage rises than when the voltage is reducing.
Under the 16-7 hertz condition the magnitude of the voltage being applied to
the motor will be equal to 0-707 times the maximum DC voltage from the
convertor. If a higher voltage than this is required then it will be necessary
to diverge from the sinusoidal shape as previously explained. Fig. 9.13
Fig. 9.12 Voltage waveforms under different output conditions
100 Hertz
100mS/cycle
voltage peak 60%
frequency ratio 0-2
667 Hertz
150m5/cycle
voltage peak 40%
frequency ratio 0 13
5"
§
Co
i
,
I i
<
Fig. 9.12 Continued
i
s
s
60mS/cycle
voltage peak 100°/o
16 7 Hertz
!
s
,
s
/
voltage peak 80°/o
13 33Hertz 75mS/cycle
current zero
Co
Co
1
§
324
The cycloconvertor
demonstrates the resulting waveform when an appropriate 20 hertz reference
wave is applied to the convertor. During part of the half cycle the reference wave
exceeds the maximum controllable value and the convertor just produces the
maximum that it can causing a flat-topped waveform to be produced.
This is the shape of waveform which is usually used at the higher frequencies
so that the maximum power rating can be achieved. By this method it is possible
to produce approximately 15 per cent more output voltage and power than if
a sinusoidal output is used.
20 hertz 50mS/cycle
voltage peak 120%
frequency r a t i o 0 - 4
Fig. 9.13 High value of output voltage
Convertor currents
Over the majority of the useful range of a cycloconvertor the currentflowingin
the motor is reasonably sinusoidal at the low motor frequency. The positive half
cycle of current flows through the one convertor and the negative half cycle
through the other as shown in Fig. 9.14. The convertor then chops up this
current at the relatively high mains frequency so that it is shared out between
the three phases supplying each bridge. The overall result is that the current
being fed into one of the reversible phase convertors will be a mains frequency
current whose magnitude will be modulated by the low output frequency as
The cycloconvertor
325
shown in the diagram. This low frequency modulation from the other two
phases will be 120 electrical degrees out of phase (referred to the low frequency)
so that the sum total current to the three convertor phases will only be
modulated to a small degree at six times the motor frequency and so it can be
considered to be a substantially constant mains frequency input current.
current in positive
convertor bridge
current in negative
convertor bridge
supply line current to convertor
Fig. 9.14 Convertor currents
9.3.2 Current reversal
With the power circuit discussed up to now and shown in Fig. 8.2 the transition
of the current from negative to positive and vice versa needs some more detailed
326
The cycloconvertor
explanation. This circuit can only be used successfully if only one of the bridges
is in operation at one time. If ever both bridges were fired up together a short
circuit between the two convertors would result and very large fault currents
would flow. As there is a risk of such a short circuit occurring during the
transition of the current from one bridge to the other it is usual to allow a short
period of time with the current at zero. This allows time for the outgoing bridge
to fully turn off before the incoming bridge receives firing pulses. The length of
the delay depends on the reliability of the zero current measurement system and
the expected time for the outgoing thyristors to regain their blocking capabilities.
The turn off time of the convertor grade thyristors used in this circuit is usually
less than one millisecond and total zero current periods of a few milliseconds are
often used, depending on the level of the circuit inductance and the degree of
precision of the zero current checking circuits.
If this switched approach to current transfer is considered to be unacceptable,
perhaps because the zero current delay to produce reliable operation is too long,
then a much smoother transition can be achieved by operating both bridges at
all times and allowing the current to flow in either direction as required by the
load. This can be achieved by feeding the two bridges from separate transformer
supplies and by including output reactors to limit the circulation of harmonic
currents between the bridges. The principle used during current transition is to
allow a small current to circulate between the two bridges (controlled by
matching the voltages of the two bridges) so that both bridges are in operation
in order that the current can transfer naturally. Very smooth current changeover
can then occur and almost true sinusoidal current can result.
As the switched system is satisfactory for the majority of motor drives, I do
not intend to go into the circulating current system more deeply here; reference
should be made to the appropriate papers listed in the Bibliography section.
9.3.3 The motor vector diagram
From the motor point of view this drive operates in the same way as a normal
mains supply or a voltage source inverter, in that the drive provides a voltage
to the motor and the motor is allowed to draw whatever value and phase of
current that it requires. Figs. 1.15 and 1.23 show the motor vector diagrams
appropriate to an induction motor and a synchronous motor. With an induction
motor the current will always lag the voltage vector and assuming a relatively
constant voltage to frequency ratio is maintained then a relatively constant
reactive or magnetising current will be drawn by the motor.
When a synchronous motor is fed from a cycloconvertor the current which
flows in the motor is directly dependent on the value of the air gap flux. The
current is dictated by the difference between the applied voltage and the induced
voltage. The induced voltage is the result of the air gapfluxand due to armature
reaction this results from the difference between the appliedfieldcurrent and the
MMF caused by the armature current. As explained in Chapter 1 the motor
system balances when the induced voltage matches the applied convertor output
The cycloconvertor
327
voltage and this only occurs when the magnitude and power factor angle of the
current are such that the resultant flux is correct. As the cycloconvertor will
allow whatever current is needed to flow, stable operation is easy to achieve.
9.3.4 Relationships and equations
The cycloconvertor is a voltage source drive and is in fact the nearest drive to
a variable frequency sinusoidal mains supply of all those studied in this book.
The voltage applied to the motor is usually chosen in relation to the frequency
to ensure a reasonably constant level of air gap flux and in the case of a
synchronous motor to ensure a good motor power factor.
The torque produced by the motor is directly given from the inphase
component of the current and the air gap flux.
mains supply
Fig. 9.15 3 phase cycloconvertor
In order to deduce relationships between the various parameters of this
system let us study the cycloconvertor as shown in Fig. 9.15 consisting of three
transformer fed switched reversing convertors, each feeding one of the star
connected windings of the motor (which could be an induction or a synchronous
motor). The variables shown on this diagram are as follows:
Vline is the RMS value of the motor fundamental line voltage
Vm is the RMS value of the fundamental phase voltage
328
The cycloconvertor
Im is the RMS value of the fundamental motor line and phase current
Vdc is the mean value of the DC voltage
Idc is the current in each of the phase convenors
Vt is the RMS line voltage to the convenor
It is the RMS line current to the convertor
Pfm is the power factor of the motor current.
Pfs is the power factor of the supply current
Vs is the RMS line voltage of the mains supply
Is is the RMS line current drawn from the mains supply
Convertor currents
Reference to Fig. 9.14 shows that the two convenors of each phase carry one
of the two half cycles of motor current. The current in one convertor is therefore
a half sine wave at motor frequency having a peak value of Im x yjl.
The average value over the full cycle will therefore be equal to I fit x I m x ^ / 2
and its RMS value will be Im/^j2.
The decision as to how to rate the convertor to accept such current pulses
depends on the magnitude of the current demanded by the motor at different
frequencies. The heating caused by the half cycle of current will depend on the
magnitude of the current and the duration of the current pulse. If the motor is
required to provide high torque at low speeds then the current can be high when
the current pulses last the longest and this will have to be taken into account.
In general, the convertor can be rated based on the RMS value of the motor half
sine wave.
i.e.
Idc = ha/y/2 amps RMS
(1)
Convertor input currents
The supply transformer to one phase of the cycloconvertor will feed both
convertors of the phase and the line current will be a mains frequency current
modulated by the output frequency as shown in Fig. 9.14 i.e. the supply current
will vary in magnitude to follow the rise and fall of the motor current. The
maximum RMS value of this current will occur at the peak of the motor current
and following the normal convertor relationships, this maximum value will be
equal to:
Im x y/2 x V2/V3
and the RMS value over the full low frequency cycle will be this maximum
value divided by y/2, i.e.
It = Im x y/2/y/3 amps RMS
It = Im x -816 amps RMS
(2)
The cycloconvertor
329
The current flowing in the primary winding of the supply transformer to the
one phase convertor will be related to the secondary value by the turns ratio and
the total input current to the three transformers will have an RMS value of
6 x yjljn times the individual transformer primary current, because the 3 phase
currents are out of phase with each other.
Convertor voltages
The maximum value of the convertor voltage required to produce a particular
voltage at maximum motor frequency depends on whether sinusoidal or
trapesoidal voltages are to be produced under this condition. The three waveshapes shown in Fig. 9.9 have the same RMS value but their peak values differ
and the level of DC voltage from the convertor needed to produce the square
or trapesoidal motor voltage waveforms will be lower than if a sinusoidal
waveform is decided on.
If a sinusoidal waveform is chosen then the convertor will have to be arranged
to give a DC voltage equal to
Vm x 72
In practice it is found that the use of an appropriate shape trapesoidal voltage
wave is quite acceptable if its shape is chosen to limit the amount of fifth and
seventh harmonics in the waveform. If this is done it is found that a saving in
DC voltage of approximately 16 per cent is possible so that the maximum DC
voltage can be given by
Vdc = Vm x 72 x 0-84
During operation the DC voltage of the convertor will be continually altering
thus producing the AC voltage.
Convertor supply voltage
As with any convertor transformer the choice of secondary voltage depends on
the conditions under which the DC voltage and hence the motor voltage has to
be maintained. It may be necessary to maintain the motor voltage at its maximum
rated level even when the supply voltage reduces due to other loading on the
system. It will certainly be necessary to allow for the load of the cycloconvertor
and this will cause some reduction of transformer secondary voltage due to its
impedance. Regulation due to load current will usually result in approximately
afiveper cent loss of volts and hence the open circuit secondary voltage required
will need to be five per cent higher than that needed to produce the above
maximum convertor output voltage.
Therefore the open circuit line voltage of the transformer will need to be given
by:
Vt = 105 x 0-742 x Vdc(max)
and therefore if sinusoidal motor voltage is to be used at the maximum motor
330
The cycloconvertor
voltage, then
Vt = 105 x 0-742 x ^ 2 x Vmr
(3)
where Vmr is the rated value of the motor RMS voltage. If trapesoidal voltage
is to be used then
Vt = 105 x -742 x y/2 x -84 x Vmr
(4)
These values assume that regeneration does not cause any other limitations.
Study of Fig. 9.11 will show that regeneration occurs for a period depending on
the motor power factor. The regeneration period starts when the motor voltage
is zero andfinisheswhen the voltage reaches a value which alters with the motor
power factor. As long as it is possible to be sure that the convertor can always
produce sufficient voltage at the end of the regeneration period, then all is well.
If the system is such that high currents at low power factors have to be catered
for it may be necessary to make additional allowances to ensure that inversion
failure does not occur.
Supply power factor
As described previously the phase angle of each convertor is continuously
changing so that only the power factor to the total 3 phase cycloconvertor really
means anything worthwhile and this can best be assessed from the input KW
and KVA.
If we assume that the supply transformers have a 1:1 ratio then the input line
current to the complete cycloconvertor will be given by
Is = 6 x yjl x -816/71 x Im amps RMS
= 2-204 x Im
amps RMS
For a fully sinusoidal voltage design the transformer primary line voltage will
from equation (3) be equal to
1.102 x Vmr
and therefore the total primary KVA equals
2.204 x 1-102 x V3 x Im x Vmr
KVA = 4-21 x Im x Vmr
(5)
The KW supplied to the cycloconvertor will be equal to the KW into the motor
plus the power losses in the cycloconvertor. Therefore
KW in = Im x Vm x 3 x Pfm + Convertor and Transformer Losses
or alternately
KW in = Im x Vm x 3 x Pfm/Efc
(6)
Where Efc is the cycloconvertor efficiency. Therefore the supply power factor is
The cycloconvertor
331
given by
__
(Im x Vm x 3 x Pfm)
(4-21 x Vmr x Im x Efc)
Pfs = 0-713 x Vm/Vmr x Pfm/Efc
(7)
If the system has been designed on the basis of trapesoidal control, the supply
voltages can be lower for the same motor voltage and the supply total power
factor will be given by
Pfs = 0-85 x Vm/Vmr x Pfm/Efc
(8)
9.3.5 Examples of calculations
1) Specification of supply transformers
Question
A low speed synchronous motor driving a large cement mill has to provide 4000
KW to the load when running at 20 RPM. The motor is designed for operation
at 1500 volts RMS per phase in sinusoidal operation at five hertz and operates
under this rated condition at an efficiency of 93 per cent and at a power factor
of unity. If a switched reversing convenor as in Fig. 9.15 is to be used and the
convertor efficiency at rated load is 98 per cent, specify the secondary voltage
and current ratings and the KVA rating of the three transformers feeding the
convertors.
Answer
The motor input KW equals
4000/0-93 = 4301 KW
The motor KW per phase = 1434 KW.
As the motor operates at unity power factor
Motor current Im = 1434 x 1000/1500
= 956 amps RMS
From equation (2) the secondary line current of the supply transformer to one
phase of the cycloconvertor will be equal to
•816 x Im = 780 amps RMS = It
as
From equation (3) the transformer open circuit secondary line voltage is given
1 05 x -742 x yjl x Vmr for sinusoidal operation
Vt = 105 x -742 x y/2 x 1500
= 1652-5 volts RMS
332
The cycloconvertor
However this does not allow for the efficiency of the convenor which will
cause some increase in the necessary supply current or voltage. Let us allow for
it in the voltage i.e.
Vt = 1652-5/0-98 = 1686 volts RMS
Transformer secondary KVA rating
= It x Vt x v^/lOOO
= 1686 x 780 x V3/1000
= 2278 KVA per motor phase
2) Supply power factor
Question
What will be the supply power factor to the total drive when the motor is
running at a frequency of 2-2 hertz with a phase voltage of 700 volts at a power
factor of 0-9 leading if the convertor efficiency under this condition is 96 per
cent.
Answer
From equation (7) the total supply power factor equals
0-713 x Vm/Vmr x Pfm/Efc
=
-713 x 700/1500 x 0-9/0-96
= 0-312 per unit.
9.4 Practical circuit design considerations
The power switching elements of a cycloconvertor will normally be naturally
commutated thyristors and the reversing bridges used will be very similar to
those used for DC motor drive applications in many cases. As usual the
thyristors will befittedwith snubber circuits to protect against high voltages and
dv/dt, they will be mounted on heatsinks and they may be protected by fuses in
series with them. If the thyristors are used in parallel to achieve the rating they
may have reactors in series with them to ensure good current sharing. If they are
used in series then some components will be included to ensure correct sharing
of the total voltage. In other words, the cycloconvertor will consist of conventional bridge connected thyristor assemblies containing all those peripheral
items essential to naturally commutated circuits. The convertors will be very
similar to those which would be used for the synchro-convertor drive described
in Chapter 7. From the practical point of view the cycloconvertor just consists
The cycfoconvertor
333
of three AC to DC converters connected to supply the windings of a 3 phase
motor.
9.4.1 Overcurrent protection
The most important overcurrent fault conditions which can occur with this
system are those associated with:
(a) Switching the full convertor voltage onto the motor winding inadvertently, and
(b) Switching both forward and reverse convenors on together or
switching one of the convertors on before the current in the other has
become zero.
Both of these conditions can be severe and they can cause large currents to
flow in the circuit sufficient to damage the components if they are allowed to
persist.
Switching the full voltage of the convertor suddenly onto the motor, whether
it is rotating or not will quickly saturate the motor core and it is only the
resistance of the winding which will limit the level of current reached. The small
inductance of the winding will limit the rate of rise of the current but this does
not usually have a great effect within the normal time scales of protection of one
to ten cycles at mains frequency.
The inadvertent conduction of the two convertors of one phase is even worse
in that this produces an almost zero impedance short circuit and it is only the
impedance of the AC supply system which will limit the level of the fault
currents which will flow.
In the smaller drives, up to say 500 KW, it is likely that the inclusion of fuses
in the circuits is the only practical and economic method of protecting the
healthy components from the effects of these faults. As, with many well designed
electronic control schemes such faults will be relatively rare, the presence of
fuses is usually acceptable and such drives can operate for many years without
fuses having to operate to protect the drive.
Drives which regenerate regularly can be more at risk from these faults,
because whenever power is being fed back to the supply system, it is essential
that the mains sine waves are present always. If, perhaps due to fault conditions
on the mains network, these supply voltages ever disappear or reduce to a low
value then a failure to commutate will occur in the convertor and the result will
be a short circuit involving the convertor and the motor winding. Clearly if the
drive is regenerating for a significant period of its operational life then there is
more of a risk of this occurring and in particularly bad supply cases fuses may
blow regularly due to such faults.
In these cases, and in others where the consequences of fuse blowing may be
too serious, other methods of protection need to be used.
The most frequent method is to include additional impedance in the circuit
to reduce the level of fault currents to values which the components can
withstand for the time needed to open a supply circuit breaker. This impedance
334
The cycloconvertor
can be built into the supply transformers or additional AC line reactors can be
introduced for this purpose. Although the inversion failure condition could be
limited by reactors in series with the motor this is not usually done for protection purposes because they would not limit the fault currents during bridge
faults, whereas input side reactors will.
9.4.2 Convenor polarity switching
One of the areas of difficulty in a cycloconvertor is in the arrangements made
for selecting the correct polarity convertor which should be in operation at a
specific time. As will be seen in Figs. 9.2 and 9.11 the current zero will usually
be delayed from the voltage zero to the inductive nature of the motor load (in
the case of a synchronous motor the phase angle of the motor current can vary
from leading to lagging but the point under discussion is not significantly
.
.
J
I
.
.
I
I
positive bridge
firing pulses
negative bridges
tiring pulses
11
|
•
t
t
X
B-current zero detected
C-negative bridge released
D current starts rising
voltage
I
1
1
voltage ref
i
I
Fig. 9.16 Practical current transfer
I
I ' ['1
"T
h
'
The cyc/oconvertor
335
affected). In general, the point when the zero crossing of current should occur
varies as the load conditions change. At this point it is necessary to change over
from the positive convertor to the negative convertor or vice versa and the ideal
would be a smooth transition from the one convertor to the other without a
hesitation in the current. With switched convertors it is not easy to achieve this
ideal objective because it is essential to ensure that there is no possible chance
of the two convertors conducting at the same time. It is necessary to check that
the current has reached zero before considering firing the incoming bridge, in
order to make sure that the thyristors in the outgoing bridge have regained their
blocking ability.
Fig. 9.16 shows the changeover from positive to negative current as it occurs
in practice. The actual voltage waveform is shown to be increasing negatively at
the point the current reaches zero — Point A — due to the inductance of the
zero
current
monitoring
rn
? ? t
j
zero
current
detection
Z.C.D
-4 -i -i
AC supply
Fig. 9.17 Zero current detection methods
motor
winding
336
The cycloconvertor
load. Current is detected at point B to be zero and this initiates changeover to
the negative bridge. The pulses on the negative bridge are released after a time
delay of approximately two milliseconds but due to the back emf from the motor
the current does not really start rising until point D.
Clearly the delay necessary to ensure satisfactory reliable changeover does
cause some distortion to the motor current and this will be most significant
under the highest operating frequency condition. However when one remembers
that some drive systems apply square wave currents to the motor quite satisfactorily, one realises that this distortion is unlikely to cause serious difficulties.
Deciding the correct point of switchover is even more difficult if the current
becomes discontinuous, which it can do under low load current conditions. In
such cases it is not sufficient to check for zero current continuously and cause
a changeover as soon as one is detected, as this will result in changeover much
too early. The method often used is to predict the point where changeover
should occur electronically and then only look for the current zero at around
this predicted point.
The most common method used to detect zero current in order to initiate
bridge changeover is to use the voltage across a diode which is carrying the
current. While the current isflowingthere will be a small forward voltage across
it. Once the current ceases a relatively large reverse voltage will appear across
the diode. This method is found to be much more satisfactory than a more
conventional linear method of current measurement. The diodes can be connected
directly in series with the main convertor bridges as shown in Fig. 9.17(a) but
in this case a significant additional power loss may be involved, particularly in
large convertors. A more economical method is to monitor the circuit current
via current transformers and to put the diode in the output of these as shown
in Fig. 9.17(b).
9A3 Alternative power circuits
In the earlier explanations in this chapter I have concentrated on the use of
bridge convertors in a switched anti-parallel system, as this is found to be the
most frequently used circuit. However there are a number of other circuits which
have been used and which do have merit in specific circumstances. Before
discussing these I should say that the use of bridge circuits is not an essential
feature of cycloconvertors. The convertors have to be capable of full regeneration
and any circuit which can do this can be used. Many of the early cycloconvertors
used mercury arc rectifiers and the use of the six phase half wave circuit
dominated these applications. However with the present universal use of
thyristors the bridge circuit is now the first choice.
The three pulse cycloconvertors
The three pulse cycloconvertor of Fig. 9.18 is probably the simplest system that
can be used with motor drives. Each phase consists of two reverse connected
three pulse half wave convertors connected to a common 3 phase four wire
The cycloconvertor
337
supply. The motor windings are effectively connected in star because one end of
each is connected to the supply neutral. When this system is operated with a
balanced 3 phase load there will be no current flowing in the supply neutral
connection so that as long as the three motor winding ends are connected
together as a star point it is not essential to connect this point to the neutral.
However it is easier to understand if drawn as in the diagram.
AC supply
Fig. 9.18 The three pulse cycloconvertor
The convertors operate in a very similar way to the six pulse bridge system
already explained except that the voltage ripple produced by the convertors is
at three times the mains frequency. This means that there is a large increase in
ripple current in the system and the acceptable range of output frequency is
limited compared to the six pulse case.
The voltage harmonics for the three pulse circuit are approximately four times
the size of those for the six pulse circuit and they are also at a half of the
frequency, hence it is difficult for the motor current to follow a true sine wave.
In fact the current usually contains a large amount of this third harmonic with
the current being discontinuous, being applied to the motor in pulses at three
times the supply frequency. The result is very non-linear control behaviour and
problems in establishing the correct points for bringing in the reverse convertor.
The performance can be improved if continuous circulation of current between
the forward and reverse bridges is allowed but then additional large reactors are
needed and significant losses occur.
It is always difficult to establish firm values for the practical operating
frequencies of cycloconvertors, because it depends on the method offiringpulse
generation and the sensitivity of the load to low beat frequencies. However it is
possible to say that the highest frequency practical for the three pulse circuit is
approximtely half that for the six pulse circuit, hence limiting the output
frequency on mains power supplies to the ten to twelve hertz region. Clearly this
may not be a limitation if a high frequency power supply system at say 400 hertz
is available as is the case in some ship and aircraft mounted systems.
338
The cycloconvertor
The delta connected cycloconvertor
An alternative six pulse circuit using the same number of thyristors as the three
pulse circuit above, is the delta-connected cycloconvertor circuit of Fig. 9.19.
With this arrangement it is not necessary to use reversing convertors in each
phase. The principle is that the negative current required in each line flows
in the adjacent convertor to that which carries the positive half cycle. The
convertor current therefore alwaysflowsthe same way around the loop of three
convertors.
mains supply
Fig. 9.19 The delta cycloconvertor
In order that the three convertors can operate independently, it is necessary
to feed them from isolated mains supply transformers. Also, as the three
convertors work at the same time, reactors will probably be needed to limit the
level of harmonics at six times the mains frequency flowing in the loop.
Although this circuit uses the convertors more effectively with the two half
cycles of output current flowing in each convertor, the size of the transformers
has to be increased by approximately 55 per cent and the average power factor
is nearly 22 per cent lower than using the six pulse reversing convertor circuit.
9.5 Overall control methods
With this direct AC to AC cycloconvertor system, control over both the
frequency and voltage to the motor is carried out using thefiringcontrol of the
convertors and hence it is necessary first of all to consider firing methods used.
The cycloconvertor
339
9.5.1 Firing control
Although other methods are possible the traditional approach has been to
choose a firing system which has a linear characteristic between convertor
output voltage and the control signal fed into the firing circuit. If a sinusoidal
control signal of the desired frequency and representing the desired magnitude
is fed into thefiringcircuit, then the result will be a correct output from the
convertor.
The relationship between the firing angle and output voltage of a bridge
convertor of the type shown in Fig. 9.1 is a cosine curve as shown in Fig. 9.3
and hence it is necessary to produce another cosine relationship between input
control signal and firing delay angle if a linear overall result is to be achieved.
convertor mains sine waves (refer diagrams 3-1 and 3-2 )
\ control
/ \signal
v
A
V
V
A\
•- r
A.
Fig. 9.20 Firing arrangements
\/'
V
\
y
K ^ - K - 7 :
•i
.
.• -v
/ \_ / \
\
;
/ \
>f
\
\- reference \.
X
K
X
•r '' r ''
Such a relationship is produced by comparing the firing circuit input signal
with reference sine waves derived from the mains supply feeding the convertor.
If the correct phase angle of reference wave is obtained and the correct part of
the wave is used, the required linear relationship can be produced.
Fig. 9.20 shows the principle as applied to the positive side of a six pulse
bridge circuit. The lower diagram shows the half sine waves used as reference
340
The cycloconvertor
waves for comparing with the control signal, the point where the control signal
cross these reference waves is the point tofirethe appropriate thyristor arm. The
diagram shows a changing control signal and points Tl, T3 and T5 for firing the
appropriate thyristors to produce the output voltage as shown in the upper
diagram. Using this method the firing delay angle is given by:
as long as the reference wave is phased to start at the free firing point of the
appropriate thyristor arm, i.e. Point X, thus leading to a linear output voltage
to control signal relationship. The other sections of the convertors are dealt with
in a similar way to produce the complete system.
With such an arrangement any output waveform can be produced as long as
it is within the limitations of the six pulse mains frequency convertors. If a sine
wave control signal is applied to thefiringcircuit, then the convertor output will
follow this sine wave shape. If a triangular or trapesoidal control signal is
applied, then the output to the motor will also follow these shapes as truly as
the six pulse system allows.
It may at this point be useful to refer back to Fig. 9.12, because they assume
this method of firing.
9.5.2 Typical control schemes
Most of the cycloconvertor control schemes therefore incorporate a linear
transfer characteristic for the convertor and then their job is to provide the
appropriate signals for frequency, voltage magnitude and polarity into the firing
circuits, so that the required performance and protection features are achieved.
The specification of the control scheme is therefore dependent on the load
duty which the drive has to perform and on the type of motor being used. When
used with an induction motor the signals fed into the firing circuit are the only
variables available and the optimum combinations of these parameters have to
be derived by the control system, if the required performance is to be achieved.
In the case of a synchronous motor there will be additional independent control
over the motor field current and this enables optimisation of the motor power
factor and in general more precision to be achieved in motor performance. The
fact that the motor speed always corresponds directly to the applied frequency
is also a benefit compared to the induction motor where the slip between motor
frequency and speed varies with the load and the magnetisation condition of the
motor.
Control scheme for a synchonrous motor
Because the speed follows the frequency, it is not essential to measure the speed
using an encoder or tacho-generator; a speed monitor based on the applied
frequency can be used if needed. Such a system can, however, have an open loop
speed control arrangement where the desired speed directly dictates the frequency
to be applied to the cycloconvertor and to the motor.
The cycloconvertor
341
Fig. 9.21 shows a typical synchronous motor scheme where the speed requirement sets the operating frequency and the field control is based on a motor
stator current measurement. In this scheme the relationship between voltage and
frequency is set by the V/f function which allows for stator resistance at the low
frequencies. This should enable motor operation at constant air gap flux to be
achieved.
mains supply
current
measurement
voltage
absolute
value
current
limit
polarity
detector
v.f.
function
frequency
firing circuit
three phase
cyclo -convenor
oscillator
direction
motor
current
measurement
ramps
frequency
reference
current
amplifier
field
system
function
generator
synchronous
motor
Fig. 9.21 A cycloconvertor/synchronous motor scheme
The required frequency signal is applied via a preset ramp to prevent the
cycloconvertor frequency being changed too fast. This required signal can be
bi-directional, i.e. it can request forward or reverse rotation and this is picked
up by the polarity detector which decides the direction of the oscillator control
waveforms. The current limit feature in the frequency control is to ensure that
overloading of the system will not damage the motor or the cycloconvertor. It
reduces the frequency if an excessive current is measured.
The field control system is chosen to maintain the motor power factor at or
near to unity under all load conditions. This particular machine has rotor
slip-rings to feed the DC field and a small convrrtor is included to provide
controlled power to it. This convertor is operated on a closed loop current
controlled basis and it is fed with the output signal from an armature reaction
function generator based on a measurement of stator current. This function
generation is set up to match the motor so that near unity power factor
conditions are achieved.
This scheme will allow full four quadrant motoring and regenerating
342
The cyc/oconvertor
performance to be achieved but due to its basically open loop approach, it
cannot achieve the high dynamic performance required for applications such as
steel rolling mills, etc.
In these cases more complex schemes are employed based on the measurement of rotor position and they usually employ a more complex motor model
within the control scheme in order that direct and quadrature axisfieldscan be
assessed independently, so that the true flux in the motor is more accurately
calculated. The cycloconvertor control may also be modified to include closed
loop current control of the stator currents.
Control schemes for induction motors
Due to slip between the motor speed and the stator rotating field it is normally
necessary to include a speed measurement when using an induction motor.
The control of the cycloconvertor can be very similar to that shown in
Fig. 9.21 except that a speed control amplifier is included prior to the current
limit circuit to allow direct comparison between the output of the speed ramp
and the actual speed (see Fig. 9.22).
mains supply
speed measurement
signal
current
measurement
v:f
current
limit
speed
amplifier
voltage
function
frequency
polarity
detector
oscillator
firing
circuit
three
phase
cycloconvertor
direction
ramps
I
speed
setting
-VSAA/WW-
induction
motor
Fig. 9.22 A cyc/oconvertor/induction motor drive
In this case the convertor is like a voltage source inverter in that the current
drawn by the motor will be decided by its magnetisation and torque needs, and
variation of the V/f ratio will decide directly the level offluxgenerated in the air
gap.
More complex schemes have been considered by some authors, using closed
loop current control of the stator currents and sophisticated ways of assessing
the motor flux and torque needs using slip and maybe position measurements
The cycloconvertor
343
of the rotor. There is little evidence of practical schemes of this type being in
service.
9.6 Performance and application
The cycloconvertor is a low frequency drive capable of performing a full
reversing and regenerative duty. It is naturally commutated throughout and it
can therefore be made at large as well as small power ratings and it can be
designed for high voltage operation if required.
Within its frequency limitations it can produce good sinusoidal currents in the
motor and hence the motor torque is relatively free from torque pulsations.
It can be used with either induction or synchronous motors and it can feed
one or a number of motors in parallel. In spite of these apparent advantages it
has not gained widespread acceptance in the general field of variable speed
drives. It has however been taken up in certain specific areas and has come to
dominate in some of them.
The main reason for this is its frequency limitation; it is ideal for drives
operating at up to 15 hertz but there are not all that many applications where
such a low maximum frequency is acceptable.
One area where this drive has become very important is for driving large low
speed cement ball mills where there are advantages in building the synchronous
motor around the large diameter mill drum and supplying it at up to 10 hertz.
Many cycloconvertors of up to 7000 KW rating have been supplied for this
purpose.
Another area where the cycloconvertor has been used over a long period of
time is for steel mill roller table drives where it allows the use of individual,
directly driven rollers operating at low frequency without gear boxes. This
arrangement can reduce maintenance on the mechanical parts of the rollers
which are exposed to red hot steel and corrosive fumes, etc. continuously during
use.
One other disadvantage which may also affect its application is its low supply
side power factor, even when operating at the higher speeds. In many large
drives it may be essential to provide some additional power factor correction
equipment in order to produce an overall acceptable installation.
This low power factor is in fact just one indicator of the low equipment
utilisation which occurs in this drive. For a specific KW output the size and
ratings of the convertors and transformers can be much larger than maybe the
case with other drives. For example, from equations (2) and (3) of Section 9.3.4
the total transformer KVA to supply the 3 phase convertors under sinusoidal
control can be over 1-5 times the motor KVA.
9.6.1 Speed range
Although there continues to be dispute about the range of frequency and speed
344
The cycloconvertor
over which the cycloconvertor can be used, it has in fact been applied successfully in the form described in this chapter at up to 40 per cent of the supply
frequency. Even at this level, however, there is evidence of distortion and
asymmetry in the output current caused by the supply related harmonics.
The performance at low speeds is usually very good because there are a large
number of convertor commutations in each output cycle leading to good
sinusoidal motor currents. A minimum speed of one per cent of top speed can
usually be obtained depending on the accuracy of the reference wave oscillators
or the quality of the control electronics used. Operation at standstill is normally
possible with DC currents flowing in the motor windings to freeze the motor's
rotating field.
9.6.2 Dynamic performance
A motor, driven by a cycloconvertor can be reversed simply by changing the
direction of the reference waves or by reversing their phase sequence. No power
switching is required.
The drive is also capable of operating under regeneration as well as motoring
conditions. Normally regeneration can be induced simply by lowering the
cycloconvertor frequency while the motor is running. The energy in the motor
is then instantly passed into the mains supply and the motor slows down rapidly.
Once the motor speed has reduced to correspond to the lower set frequency then
regeneration will stop and the drive take up its normal operating condition.
With an induction motor, where there is always a slip frequency difference
between applied frequency and the speed, it is necessary to alter the applied
frequency by more than the slip frequency to cause torque reversal to occur. In
the case of a synchronous motor, however, only very small changes in frequency
are required to produce instantaneous torque reversal and very high quality
dynamic performance is possible as a result.
9.6.3 Supply power factor
Reference back to Section 9.2.4 will show that the input power factor to a
cycloconvertor will depend on the way in which the ratings are chosen. If
operation under sinusoidal voltage conditions to the motor is to be achieved
over the complete speed range then the average power factor will be lower than
if operation under trapesoidal is allowed at the higher speeds. The curves of
Fig. 9.10 shows the average power factor of the fundamental supply current for
a drive motor operating at unity power factor.
As I have said before, the concept of power factor is difficult when the values
of KVA and KVAR input are continually changing but equations (7) and (8)
of Section 9.3.4 show that the total power factor, i.e. input KW divided by
average input KVA, varies with the motor voltage and the power factor of the
motor. This means that in general the input power factor to a synchronous
motor/cycloconvertor drive will be higher than that for an induction motor and
this is shown in the graphs of Fig. 9.23.
The cycloconvertor
345
If power factor is of particular concern it is possible to use the same techniques
of power factor improvement that are used with AC/DC convertors. The most
useful one is known as sequence control where each convertor polarity is split
into two series connected bridges each being fed from a separate transformer
1-0
09
(a) unity PR synchronous motor
08
07
Z °6
a
o
vH 0.A
a5
0 0.3
a
0.2
0.1
50
percent speed
10
09
100
(b) typical induction motor
08
07
- 06
c
trapesoidal
control
1 0-5
a
o 0 A
u
a
t 0-3
sinusoidal
control
a 0-2
0-1
50
percent speed
100
Fig. 9.23 Supply power factor
winding. The output voltage across the two bridges is then the sum of the two
individual voltages and if one of them is always operated in either the full
rectification or full inversion condition the effective supply power factor is
improved. This technique allows a small improvement in supply power factor
346
The cyc/oconvertor
but it is only really practical on large drives where it is sensible and economic
to split each phase convertor into two sections.
With a normal six pulse cycloconvertor the KVAR drawn from the supply
does vary at six times the output frequency with a magnitude of cyclic variation
of approximately 13 per cent of the average KVAR being drawn. This does not
cause any problems in the operation of the drive but it can cause a low frequency
disturbance on the supply voltage, particularly if the supply capacity is not large
in relation to the drive. Because it occurs at a low frequency it can cause a
disturbing 'flicker' in the lighting loads connected to the supply and it can cause
harmonic imbalance to other connected equipment. It is therefore normal to
supply cycloconvertors from a supply system of adequate capacity.
9.6.4 Harmonics
Motor harmonics
The motor voltage waveform will always contain a proportion of harmonics
originating from the operation of the converters and these harmonics will be
related to the supply mains frequency. Fig. 3.11 shows the level of such harmonics
produced in the output voltage of a six pulse bridge convertor as a proportion
of the maximum voltage produced by the bridge. From this you will see that
when this circuit is used in a cycloconvertor the level of harmonics will vary with
proportion of harmonics
to fundamental
range of variation
approx magnitude of
harmonics in the motor
voltage
100
percent speed
Fig. 9.24 Harmonics in the motor voltage
The cycloconvertor
347
the firing delay angle and they are a maximum at 90 degrees delay, which
corresponds to zero output voltage. Under low output frequency conditions the
motor voltage is very low and the firing angle of the convertors does not alter
far away from 90 degrees. Under this condition there will be a harmonic content
in the voltage output equal to 24 per cent of the maximum convertor voltage and
this will be very much larger than the fundamental output sine wave. This is
clearly the worst case of harmonics in the motor voltage waveform and the
result is a large harmonic component in the motor current which at low loads
may become discontinuous.
As the motor frequency and voltage are increased then the range of phase
shift increases and the proportion of harmonic in the output voltages reduces
considerably. In addition, as phase shift is changing throughout the output cycle
then the proportion of harmonic alters. Fig. 9.24 shows the degree of harmonic
content in the motor waveform over the speed range of a cycloconvertor in
comparison to the fundamental motor voltage, showing that the proportion of
harmonic to fundamental reduces as the speed increases.
The simplest way to appreciate the consequences of this is to realise that there
is approximately 24 per cent of the maximum convertor voltage of six times the
supply frequency applied to the motor stator leakage inductance. This will cause
an approximately constant amount of harmonic current to flow in the motor
which will be added to the fundamental current. Study of the leakage reactance
of a number of standard motors shows that this will normally cause a harmonic
current in the motor of between 20 and 30 per cent of the rated fundamental
current. This means that at low levels of fundamental current the harmonic
content will be large and there is a strong possibility of discontinuous current.
Supply harmonics
The harmonic content in the input current to a cycloconvertor is particularly
complicated because the currents and phase convertors are continually varying
at the motor frequency rate. There will be varying degrees of cancellation
between the harmonics drawn by the three convertors and the phase position of
the harmonics is continually changing.
The detailed study of these conditions could involve extensive mathematics
and the result would not be very meaningful to practical drive students, designers
or users, so I do not propose to pursue this here.
It is possible however, to come to some general conclusions which can be
practically useful:
1) The total input current is converted by the mains commutated thyristor
switches before flowing into the motor and therefore the worst case
supply harmonics will be on the basis of a constant firing angle where
the harmonics from the three phases add together arithmetically.
With full wave convertors as described then the total input current
will contain approximately.
20 per cent of fifth harmonic
348
The cyc/oconvertor
14 per cent of seventh harmonic
9 per cent of eleventh harmonic
8 per cent of thirteenth harmonic, etc.
under the worst case condition.
2) The nearest condition to the worse case is at low speed, low voltage,
where the phase angles hardly move from 90 degrees delay.
3) Under other conditions the supply frequency related harmonics may
be less than the above but their value will be changing at a frequency
related to the motor frequency. This means that there will be other
harmonic frequencies present which are related to the motor frequency
or both motor and supply frequencies together.
Chapter 10
Slip energy recovery
10.1 Introduction
It is well known that the speed of a wound rotor induction motor can be reduced
by inserting a resistance into the rotor circuit and that this method is often used
using a variable resistance to assist in the starting of otherwise fixed speed
machines. In some cases, where the extra loss in the resistors is acceptable this
method may be used for continuous speed variation.
The result of adding the resistance is to reduce the amount of the rotor energy
which is passed into the mechanical load and this causes the slip of the motor
to increase and the speed of the rotor to reduce.
Clearly with such a method large power losses can be produced in the
additional resistors and the overall efficiency of the system will be quite low.
An alternative application of these principles is to extract power from the
rotor by another means which would enable the power to be returned to the
supply, rather than dissipating it in resistors. These more efficient methods are
referred to as slip energy recovery systems or static Kramer systems and these are
the subject of this chapter.
The overall principle of these systems is to insert a variable back emf into the
rotor circuit in such a way that the resultant energy can be recovered and fed
back into the AC mains network which is feeding the stator of the induction
motor. The result can then be an efficient method of reducing the speed of the
motor.
These principles were initially established using motor generator sets to
achieve the energy recovery and feedback, with a DC motor to absorb the
energy and an AC generator to return the power to the mains network. Fig. 10.1
shows such a system where the separate AC generator/DC motor set is run at
fixed speed, with the generator synchronised to the mains supply and variation
of the voltage on the DC miotor byfieldvariation altered the level of back emf
in the rotor circuit of the main motor and hence varied its speed. However due
to high cost and the relatively high power losses in such schemes, the principles
of slip energy recovery are now universally applied using static convertors.
This system requires the use of a wound rotor induction motor with slip rings
350
Slip energy recovery
to connect into the rotor circuit. It therefore tends to be used in custom designed
systems where the motors and convertors are specifically chosen for the application. It is used for drives in the hundreds or thousands of kilowatts ratings
where the cost of a specially designed system can be justified.
mains supply
wound rotor
induction motor
A C generator
Fig. 10.1 A Kramer drive using rotating machines
10.2 Principles of operation
Fig. 10.2 shows the basic scheme adopted in the majority of systems.
The stator is connected directly to the mains network so as to produce a
constant speed rotating field in the motor.
The rotor slip rings can usually be connected either to a starting system
consisting of resistors or to the static slip power recovery equipment and
changeover contractors are usually included for this purpose.
When slip recovery is in use the motor current is rectified by a diode bridge
and made to flow against the DC link voltage level set by the naturally commutated thyristor inverter which feeds the recovered power back to the mains
via an interposing step-up transformer.
The reactor in the DC link is provided to ensure that the DC current flows
continuously to allow the inverter to operate correctly to invert the power back
to the mains.
The principle of operation is that the stator rotating field induces voltages in
the rotor which cause rotor currents toflow.The current is rectified by the diode
bridge to produce DC and this has to pass through the thyristor inverter to
complete the DC circuit. In passing through the inverter it has to overcome the
reverse voltage set up in the inverter and hence the inverter extracts energy from
the rotor. When running, the system balances itself so that the rectified rotor
voltage equals the reverse voltage of the inverter and the rotor current being
circulated will produce the necessary load torque, which the motor has to
produce. In order to generate voltages in the rotor, it has to slow down with
Slip energy recovery
351
respect to the stator rotating field. The speed is varied by altering the DC link
voltage, using the inverter phase angle control.
In the majority of such systems the inverter is simply a naturally commutated
thyristor convertor operated in its inversion mode (see Section 3.2.2).
mains supply
feedback
transformer
induction
motor
Fig. 1 0 . 2 The static slip energy recovery system
The following points are important to the understanding of this system and
they should be noted at this stage:
1) It is usual for the feedback rectifier/inverter system to be rated at only
a fraction of the motor power rating and as a consequence it is then
only suitable to allow a modest speed reduction from the normal
motor speed.
2) The presence of the DC link reactor means that the DC current will
be reasonably smooth and continuous and therefore the rotor currents
will be of quasi-square shape instead of sinusoidal as would be the
case when the motor was running in its normal fixed speed mode.
3) The rotor frequency of an induction motor is dependent on the speed
difference between the rotor and the stator rotating field. Hence
during normal operation of the slip power recovery system the rotor
frequency is relatively low and reduces to zero at the synchronous
speed.
4) The voltage induced into the rotor windings of an induction motor is
also proportional to the speed difference between stator field and the
rotor and it reduces to zero at synchronous speed.
5) Speed control over the motor is achieved by varying the DC link
voltage by using the thyristor inverter. It is then necessary for the
rotor voltage to match the DC link voltage in order to maintain
current flow and this causes the motor to slow down or speed up as
appropriate.
352
Slip energy recovery
6) The feedback transformer is necessary for two reasons
a) The rotor voltage of an induction motor is usually chosen by
design considerations on the motor only, and
b) The level of the DC link voltage required depends on the speed
range over which the slip recovery system has to work.
7) It will be shown later in this chapter that the motor torque generated
is roughly proportional to rotor current and hence DC link current.
As a result the operation near to synchronous speed will correspond
to high convertor currents at very low voltages and the operation at
minimum speed will usually correspond to high voltages and lower
currents (assuming torque reduces with speed).
rotor frequency
speed
Fig. 10.3
Variations with speed
synchronous
speed
The basic curves of the electrical system are therefore shown in Fig. 10.3,
drawn in this case for a pump load having a specific torque speed relationship.
From these curves it can be seen that the rating of the static equipment is directly
Slip energy recovery
353
dependent on the range of speed over which they have to operate. Operation
over the full range from zero speed to synchronous speed would require a
convertor capable of accepting the full rotor standstill volts and capable of
carrying the current which would occur at top speed; this would normally be a
significantly higher power rating than the motor itself. In practice the majority
of systems are designed to allow the speed to be reduced by a fraction of the full
speed range, for example down to 70 per cent speed and hence there is an
essential need for some separate means of accelerating the motor into the range
of operation of the slip recovery equipment.
Normally this drive system is started up by conventional means such as
variable rotors resistance, either by contactor switched resistors, reostats or by
liquid resistors and when the speed has come within the correct range of
operation of the slip recovery system then the changeover switch is operated to
connect it into circuit. This means that the slip recovery system is never exposed
to the much higher voltages present in the rotor circuit at low speeds.
Study of Fig. 10.3 will also allow an appreciation of the power factor performance of the system. The thyristor inverter mains power factor will be directly
dependent on the level of DC voltage at which it is working (see Chapter 3) and
hence it will operate at zero power factor at synchronous speed and at a high
power factor at its minimum speed operating condition. The total power factor
of the system will be the result of both the input to the motor and the feedback
from the static recovery system. Hence this will be relatively good at the
minimum speed point but the power factor at near to synchronous speed will be
quite a bit lower than that which would be obtained with the motor operating
without the recovery convertor connected (for further details see Section 10.6.2).
The operating efficiency however is good over the whole of the speed range.
The only additional losses are those associated with the convertors and these are
usually small compared to the amount of energy saved using this system.
The convertor losses are however dependent on current and therefore they
will be present even when, at near to full speed, very little feedback of energy
is taking place.
The overall viability of this scheme is clearly dependent on the amount of time
for which the speed of the motor is to be reduced. If the motor runs for most
of its time at reduced speed then the saving of energy which can be made using
this system may be substantial. If however the drive only runs occasionally at
reduced speed, then the convertor losses and the poor power factor at high speed
may outweigh the schemes advantages.
The square wave currents which flow in the rotor winding cause some reduction in the performance of the motor. The total RMS value of the current at any
specific torque and speed condition will be increased above the equivalent
sinusoidal value and therefore some derating in thermal performance is needed.
In addition the result of the harmonics contained in the rotor current is the
introduction of a small amount of harmonic torque ripple, which can set off
mechanical resonances into the load system.
354
Slip energy recovery
The current injected back into the supply by the feedback converter is also
square in shape and contains harmonics. However, as the KVA feedback is
usually small compared with that drawn by the motor the level of harmonics is
usually quite acceptable.
The system as being described here is only capable of reducing the speed down
from the synchronous value by drawing energy from the rotor. Operation at
higher speeds than synchronous would require energy to be fed into the rotor
and it is not possible to do this with a diode rectifier. Although modified circuits
have been proposed to allow operation at above synchronous speed, the majority
of practical systems in operation are of the type being described here.
The slip recovery drive is not capable of reversing the direction of the motor;
this is set by the stator phase connections and by the direction of rotation of the
stator field.
In this system the motor conditions i.e. itsfluxand operating power factor etc.
are set by the mains supply connected to the stator. The only features of this slip
recovery system which directly affects the motor's electrical conditions are the
higher slip speed under which it works and the presence of harmonics in the
rotor current.
Starting
As most of the slip recovery drives in service are only rated for a limited speed
range they usually include another means of starting. The normal arrangement
is to connect a set of variable resistors to the rotor slip rings and then to run the
motor up in the normal way reducing the rotor resistance as the speed increases.
When the rotor voltage has reduced to a value which is within the rating of the
slip recovery diodes and thyristors, then a changeover to this system can occur.
Either a voltage or frequency measurement can be used to initiate the switchover. It is normal to release the pulses on the inverter bridge after switchover,
with the pulses phased to give a higher DC link voltage than would be expected
at the switchover speed. A gradual advance of the firing angle will then lead to
the current starting to flow in the DC link and torque to be developed in the
motor.
Once running, variation of the DC link voltage using the inverter causes the
motor speed to alter. The minimum speed will be dictated by the maximum
voltage which the inverter has been designed to operate at and the maximum
speed will be synchronous speed.
If operation at synchronous speeds for prolonged periods is expected, the
system may contain means to short circuit the slip rings so as to avoid the losses
in the convertor equipment. Clearly if speed reduction is later required then the
slip recovery convertor needs to be switched back in.
10.3 Detailed analysis of the system
This system naturally splits itself into three main areas:
1) The induction motor itself connected to the fixed frequency supply.
Slip energy recovery
355
2) The diode rectifier in the rotor, operating at rotor frequency.
3) The naturally commutated inverter operating at mains frequency.
The diode rectifier and the inverter are effectively isolated from each other by
the presence of the DC link reactor and it is the DC link which effectively ties
the three areas together.
In order to study it further let us assume that the motor has already been
accelerated up to high speed and the slip recovery system has been brought into
operation, so that the current isflowingin the DC link and the inverter is feeding
power back into the mains supply.
10.3.1 Circuit waveforms
Rotor and diode rectifier conditions
If we assume that the DC link inductance is high, then steady DC current will
beflowingin the DC link. This current originates from the rotor windings and
is rectified by the rotor diode bridge. The rotor frequency will be low and will
depend on the actual running speed compared to the rotor stator rotating field.
rotor voltages - at say 5 hertz
ph lse rotor
jrrent
overlap
period
7
\
DC voltage
Fig. 10.4 Rotor waveforms
The conditions in the rotor and rectifier will therefore be quite conventional
except that the operating frequency will be low. The waveforms of Fig. 10.4
show the 3 phase rotor voltages, the rotor currents and the DC voltage which
would typically occur under these conditions. The rotor voltage level will also
356
Slip energy recovery
depend on the slip speed and the induced voltage waveforms will be sinusoidal
due to the characteristics of the induction motor as described in Chapter 1.
The overlap distortion produced on the slip ring voltage waveform will be due
to the leakage reactance of the rotor and the reflected effect of the stator and it
is caused by the diodes temporarily shorting out two rotor windings, while the
current is transferring between them.
In the diagram the overlap angle is shown to be relatively long at around
30 electrical degrees making the rotor current waveform trapesoidal in shape.
This overlap angle depends on the magnitude of the effective inductance of the
rotor circuit and the value which is available to cause the transfer of the current
from one diode arm to the next. Most wound rotors have a value of inductance
which causes overlap angles in the 20 to 40 electrical degree range.
In practice the shape of the rotor current waveform does not alter very much
as the speed is changed because, although the voltage in the rotor increases as
the speed reduces, the frequency also increases so that the overlap time represents
a similar electrical angle over a wide speed range. The magnitude of the current
does alter the overlap angle so that at low levels of current the waveform is more
square in shape.
As indicated in Chapter 1 the induced voltage in an induction motor is
generally sinusoidal in shape and this is certainly true in all operating conditions
of this system, leading to the slip ring voltage waveform as shown in the figure.
The DC link voltage on the rotor side of the reactor is a typical diode rectifier
shape, with a main ripple at six times the rotor frequency, having a peak to peak
ripple magnitude of between 13 and 20 per cent of the peak voltage depending
on the overlap and the current level.
Motor stator current waveforms
The motor stator current is the sum of the magnetising current required to
generate the necessary back emf plus the transformed rotor current. The
magnetising current will be generally sinusoidal but the rotor current will
contain a small amount of harmonics which will produce some distortion in the
stator current.
The situation is somewhat complex due to the speed difference between the
rotor and stator rotating field. For example the fifth harmonic in the rotor
current will cause an air gap field rotating around the rotor at five times rotor
frequency. This will interact with the stator rotating field rotating at 50 or
60 hertz frequencies. As the speed is changed then the rotor frequency and the
fifth harmonic frequency is changed while the stator frequency remains constant. This is clearly a fruitful area for complex mathematical calculation and
assessment.
In practice however the result is a relatively small amount of waveform
distortion and the possibility of some degree of low frequency beating effects
around specific speeds related to the rotor harmonics. For all practical purposes
the stator current waveforms are very close to sinusoidal in shape.
Slip enerav recovery
357
Inverter waveforms
The conditions in the feedback inverter are almost identical to those explained
in Section 3.2.2, with the DC current being chopped up into quasi-square wave
AC currents and the ripple occurring on the DC voltage being dependent on the
firing angle of the thyristors.
The only real difference is in the specific conditions of use in the slip recovery
system. With most loads the maximum speed condition occurs when the current
is the highest and this corresponds to the point where the voltage on the DC link
is zero. The inverter thyristors therefore operate at a delay angle of 90 degrees
with maximum DC voltage ripple at this point. At the low end of the speed
range the inverter will be operating at its maximum inverting voltage and the
DC voltage ripple from the inverter will be lower. Usually the level of load
current required at the lower speeds is reduced. The ripple in the DC voltage
always occurs at six times the mains frequency.
R.M.S harmonic in
inverter voltage
at 6x mains frequency
30
voltage
J2
20
c
o
1 01— /
/
RMS harmonic
voltage from rotor
at 6xrotor frequency
a
si
top speed
minimum speed
i
o
20*/o speed range
due to '
rotor /
diode \
rectifier
f
y/ ^
Q.
a
A
current
50°/.
to inverter
C/3
cc
Fig. 10.5 Harmonic effects on the DC link current
The DC link current
The assumption of a smooth and steady level of DC current is only true if a large
DC link reactor is employed. In practice this is rarely the case and the ripple
358
Slip energy recovery
voltages from both the motor rotor and the feedback inverter usually leave their
mark on the DC current. In assessing this it is necessary to appreciate that the
sum of the ripple voltage from the rotor rectifier and that from the inverter,
appear across the DC link reactor. The rotor ripple is only approximately six per
cent RMS of the DC link voltage (which reduces as the speed increases) at a
varying frequency of six times the rotor slip frequency, whereas the inverter
ripple is at a constant frequency but its magnitude reduces with the speed. These
two effects are shown on Fig. 10.5 and this may imply that the inverter ripple
is dominant. This is not however the case; they can both cause similar proportions of harmonic current in the link due to the frequency difference.The ripple
current at any one frequency will be given by:
Iripple
=
V r i p p l e /(2 X U X fripple X L )
where L is the inductance of the DC link reactor.
What this tells us is that the ripple current caused by the rotor is of approximately constant magnitude, whereas the ripple caused by the inverter increases
with the speed. The current graphs show that the inverter ripple is dominant
only if the speed range is greater than 25 per cent and that at the lower speeds
the rotor ripple is more important.
The magnitude of the ripple current is not affected significantly by the level
of meairDC current flowing, so that the ripple is more noticeable at low levels
of current and torque.
10.3.2 Motor equivalent circuit
When the total system is looked at from the motor stator side, then the most
useful technique to aid in its understanding is the equivalent circuit. Clearly
there will be many similarities between the equivalent circuit of this system and
a normal induction motor, so reference to Chapter 1 may be useful here.
The single phase equivalent circuit of this system is shown in Fig. 10.6(a)
which shows the addition of the rotor diodes and the back emf caused by the
inverter.
The addition of these components makes the operation of the motor fundamentally different, in that current cannot flow in the rotor circuit at all unless
the voltage generated in the rotor exceeds the additional back emf from the
inverter. Therefore current flow isv dependent on the necessary generation of
rotor voltages and this is only produced by the rotor slowing down in order to
increase the slip speed.
Once sufficient voltage is being produced current willflowin the rotor circuit,
limited only by the resistances and inductances in the current flow path — the
diodes cease to have real relevance to the circuit's operation. They do cause the
rotor currents to be of quasi-square wave shape but as the torque in the motor
is generated by theflowof the fundamental rotor current, the harmonics will be
ignored for the present.
With these assumptions then, the equivalent circuit of the motor when current
Slip energy recovery
359
is flowing can be reduced to that of Fig. 10.6(b) where the diodes have been
removed and the DC voltage replaced by an equivalent AC voltage Vr which
always has to*be in phase with the current flowing. The circuit resistances and
reactances have been lumped together into R2' and L2'.
The final simplification of the circuit is carried out in the same way as with
a normal motor resulting in Fig. 10.6(c) where the rotor circuit has been
converted from one operating at slip frequency to one working at the stator
frequency.
R1
VI
LI
F1
11 R1
El
*l
VI
F1
R2'/sl
L2
LI
t ^mag
v
E1
^ ^
F1
nJ
Fig. 10.6 Equivalent circuits
The effect of adding the slip recovery convertor system to the motor is
therefore to introduce an additional rotor voltage equal to — Vr/Sl, this voltage
always being in phase with the rotor current (SI is the slip of the motor related
to the normal synchronous speed).
An important point which has a great effect on the understanding of this
system is that the magnitude of this additional rotor voltage is approximately
equal to the rotor referred induced voltage El under most normal operating
conditions. At the point where the rotor diodes are just overcome by the induced
rotor voltage, then Vr/Sl will be equal to El and no rotor current willflow.As
torque is applied the slip increases thus reducing Vr/Sl so that there is a
difference voltage in the rotor to allow the rotor current to increase. This
360
Slip energy recovery
situation is true whatever the value of the feedback voltage because the critical
slip rises in proportion to this value.
The other effect which should be appreciated is that the effective values of
rotor resistance and inductance have now increased because they must take
account of the complete flow path of the rotor current and this includes the
diode rectifier, the DC link, the inverter and the feedback transformer (assuming
that the value of Vr does not already take these into account).
So now Fig. 10.6(c) represents the motor equivalent circuit under all conditions
while rotor current isflowing.This means that it covers all conditions where the
rotor induced voltage is sufficient to overcome the back voltage from the
inverter. In this equivalent circuit, this means all conditions where Vr/Sl < El,
i.e. conditions where SI > Vr/El.
This large value of slip also causes the rotor resistance R2 to be less dominant
and to have less direct effect on the rotor current.
-11X1
-I1R1
11
Vr/Sl
flux
Fig. 10.7 The vector diagram
10.3.3 The motor vector diagram
The vect6r diagram of the motor expresses the relationships between the fundamental sinusoidal currents and voltages and is best developed from the simplified
equivalent circuit as above. It has many identical features to that for a motor
fed from a more normal source as explained in Chapter 1, the differences being
the rotor conditions.
Fig. 10.7 shows a typical vector diagram of this scheme showing the additional
voltage vector Vr/Sl which reduces the effective voltage in the rotor circuit to E2.
This additional vector is always very similar in length to El, because the slip has
to increase in proportion to the feedback voltage before any rotor current can
flow at all e.g. if an equivalent feedback voltage of 20 per cent of rotor voltage
Slip energy recovery
361
is applied then the slip will have to increase to 0-2 to allow current to flow, if
a 30 per cent voltage feedback is used then the slip has to increase to 0*3 etc.
Although this may appear to cause a dramatic change in the effective rotor
voltage to Er, it should be remembered that the effective rotor resistance is also
affected by the slip, so it reduces also as the feedback voltage is increased.
The remaining features of the vector diagram are relatively unaffected. The
stator current II is the result of the sum of the rotor current and the magnetisation and iron loss currents and VI differs from the induced voltage by the
small voltage drops in the stator resistance and leakage reactance.
10.3.4 Circuit equations and relationships
We can now consider how all the parameters of the total motor/rectifier/inverter
system are related in the operation of this drive.
The first important fact is that the system only produces motor torque if the
induced rotor voltage (when rectified by the rotor diodes) exceeds the DC
voltage set by the inverter. The slip of the motor increases until this critical value
is reached. If the open circuit rotor voltage induced in the motor at standstill
when normal mains supply voltage and frequency are applied to its stator is
E2max, then this critical value of slip is given by:
Critical slip = Vdc/(l-35 x E2max)
(1)
Only slip values above this will cause torque to be generated, values below this
are of no importance.
Let us now consider the other parameters with reference to Fig. 10.2, which
define some of the parameters with which we are concerned.
Let us assume that the drive is running at a speed below the critical slip point
and that rotor currents areflowingto generate the required torque. Current will
then beflowingin the DC link and through the inverter, which will return some
of the power passed from stator to rotor back into the mains supply.
If we start with the inverter and initially neglect the effects of inverter or
transformer resistance and reactance, then the mean value of the DC voltage
will be given by the following, if smooth and continuous DC current is flowing:
Vdc = 1-35 x Vt x COS (Beta)
(2)
where Vt is the transformer secondary RMS line voltage and Beta is the inverter
firing advance angle.
In practice there will be some voltage drop in the transformer and inverter
resistance and reactance and reference back to Chapter 3 will also show that the
DC voltage is actually given by:
Vdc = 1-35 x Vt x COS (Beta) + 1-35 x Vt x Xt/2
+ Idc x R + 2 x Vth
where Xt equals the per unit circuit reactance
(2a)
362
Slip energy recovery
R equals the equivalent DC resistance of the circuit
Vth equals the voltage drop in the thyristors.
If the current in the DC link is equal to Idc mean then the power being fed back
by the inverter is given by:
Feedback power = Idc x Vdc
= 1-35 x Vt x Idc x COS (Beta) approximately
(3)
Now let us look at the rotor side of the DC link. The currentflowingin the DC
link also flows in the rotor windings and reference to Chapter 3 will show that
the rotor line current will be given by:
Ir = Idc x 0-816 amps RMS
(4)
This is the total RMS value and the current waveshape will be trapesoidal. So
the fundamental value of the current, the value which produces the motor
torque, will be slightly lower at say:
12 = Ir/1-05
12 = 0-78 x Idc
(5)
As the DC link current is being forced by the rotor voltages, the DC link
voltage on the rotor side will be slightly above that on the inverter side due to
the small resistance of the DC link reactor. In addition it is necessary to
overcome the forward drop of the diodes.
The slip ring phase voltage can therefore be given by:
Vr x y/3 x 1-35 = Vdc + 2VD + Idc x Rdc
where VD = the forward voltage drop of a diode arm, and
Rdc = the resistance of the DC link
i.e.
Vr = (Vdc + 2VD + Idc x Rdc)/(l-35 x 1-732)
(6)
We can now insert this value into the equivalent circuit of the motor in order
to proceed with the analysis.
Let us now therefore refer to Fig. 10.6(c) and to the vector diagram of
Fig. 10.7.
The rotor current 12 is the result of the difference voltage between the induced
voltage El and the feedback voltage Vr/Sl, i.e. Er acting on the circuit
impedance Z2 where
Z2 =
2
V/(R27S1) +
{XT)2
E2 = 12 x Z2
and angle
An3 = ATN{(X2' x Sl)/R2'}
(7)
Sfip energy recovery
363
Once the value of the slip has been established the vector diagram can be
solved by normal geometrical means to establish the other parameters in a
similar way to that carried out in Chapter 1.
The way of establishing the slip and therefore the speed is to consider the
power relationship in the rotor.
The total power per phase in the rotor is given by:
Pr = (I2)2 x R27S1 + 12 x Vr/Sl
= (I2)2 x R2' + (I2)2 x R2' x (1 - S1)/S1 + 12 x Vr/Sl
and this must equal Rotor resistance loss + Mechanical power + Feedback
power.
The total feedback power when looked at from the rotor equals the rectified
value of the rotor current (from equation (5)) multiplied by the rectified value
of the rotor volts Vr, i.e.
Feedback power = I2/-78 x Vr x 1-35 x 1-732
= 12 x Vr x 3.0
i.e. the feedback power per phase equals 12 x Vr. Therefore the total mechanical power per phase equals
Pm = {(I2)2 x R2' + 12 x Vr} x (1 - S1)/S1
Therefore the total motor mechanical power
= 3 x {(I2)2 x R2' + 12 x Vr} x (1 - S1)/S1
(8)
This power is also related to the torque by:
Pm = (2 x PI x Speed x Torque)/60
(9)
where speed is in RPM and torque in Newton metres, and
Speed = 120 x Fl/P(l - SI) RPM
(10)
where Fl is the stator frequency in hertz and P is the number of motor poles.
From these equations it is possible to establish the values of the parameters
under a specific set of operating conditions with a motor and convertor system
with specific constants, etc.
The curves of Fig. 10.8 and 10.9 show the resulting characteristics as torque
is varied with a specific feedback voltage and the relationships between speed
and torque at different values of feedback voltage. These results were obtained
from a 55 KW, 415 volt, 50 hertz, 4 pole system with a rated torque of 372
Newton metres at 1480 RPM.
Fig. 10.8 is drawn for a constant feedback voltage of 50 volts/phase —
referred to the stator. From this you can see that rotor current is almost
proportional to torque and the amount of power fed back is also directly
proportional to the torque. With a constant feedback voltage the speed reduces
364
Slip energy recovery
slightly as torque is applied in a similar way to a normal induction motor. In
practical systems the speed is maintained constant by varying the feedback
voltage and Fig. 10.9 shows the relationship between speed, feedback voltage
and torque.
150 -
1300
30
2 1200
20
1100
10
Q_
200
Fig. 10.8 Variation with torque
400
600
800
torque- N.M.'s
1000
1500
speed at zero torque
/
1000 >
^ N
/
speed at rated torque
500
Fig. 10.9 Variation of link volts
i
100
i
200
DC link voltage
1
300
Slip energy recovery
365
10.3.5 Examples of calculations
1) Calculation of ratings
Question
A 750 KW wound rotor induction motor has a standstill open circuit slip ring
voltage of 1000 volts AC, RMS line when supplied from its normal stator rated
voltage and frequency. At its full speed, rated torque short circuited slip ring
condition, it carries a rotor current of 450 amps RMS line. Decide the approximate ratings of a slip power recovery diode rectifier and inverter system capable
of reducing the speed to 70 per cent of the synchronous speed.
Answer
At zero rotor speed the rotor line voltage equals 1000 volts. At synchronous
rotor speed this voltage will be zero. At 70 per cent of synchronous speed the
rotor induced voltage will be
1000 x
30
T100
HH = 3 0 ° v o l t s A C l i n e
From equation (6), ignoring the circuit resistance but allowing 2 volts voltage
drop per diode we get
Vdc = 1-35 x 300 - 4 = 401 volts DC
When runnning at or near to full speed the rotor fundamental current could be
at the rated 450 amp value as given.
Then from equation (5) the DC current will be given by:
Idc = 450/78 = 577 amps mean.
The slip power recovery convertor therefore has to be rated at
401 volts DC and 577 amps DC
i.e. its KW rating will be 231-4 KW
Question
If the rotor current at the 70 per cent speed had a fundamental value of 100 amps
and it varied linearly with speed up to 450 amps at synchronous speed, what
value of power would be fed back from the DC link at the 70 per cent speed and
what would be the highest power fed back over the 70 to 100 speed range.
DC voltage at 70 per cent speed as above = 401 volts DC
From equation (5)
Idc at 70 per cent speed = 100/78 = 128-2 amps.
Therefore power fed back
= 401 x 126-6 = 51.4 KW
366
Slip energy recovery
The application of simple mathematics to a situation where the current and
voltage both vary linearly, one rising and the other falling with the voltage
reaching zero at maximum speed, can show that the maximum power will occur
when the fundamental rotor current equals 225 amps and this occurs when the
speed is 80-7 per cent of maximum.
At this point the rotor voltage will be equal to
1000 x ^
1UU
= 193 volts
Therefore Vdc = 1-35 x 103 — 4 = 257 volts. Therefore maximum power fed
back from the DC link equals
Maximum power = 257 x 225/-78 = 741 KW
2) Rotor conditions
Question
If the rated slip of the above 4 pole, 50 hertz motor is 3-2 per cent and the
rotor referred inductance is 0-2 mH. Find the DC voltage and DC current
flowing when the speed is 80 per cent of synchronous speed and the torque is
2000 Newton metres. Ignore the stator resistance and leakage inductance.
Answer
From equation (8) at rated load and with Vr at zero
750,000 = 3 x (450 x 450 x R2' x (96-8)/(3-2))
R2' = 750,000 x 3-2/(3 x 450 x 450 x 96.8)
= 04 ohms
At 80 per cent of synchronous speed the slip equals 0*2 per unit. As the motor
is a 4 pole 50 hertz motor then the synchronous speed equals 1500 RPM. From
equation (9)
Pm = (2 x PI x 0-8 x 1500 x 2000)/60
= 251,327 watts
From equation (8)
251,327 = 3 x {(I2)2 x -04 + 12 x Vr} x 4
.*. {(I2)2 x -04 + 12 x Vr} = 20,944
If we now refer to the vector diagram Fig. 10.7
El = 1000/1-732 = 577
and Vr/Sl will be slightly less than this value i.e.
Vr equals approximately 577 x -2 = 115 volts.
Slip energy recovery
367
Let us assume Vr = 110. From above therefore
{(I2)2 x -04 + 12 x 110} = 20944
This gives
12 = {-110 x j\10
x 110 + 4 x 04 x 20944}/08
= 178-8 amps
Therefore from equation (5)
E2 = 178-8 x ^/(-04/-2)2 + (2 x PI x 50 x -2/1000)2
= 37-5 volts
and
angle An3 = ATN(-0628 x -2/04)
= 17-43 degrees
Geometric study of the vector diagram will then give
Vr = 111 volts, because
El x SIN An2 = E2 x SIN An3
If we take this value of Vr then from above
12 = 177-4 amps fundamental
from equation (5)
Idc = 224-6 amps mean
and from equation (6) approximately
Vdc = 111 x 1-732 x 1-35 - 4
= 255 volts.
10.4 Practical circuit designs
As this drive system is naturally commutated i.e. switched using the reversing
sinusoidal rotor voltages and main supply voltages to effect switching between
the diode and thyristor switches, the majority of the additional items required
are associated with protection and cooling etc. The components need to be
protected against any overvoltages which may occur and against fault current
which may arise due to circuit maloperation.
However before dealing with these we should consider the situation regarding
the start up of such drives. Because it is often uneconomic to design the feedback
convertor system to accept the full standstill rotor voltage it is not normally
368
Slip energy recovery
possible to have the converter connected during the starting period. To achieve
satisfactory starting torque, the rotor slip rings are usually connected to a
variable resistance which will enable the motor to operate at high slip values and
with acceptable levels of current. The resistance may be infixedvalues switched
with contactors or it may be a fully variable liquid type resistor. This will usually
be connected to the slip rings via a changeover switch which will be used to bring
the convertor into operation. The drawing in Fig. 10.10 shows a typical arrangement for a starting and switchover scheme. Whenfirststarting, the motor switch
SW1 connects the starting resistor to the slip rings. When the motor has
accelerated up to the appropriate speed to suit the convertor, the slip frequency
detector initiates changeover of SW1, as long as SW2 is closed. The convertor
is now connected and the inverter firing angle can now be increased from its
initially low value in order to allow current to flow in the rotor, DC link and
inverter.
starting
resistor
isolator
5W2
SW1
feedback
rectifier
and
inverter
etc.
rotor
control
slip
frequency
detector
control
rotor
shorting
switch
Fig. 10.10 Switching and contactor arrangements
This diagram also shows an additional feature of such schemes, a rotor
shorting switch which is used for running at top speed without the convertor
connected. If running at full speed for long periods of time is anticipated, then
the use of this switch will allow the most efficient running condition as no
convertor losses will then be produced, the convertor being disconnected
automatically by SW2.
10 A A Overcurrent protection
Under normal circumstances the voltages produced at either side of the DC link
reactor are equal and balanced, the current circulating being the result of a small
difference between them. If ever this balance gets seriously disturbed, high
Slip energy recovery
369
currents can circulate in the system. As the whole of the DC link voltage is
directly dependent on the speed of the motor, the worst conditions for fault
current occur at minimum speed where the rotor voltage is at its maximum
value. This is also the point at which there is maximum dependence on the
supply voltages. If there is any disturbance in this supply voltage, the resulting
surge of current in the DC link and rotor circuit may be sufficient to cause
commutation failure in the inverter and then the inverter appears as a direct
short circuit on the DC link, causing large currents to flow. The direct result is
to cause the motor to accelerate up to full speed in a DOL manner and the high
rotor currents are likely to damage the convertor unless specific allowance has
been made in the choice of the components.
In most cases the current will be too high and protective fuses or circuit
breakers will be used to prevent damage to the more important components in
the system. If this happens, the rotor circuit will be effectively open circuited and
torque generation will immediately cease, causing the motor to coast down to
rest under the influence of the load. All this can be caused by a small disturbance
in the AC mains sine waves. Hence in well designed systems a margin will be
built into the convertor and the transformer ratio, to allow a modest reduction
in voltage to occur while running at the minimum speed in order to reduce the
risk of inversion failure.
Clearly the presence of the DC link reactor is a help in this situation because
it reduces the rate of rise of current and it therefore increases the allowable size
of disturbances which may be allowed to occur, without causing inversion
failure.
This problem of supply disturbances has led to the investigation of a number
of more complicated solutions like the provision of forced commutation on the
feedback inverter or the use of static switching on the DC link to remove the
fault very quickly. However these methods tend to add complexity and cost to
an otherwise simple system and none have proved to be practical enough to be
applied widely. The most common approach is to include safety margins on the
current ratings of the semiconductors and to include a DC circuit breaker to
clear the fault.
10.4.2 Overvoltage protection
When the slip power recovery convertor has been chosen only for use over a
limited range of speed, some means must be included to ensure that under no
circumstances can the convertor remain connected to the motor at lower speeds,
where the rotor voltage will be in excess of its capability. This may take the form
of a slip frequency or rotor voltage detector to initiate the opening of the rotor
switch and the introduction of the rotor resistance. A back up measurement
based on speed can be included if it is felt that the other electrical measurements
could fail.
As the feedback convertor is directly connected to the mains supply, probably
via a transformer, it is necessary to cater for the variations and transients which
370
Slip energy recovery
can occur on the mains voltage. Snubber circuits and surge suppression devices
will be needed in the inverter to ensure that the thyristor switches can cope at
all times.
10A3 Circuit variations
The main limitations of this drive system is the low overall power factor (see
Section 10.6.2) and the poor utilisation of the convertor.
The power fed back via the convertor is at a power factor proportional to the
DC link voltage. At the higher speeds (where the torque and current are often
higher) the power factor is very low and the appropriate reduction of overall
power factor takes place.
The convertor has to be normally rated for the maximum circulated current
which usually occurs at the top speed (when the DC volts are very low) and for
the maximum rotor voltage (which will occur at the minimum speed). Therefore
the maximum volts and current occur at opposite ends of the range and
hence the convertor capability is usually well in excess of the actual power fed
back.
Fig. 10.11 (a) shows the typical conditions related to the feedback convertor
(in this case for a fan load).
The power factor in a simple arrangement is proportional to the DC link
volts. If the transformer ratio is changed at, say, the halfway point in the speed
range, the feedback power factor at this point could be improved from say 0-5
to 0-9 per unit and at the same time the current in the primary winding would
reduce so lowering the supply KVAR. Fig. 10.1 l(b) shows what the result would
be if the ratio was changed by a factor of two at midway in the speed range. In
this case KVAR is halved, the feedback KW being unchanged.
In practice this has to be done by having a half voltage tap on the inverter
transformer secondary or by a delta-star transformation on the primary side.
A similar improvement can be made using dual inverter bridges, connecting
them in series at the low speeds and in parallel at the high speeds and this
approach has the additional benefit of increasing the utilisation of the inverter
thyristors. Fig. 10.12 shows a typical double bridge inverter arrangement with
switching to allow the series/parallel operation of the bridges.
10.5 Overall control methods
With this drive system there is only one controllable variable and that is the
phase angle offiringof the inverter thyristors. This directly controls the level of
the DC voltage and hence the speed of the motor at which torque can be
developed.
In addition, reference to Fig. 10.9 shows that at any voltage setting, variation
of load torque only produces a small change in speed and as a result the system
is inherently stable with only minor alterations of the DC voltage values being
Slip energy recovery
mm
speed
Fig. 10.11 Transformer ratio changing
synchronous
speed
371
372
Slip energy recovery
necessary to maintain a steady working condition against normal supply, load
and temperature variations.
At all speeds the torque developed in the motor is directly dependent on the
rotor current and hence on the DC link and inverter current and so a measurement of this value can be reasonably used if control of the load torque is desired.
In most practical cases the speed torque characteristic of the load is decisive in
deciding the torque which must be generated and the only way torque can be
controlled is by altering the DC voltage and hence the speed.
inverter
feedback
transformers
Fig. 10.12 Series/parallel inverter switching
Hence the inverter voltage controls the motor speed and it can only do this
within its range of designed voltage. Maximum voltage will correspond to
minimum speed and zero voltage will cause maximum synchronous speed. If the
motor is overloaded causing excessive currents, all the inverter can do is to lower
the speed to its lower limit. If this is insufficient to reduce the current, then it will
be necessary to operate the overcurrent protection systems employed.
However, study of Fig. 10.9 will show that only small changes in the DC
voltage would cause large changes of torque and current and this would cause
severe acceleration conditions to occur. It is therefore usual to incorporate a
current limiting system in the inverter, so as to reduce the accelerating torque
and current to acceptable levels. The current limiting system effectively prevents
the fast reduction of the voltage.
Under speed reduction conditions, the current will immediately reduce to zero
as soon as the DC voltage is increased and the motor will slow down in response
to the load, until the new speed set point is reached.
The control systems normally employed for the slip energy recovery scheme
are therefore quite simple and Fig. 10.13 shows a typical example having a high
speed inner current loop based on a rectified AC current measurement and an
Slip energy recovery
373
outer speed or DC link voltage loop. If a speed control loop with a tachogenerator is employed then the speed will be kept constant irrespective of load
and supply changes. If a voltage loop is used then load changes will not be
compensated for and some speed change with load could be expected. However
this is usually quite acceptable and the ability to work without a small tachogenerator on a large motor may be beneficial.
rectifier
inverter
tttttt
motor
firing
circuits
mesurement
tacho -generator
for alternative
speed control
speed
setting ;
ramp
current
amplifier
voltage
/speed
amplifier
<§>
feedback
transformer
current
measurement
r
signal
limit
Fig. 10.13 Slip power recovery control system
10.6 Performance and application
This drive gives very efficient, limited speed range performance and has been
applied most widely to pumps because of the narrow speed range which is then
required. Where the speed range is only 20 to 40 per cent of synchronous speed
this drive can be economic in first cost compared to other systems.
It has a number of disadvantages which have restricted its widespread
application.
1) It requires a wound rotor slip ring induction motor and these are
usually made specially to order.
2) Its overall power factor is quite low under the rated operating
condition.
3) It is necessary to provide conventional starting equipment and
switches, etc.
4) It is susceptable to supply disturbances.
However, it is a simple drive, which is very sU ble in operation and which
saves considerable energy compared with the long established variable rotor
resistance method of control.
10.6.1 Efficiency
The main advantage of this scheme is the overall high efficiency which results
from recovering the rotor energy not needed to satisfy the load. Under rotor
374
Slip energy reco very
resistance control, this energy would be dissipated in heat in the resistance. The
result is that the efficiency of the drive remains high over the complete speed
range.
There will be a small loss of energy in the rotor diodes, in the inverter and the
feedback transformer and there will also be a slightly increased motor loss due
to the harmonic content of the rotor current. But, in a well designed system these
will only be a small proportion of the total energy saved. Although this is true
in general the position at the high end of the speed range is not quite so good.
At this point the circuit current is usually at its maximum and hence these
additional losses are at their highest level. The feedback power at this point will
however be very small and hence the additional losses may in fact exceed the
feedback power resulting in an overall loss of energy.
The actual energy saved depends on the torque/speed characteristic of the
load as mentioned previously and Fig. 10.14 shows the way in which these may
vary over the speed range for a constant torque and a variable torque load.
When considering the use of this scheme it is usually the energy saving which
has to pay for the extra cost of the convertor equipment. Some knowledge of
the likely load duty is needed, i.e. the periods of time at different speed and load
torque setting which are expected, before an accurate estimate of the energy and
therefore cost savings which can be made.
10.6.2 Power factor
Unfortunately the total input power factor to this system i.e. including both the
input current to the motor stator and the supply current of the feedback
convertor, will always be lower than the motor would be in its normal fixed
speed state. The feedback inverter always draws a lagging KVAR from the
supply system and increases the total lagging KVAR of the combined system.
At the same time the total KW drawn from the supply will be reduced by the
rotor power and fed back through the slip recovery inverter. The combined
reduction of KW and increase in KVAR will always therefore reduce the
operating power factor.
The amount of the reduction depends on the size of the feedback convertor
and the range of speed over which it can operate.
The general situation is shown by the vector diagram of Fig. 10.15. This
represents the total input condition to a slip recovery system which is operating
over a 100 per cent to 53 per cent speed range on constant torque. OA represents
the stator input vector and this is almost constant throughout the speed range.
AB represents the input current to the feedback convertor at the maximum
speed when no power is being recovered, OB is then the total vector under this
condition. AC represents the minimum speed input current to the convertor and
hence OC represents the minimum speed condition. The curve BC therefore
represents the locus of the total current vector as the speed is changed.
The length of the AB and AC vectors depends only on the size of the feedback
convertor and hence the available speed range. If the convertor is small then the
extra convertor losses
(b) variable torque
Fig. 10.14
Energy saving
synchronus
speed
376
Slip energy recovery
feedback
convertor
vectors
total
maximum
speed input
current
total
minimum
speed input
current
KVAR
Fig. 1 0 . 1 5 Constant torque input vector diagram to a slip power recovery system
area in which
total current
vector c a n be as
torque varies
KVAR
Fig. 1 0 . 1 6 Range of power factor variation
Slip energy recovery
377
(a) at rated motor torque
range of speed control
10 -
0-8
06
02
50
percent synchronous speed
100
(b) at mid- range speed
range of speed control
I
c 08
I
o
o 06
50
100
percent rated torque
Fig. 10.17 Variation of total input power factor
reduction in power factor will be small; if the convertor is large the reduction
will be large. In the limit with a full size feedback convertor capable of allowing
the full speed range to be covered, the vectors AB and AC will be almost equal
to OA and the overall power factor will be low at all times and could approach
zero at very low speeds.
378
Slip energy recovery
The chart of Fig. 10.15 is drawn for constant torque conditions over the speed
range but it can in fact represent any condition of load if the scale is changed,
because if the torque is reduced so the length of all the vectors reduces. The OA
vector will follow the dotted motor circle line XY and the lengths of AB and AC
will depend on the rotor and DC link currents which will reduce with torque.
The chart of Fig. 10.16 has been drawn to show the conditions for any load and
speed for a system with a 100 to 70 per cent speed range. The shaded area shows
the limited area in which the total supply current vector an be and the diagram
OPQR shows the general case at approximately 60 per cent torque showing the
locus of the vector as the speed is changed. OB represents the full speed rated
condition and ON is the zero torque condition. Fig. 10.17 shows the overall
system power factor for a drive with different size feedback convertors compared
to the original fixed speed power factor of the motor.
10.63 Torque capability
As the feedback convertor has to be capable of carrying the full rotor current
when the motor is running near to its top speed then the convertor system will
be capable of full current over the whole of the designed speed range and hence
the slip power recovery drive is essentially a constant torque drive capable of
accepting the full torque at any point over the speed range.
However this may not be within the capability of the motor due to the reduced
cooling which may be the result of speed reduction. If continuous operation at
full torque and current at the lower speeds is expected, then the motor cooling
will have to be checked to ensure that it is capable of sustaining this without the
temperature of the motor being exceeded.
10.6.4 Harmonics in the system
Due to the operation of the diode rectifier and the feedback inverter, there will
be additional harmonics in the rotor of the motor and in the electrical supply
to the inverter.
The rotor currents will be quasi-square in shape particularly at the reduced
speeds and this will result in increased heating in the rotor due to the harmonics.
The rotor harmonics also set up parasitic harmonic MMFs in the air gap and
these do interact with the stator causing some degree of stator harmonics.
However, due to the slip between rotor and stator MMF, the harmonics induced
into the stator are not directly related to the supply frequency and they are in
general of a lower relative magnitude than in the rotor.
The main effect of the rotor current harmonics is to introduce a ripple into
the torque generated by the motor and in some instances this may require
investigation to ensure that this ripple does not set up torsional resonances in
the mechanical load system. In such cases the use of a torsionally flexible
coupling will usually isolate the load from this torque ripple.
The current in the windings of the inverter transformer will contain the
normal degree of harmonics related to the mains frequency connected to the
S/ip energy recovery
379
transformer and these will have to be accepted by the mains system. However
as the size of the convertor is usually low in relation to the motor size the
amount of distortion in the total current to the motor plus the slip recovery
converters is usually quite small. Clearly the worst case will be when the rotor
current is at its highest level and this usually corresponds to operation at near
to synchronous speed.
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Section 5.2
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Section 5.3
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Section 5.5
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Section 6.3
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Section 7.2
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Section 7.3
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Section 7.5
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Section 7.6
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Sources and Effects of Power System Disturbances, May, pp. 119-122
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Section 8.3
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Section 8.4
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Section 9.1
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Conf. 3, Paper 5
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twelve-pulse cycloconverter for the speed control of induction motors', IEE Conf. Publ. 234,
May, pp. 257-260.
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noncirculating cycloconverter based on dynamic characteristics of thyristor-diode series circuit',
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122-133
Section 9.5
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Symposium, Diisseldorf, W. Germany, April, pp. 537-546
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dynamic requirements', Siemans Power Engineering, 11, No. 12, pp. 339-343
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Section 9.6
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Section 10.2
GOLDHAMMER, A. B. (1967): 'Modern Kramer drives', Electrical Review (UK), August
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use of a thyristor inverter', IEEE Trans, on Industry and General Applications, IGA-5, No. 1,
January/February, pp. 74-82
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Section 10.3
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rotor', IEEE Trans, on Power Apparatus and Systems, PAS-85, No. 1, January, pp. 76-84
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Section 10.6
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Index
Alpha angle, 143, 216, 254, 260
Amplifying gate thyristors, 65
Armature reaction, 34, 37, 42, 43, 251
Assymmetric thyristors, 65
Earth voltages, 52
Efficiency,
induction motor, 26
Excitation losses, 51
Base current, 79
Base drive circuits, 89
Beta angle, 246, 253, 261, 278, 287, 301, 361
Brushless excitation, 248
Fault conditions, 60, 100, 152
Field control, 269
Forced commutation, 70
Friction losses, 51
Capacitor,
commutation, 222, 232, 297, 300
DC link, 135, 154, 176, 184, 194
motor, 277-293
Circle diagram, 20
Collector emitter sustaining voltage, 76
Commutation,
CSI, 221
DC link, 277, 296-298
forced, 70, 279
low speed, 243
natural, 68, 107, 241, 245, 278
self, 119
Control,
sinusoidal, 330
trapesoidal, 317, 331
Convertor grade thyristors, 63, 66
Copper losses, 49
Current capabilities,
G.T.O. thyristors, 94
thyristors, 58
transistors, 77
Gamma angle, 253, 261
Gate firing, 61, 68, 100
Gear changes, 174, 190
Damper cage losses, 51
Delay angle, 113, 339
Delay time, 61
Delta connected cycloconvertor, 338
Diodes
fast, 123
feedback, 120, 133, 140, 175, 187
reactive, 120, 133, 140, 175, 187
Discontinuous current, 145
Harmonics, 45, 115, 138, 141, 164, 200, 283,
346, 357, 377
Holding current, 61
Induced voltage, 19
Inverter grade thyristors, 64, 67
Iron losses, 50
Kramer system, 349
Latching current, 61
Load commutated inverter (LCI), 239
Magnetising circuit, 47, 217
Magnetising current, 12, 17, 19
Magnetising saturation curve, 19
Magneto-motive force (MMF), 4-7, 32-39,
161, 205
Motor capacitor, 277-293
Motor magnetisation, 149, 158, 277, 303
Off-state, 55, 81, 8
On-state, 55, 77, 80
Overlap, 112, 254, 355
Overvoltage suppression, 230
Poles,
motor, 4, 28, 35
Index
Power factor,
induction motor, 9, 27, 364
motor, 253
supply, 164, 237, 273, 307, 319, 330, 345,
374-377
synchronous motor, 37, 42, 43
Pulse dropping, 172
P.W.M. - gear change, 173
P.W.M. — unsynehronised, 171
Rate of rise of voltage (dv/dt), 113
Reactive current, 139
Reactor DC link, 135, 155, 174, 203, 231, 241,
263, 277, 299, 351, 369
Regeneration, 111, 124, 126, 156, 193, 204,
247, 313
Reverse conducting thyristors, 65
Ripple current, 140, 146, 357
Rise time, 61
Rotor,
induction motor, 6
synchronous motor, 36
Safe operating area (SOA), 82
Saturation current, 17
Saturation voltage, 17
Sinusoidal control, 330
Slip, 9, 13, 15, 350
Slip compensation, 160, 197
Slip - critical, 360
Slip speed curves, 15
Snubber circuit, 68, 97
391
Speed control accuracy, 164, 201
Stator,
rotating field, 7, 251
Stator windings, 3
MMF waveforms, 6, 34
rotating field, 7, 34
Supply side convenor, 133,154,157,174,204,
241, 266
Three pulse convertor, 336
Torque production, 7, 35
Torque pulsations, 46, 205, 236, 274
Transient performance, 201
Transistor,
darlington, 80, 85
MOSFET, 85
silicon bipolar, 75, 84
Transistor parallel operation, 87
Transistor saturation, 78
Trapesoidal control, 317, 331
Turn off, 61, 81, 96
Turn on, 61, 81, 96
Vector diagram, 20-25, 39, 44
Voltage capabilities,
G.T.O. thyristors, 93
thyristors, 59
transistors, 75
Windage losses, 51
Zero current detection, 335
IET Power and Energy Series 8
This book is intended to explain the technical principles involved in
the many AC variable speed drive systems available today. It deals
with all the DC link inverter and direct AC to AC converter systems
that are in commercial use. The principles of AC motors are
considered specifically from the variable frequency point of view,
and this chapter concentrates on the effects of harmonics. The
different types of power semiconductor switches are considered
separately from the drive systems in which they are used.
A total of seven separate and technically different drive systems
are considered in such a way that their principles can be fully
understood and their performance capabilities explained. Square
wave and pulse width modulated DC link inverter systems,
cycloconverters and slip power recovery drives are all included in
this comprehensive book.
This book has been written so that it can be understood by general
engineers, not just by experts in the field. It should therefore be
of great use to any engineer involved with variable speed drives in
any capacity. It should also be of interest to university and college
electrical engineering departments and students.
David Finney, B.Sc., CEng., FIEE, is
division manager and chief engineer,
responsible for large variable speed drive
systems, at the G.E.C. Industrial Controls
plant in Rugby, England. In this position he
is responsible for the development, design
and manufacture of large drive systems
for use in mining, metals, paper, oil, and
chemical industries throughout the world.
He has been technically involved in the
power semiconductor field since 1958,
when thyristors were only just emerging,
and during this time he has worked on all
types of thyristor converters and inverter
drives from a few kilowatts up to 10,000
kW using natural and forced commutation
techniques and operating in square wave
and pulse modulated modes.
He has published a number of articles
and given lectures around the world in his
chosen subject.
Variable Frequency
AC Motor Drive Systems
Variable Frequency
AC Motor Drive Systems
David Finney
Finney
The Institution of Engineering and Technology
www.theiet.org
0 86341 114 2
978-0-86341-114-4
Variable Frequency
AC Motor Drive
Systems